US20040190645A1 - System for processing digitized communication signals provided over plural RF channels - Google Patents

Abstract

A wireless digital subscriber communications system includes a base station and a subscriber unit and uses communications signals provided over a plurality of radio frequency channels. A processor provides an output phase signal corresponding to a selected output digital frequency, and a lookup table is used having two sets of predefined stored values pertaining to the amplitude of a signal for a single quadrant. In particular embodiments, predefined stored values include coarse angle approximations and fine angle approximations and a sine and cosine generator receives the phase signal and generates sine and cosine waveforms utilizing amplitude values obtained from the lookup table. In a further embodiment, the phase signal includes phase data and specifies the quadrant and the algebraic sign of the phase data, with the sine and cosine generator accessing the lookup table differently depending upon the quadrant and sine of the phase data, such that the lookup table provides an amplitude value from the sets of predefined stored values based on the phase data. A modulator combines the sine and cosine waveforms to produce the selected output digital frequency and modulates digital frequency.

Description

BACKGROUND
• This application is a continuation of U.S. application Ser. No. 10/412,456, filed Apr. 11, 2003, which is a continuation of U.S. application Ser. No. 10/223,750, filed Aug. 20, 2002, now U.S. Pat. No. 6,587,516, issued Jul. 1, 2003, which is a continuation of U.S. application Ser. No. 09/593,307, filed Jun. 13, 2000, now U.S. Pat. No. 6,449,317, issued on Sep. 10, 2002, which is a continuation of U.S. application Ser. No. 09/228,140, filed Jan. 11, 1999, now U.S. Pat. No. 6,078,629, issued on Jun. 20, 2000; which is a continuation of U.S. application Ser. No. 08/881,339, Jun. 24, 1997, now U.S. Pat. No. 5,859,883, issued on Jan. 12, 1999; which is a continuation of U.S. application Ser. No. 08/587,008, filed Jan. 11, 1996, now U.S. Pat. No. 5,694,430, issued Dec. 2, 1997; which is a continuation of U.S. application Ser. No. 08/445,082, filed May 22, 1995, now U.S. Pat. No. 5,644,602, issued Jul. 1, 1997; which is a divisional of application Ser. No. 08/222,670, filed Apr. 4, 1994, abandoned; which is a continuation of U.S. application Ser. No. 07/940,662, filed Sep. 4, 1992, now U.S. Pat. No. 5,325,396, issued Jun. 28, 1994; which is a continuation of U.S. application Ser. No. 07/658,065, filed Feb. 20, 1991, Now U.S. Pat. No. 5,146,473, issued Sep. 8, 1992; which is a continuation of U.S. application Ser. No. 07/394,497, filed Aug. 14, 1989, now U.S. Pat. No. 5,008,900, issued Apr. 16, 1991, which are incorporated by reference as if fully set forth.[0001]
FIELD OF INVENTION
• The present invention generally pertains to subscriber communications systems and is particularly directed to an improved subscriber unit for wireless communication with a base station in a wireless digital subscriber communication system.[0002]
• A typical subscriber unit is described in U.S. patent application Ser. No. 06/893,916 filed Aug. 7, 1986 by David N. Critchlow et al. Now U.S. Pat. No. 4,825,448. A base station used with such a subscriber unit in a wireless digital subscriber communication system is described in U.S. Pat. No. 4,777,633 to Thomas E. Fletcher, Wendeline R. Avis, Gregory T. Saffee and Karle J. Johnson. The subscriber unit described in U.S. Pat. No. 4,825,448 includes means for transcoding a digital voice input signal to provide digital input symbols; means for FIR filtering the digital input symbols; means for deriving an analog intermediate frequency input signal from the filtered input symbols; means for combining the intermediate frequency input signal with an RF carrier for radio transmission to the base station; means for demodulating an output signal received from the base station to provide digital output symbols; and means for synthesizing a digital voice output signal from the digital output symbols. The subscriber unit includes a baseband processor chip and a modem processor chip. Both are TMS32020 digital signal processors. The baseband processor chips perform the transcoding of the digital voice input signal, the synthesis of the digital output symbols, and various baseband control functions; and the modem processor chip performs the FIR filtering of the digital input symbols, and the demodulation of the output signal received from the base station. The modem processor chip generally acts as the master for the system.[0003]
SUMMARY OF THE INVENTION
• A wireless digital subscriber communications system includes a base station and a subscriber unit and uses communications signals provided over a plurality of radio frequency channels. A processor provides an output phase signal corresponding to a selected output digital frequency, and a lookup table is used having two sets of predefined stored values pertaining to the amplitude of a signal for a single quadrant. In particular embodiments, predefined stored values include coarse angle approximations and fine angle approximations and a sine and cosine generator receives the phase signal and generates sine and cosine waveforms utilizing amplitude values obtained from the lookup table. In a further embodiment, the phase signal includes phase data and specifies the quadrant and the algebraic sign of the phase data, with the sine and cosine generator accessing the lookup table differently depending upon the quadrant and sine of the phase data, such that the lookup table provides an amplitude value from the sets of predefined stored values based on the phase data. A modulator combines the sine and cosine waveforms to produce the selected output digital frequency and modulates digital frequency.[0004]
BRIEF DESCRIPTION OF THE DRAWINGS
• FIG. 1 is a block diagram of a preferred embodiment of the subscriber unit of the present invention.[0005]
• FIG. 2 is a block diagram of the FIR chip included in the embodiment shown in FIG. 1.[0006]
• FIG. 3 is a block diagram of the DIF chip included in the embodiment shown in FIG. 1.[0007]
• FIG. 4 illustrates the processing tasks performed by the processor chip shown in the embodiment of FIG. 1.[0008]
• FIG. 5 illustrates the processing routines included in modem processing task shown in FIG. 4.[0009]
DEFINITION OF ABBREVIATIONS AND ACRONYMS
• The following is a definition of abbreviations and acronyms used herein:[0010]

  	A/D	Analog to Digital
  	AGC	Automatic Gain Control
  	ASIC	Application Specific Integrated Circuit
  	BPSK	Binary Phase Shift Keying
  	CCT	Channel Control Task
  	CCU	Channel Control Unit
  	CRC	Cyclic Redundancy Check
  	DAC	Digital to Analog Converter
  	DDS	Direct Digital Synthesizer
  	DIF	Digital Intermediate Frequency
  	DIP	Dual In-line Package
  	DOR	Data Output Ready
  	DPSK	Differential Phase Shift Keying
  	DSP	Digital Signal Processing
  	EPROM	Erasable Read Only Memory
  	FIR	Finite Impulse Response
  	I/O	Input/Output
  	LSB	Last Significant Bit
  	MPT	Modem Processing Task
  	MSB	Most Significant Bit
  	MUX	Multiplexer
  	PCM	Pulse Code Modulation
  	PLL	Phase Locked Loop
  	PWM	Pulse Width Modulation
  	QPSK	Quadrature Phase Shift Keying
  	RAM	Random Access Memory
  	RCC	Radio Control Channel
  	RELP	Residual Excited Linear Predictive
  	RF	Radio Frequency
  	ROM	Read Only Memory
  	RX	Receive
  	RXCLK	Receive Clock
  	RXSOS	Receive Start of Slot
  	SCT	Subscriber Control Task
  	SLIC	Subscriber Line Interface Circuit
  	SPC	Signal Processing Control
  	SPT	Signal Processing Task
  	SPTCTL	Signal Processing Task Controller
  	SSB	Switch-hook Sample Buffer
  	TDM	Time Division Multiplexing
  	TX	Transmit
  	TXCLK	Transmit Clock
  	UART	Universal Asynchronous Receiver Transmitter
  	VLSI	Very Large Scale Integration
  	XOR	Exclusive Or

DESCRIPTION OF THE PREFERRED EMBODIMENT
• Referring to FIG. 1, a preferred embodiment of the subscriber unit of the present invention includes a[0011]telephone interface circuit10, a SLIC andcodec circuit11, aprocessor chip12, afast memory13, aslow memory14, anaddress decoder15, aFIR chip16, aDIF chip17, aDAC18, an A/D converter19, aradio20, aringer circuit21, and anoscillator22.
• The[0012]FIR chip16, which is an ASIC chip, is interfaced with theDIF chip17 bylines23 and24, to theprocessor chip12 byprocessor bus25 andline26, to the A/D converter19 byline27, to the SLIC andcodec circuit11 byline29, to theradio20 byline30, and to theringer circuit21 byline31.
• The[0013]telephone interface circuit10 is interfaced with atelephone32, which converts sound waves into an input voice signal, and converts an output voice signal into sound waves.
• The SLIC and[0014]codec circuit11 is coupled to thetelephone interface circuit10 for converting the input voice signal into a baseband digital input signal, which is provided to theprocessor chip12.
• In an alternative embodiment (not shown), the processor chip is also interfaced directly with an UART for alternatively receiving digital input signals directly from and sending digital output signals directly to a digital signal I/O device.[0015]
• The[0016]processor chip12 includes a model TMS320C25 digital signal processor, which transcodes the baseband digital input signal in accordance with a RELP algorithm to provide TX data digital input symbols on theprocessor bus25. The use of a digital signal processor to perform a RELP algorithm is described in International Patent Application No. PCT/US85/02168, International Publication No. WO 86/02726, published May 9, 1986.
• The[0017]FIR chip16 FIR filters the digital input symbols and provides I,Q data to theDIF chip17 onlines24.
• The[0018]DIF chip17 interpolates the filtered digital input symbols, and modulates a digital intermediate frequency signal with the interpolated input symbols to provide a modulated digital input signal.
• The[0019]DAC18 converts the modulated digital input signal into a modulated analog input signal.
• The[0020]radio20 transmits the modulated analog input signal to the base station; and receives and demodulates a modulated analog output signal from the base station.
• The[0021]oscillator22 is a free running oscillator, that provides clock signals for theprocessor chip12.
• A description of the relationship between the subscriber unit and the base station is contained in U.S. Pat. No. 4,777,633.[0022]
• The A/[0023]D converter19 converts the demodulated received analog output signal into a digital output signal containing digital output symbols.
• The[0024]processor chip12 synthesizes a baseband digital output signal from the digital output symbols. Synthesis of RELP transcoded symbols by a digital signal processor also is described in International Publication No. WO 86/02726. Theprocessor chip12 further performs echo cancellation as described in U.S. Pat. No. 4,697,261 to David T. K. Wang and Philip J. Wilson.
• The SLIC and[0025]codec circuit11 converts the baseband digital output signal into the output voice signal that is provided by the telephone interface circuit to thetelephone32.
• The[0026]FIR chip16 consolidates circuit functionality into a VLSI device in order to reduce production cost of the subscriber unit by eliminating many separate medium scale integration parts.
• Referring to FIG. 2, the[0027]FIR chip16 includes afanout buffer33, aninternal decoding module34, anRX sample buffer35, control and status registers36, an externaladdress decoding module37, awatchdog timer module38, anRX timing module39, aTX timing module40, aTX FIR filter42, acodec timing module44, and aringer control module45.
• The[0028]FIR chip16 provides45 millisecond frame marker generation, 11.25 millisecond slot marker generation, 16 KHz symbol clock generation, timing adjustment circuits, RX sample buffering, TX symbol buffering, 8 KHz codec timing generation, processor interface decoding, ringer timing generation, external address decoding and watchdog timer reset generation. TheFIR chip16 buffers two 5-bit TX symbols at a 8 KHz rate. TheFIR chip16 converts and filters the TX symbols into I and Q data symbols, with each such symbol being 10-bits at a rate of 160 KHz. The I and Q data are interleaved and output to theDIF chip17 at a rate of 320 KHz. TheFIR chip16 also buffers RX data samples at a 64 KHz rate; and four RX data samples are read by theprocessor chip12 at a 16 KHz rate. Timing clocks and signals are generated by theFIR chip16 from an incoming 3.2 MHz master clock signal. Theprocessor chip12 is synchronized to these data rates by slot and symbol interrupts generated by theFIR chip16. The codec and processor 8 KHz timing strobe and codec clock are generated by theFIR chip16 and synchronized to the time of the incoming RX samples. TheFIR chip16 also generates control and timing signals for controlling the shape and timing of the ringing voltage provided by theringer circuit21. Thewatchdog timer module38 provides a reset signal in the event that theprocessor chip12 does not execute instructions properly.
• The[0029]fanout buffer33 buffers a 3.2 MHz master clock signal received online23afrom theDIF chip17, an advanced 3.2 MHz clock signal received online23bfrom theDIF chip17, and a reset signal received online51 from thewatchdog timer38. Unless otherwise indicated, all timing within theFIR chip16 is derived from the 3.2 MHz clock signal online23a. The advanced 3.2 MHz clock signal online23bleads the 3.2 MHz clock signal online23aby one cycle of a 21.76 MHz reference signal that is present within theDIF chip17. The 3.2 MHz clock signal is derived from the 21.76 MHz reference in theDIF chip17 and the minimum pulse width is therefore 276 nanoseconds. The advanced 3.2 MHz clock signal fromline23bis provided from thebuffer33 viainternal line47 to theTX FIR filter42, and thecodec timing module44. TheTX FIR filter42 is implemented in part by a ROM, which is pseudo-static and requires its enable input to be deactivated by the advanced 3.2 MHz clock signal online47 between successive accesses.
• The HW reset signal on[0030]line51 resets all internal circuits of theFIR chip16 and provides a hardware reset to the modules of FIG. 1.
• The internal clocks are either buffered versions of the 3.2 MHz master clock signal received on[0031]line23aor divisions of this clock.
• The internal[0032]address decoding module34 allows theprocessor chip12 to access the internal functions of theFIR chip16 for the purpose of controlling such functions and determining their status. The internaladdress decoding module34 receives processor addresses and processor strobes onbus25. The internaladdress decoding module34 provides output signals on internal bus48.
• The output signals on bus[0033]48 from the internaladdress decoding module34 include a read enable signal to theRX sample buffer35, a control write signal and status read signals to the control and status registers36, a write signal to theTX FIR filter42, slot and clock write signals to theRX timing module39, a write signal to theTX timing module40, and control signals to the TXFIR filter module42 and theRX sample buffer35, and an AM Strobe signal, which causes theRX timing module39 to reset slot timing. Only one of the respective read or write signals on bus48 from the internaladdress decoding module34 is active at any one time.
• The[0034]RX sample buffer35 receives four samples for each RX symbol time from the A/D converter19 vialine27aat a 64 KHz rate; buffers up to two symbols of data, which is eight samples total; and then sends such data samples to theprocessor chip12 via theprocessor bus25. TheRX sample buffer35 is implemented in a dual-page RAM. TheRX sample buffer35 receives a read enable signal on internal bus48 from the internaladdress decoding module34 and a write strobe signal oninternal line49 from theRX timing module39.
• The control and status registers[0035]36 allow theprocessor chip12 to control the internal functions of theFIR chip16, and allow theprocessor chip12 to read the status of theTX FIR filter42 andRX sample buffer35, and other internal signals. The control signals are provided by theprocessor chip12 via theprocessor bus25 and the status indications are derived from various internal modules of theFIR chip16. The status indications are provided to theprocessor chip12 via theprocessor bus25. The status indications are RX Underrun RX Overrun, TX Underrun, TX Overrun, Start-of-Frame, RX Start of slot, TX symbol Clock, RX Symbol Clock and TX FIR filter Overflow. The control signals, which are provided by the control registers36 to the internal circuits via the internal bus48, include the following: TX Enable, Modulation Level, Ringer Enable, Software Reset, Tristate, and Watchdog Strobe.
• The TX Enable signal indicates the beginning of a TX slot based upon the TX delay established in the[0036]TX timing module40.
• The Modulation Level signal is provided to the[0037]RX timing module39 and determines whether a slot length is 180 or 360 symbols.
• The Software Reset signal allows the[0038]processor chip12 to reset internal functions within theFIR chip16.
• The Tristate signal allows the[0039]processor chip12 to disable the outputs of theFIR chip16.
• The Ringer Enable signal allows the[0040]processor chip12 to turn theringer circuit21 on and off. This signal provides a two-second and four-second cadence for the ringing signal.
• The Watchdog Strobe allows the[0041]processor chip12 to reset the watchdog timer module in order to keep a hardware reset from occurring.
• The[0042]processor chip12 receives a RX clock interrupt (RXCLKINT) signal from theRX timing module39 vialine26cwhen data has been written into the first four locations of the dual-page RAM of theRX sample buffer35. Theprocessor chip12 then reads the RX samples from the first four locations of the dual-page RAM viaprocessor bus25. At this time samples are being written into the next four locations of the dual-page RAM at a 64 KHz rate. The 16 KHz event is a derivative of the 64 KHz event, which keeps the read and write events synchronized. This ensures that read and write operations do not occur at the same time at any one memory location and also ensures adequate response time from theprocessor chip12.
• A TX symbol buffer in the[0043]TX FIR filter42 receives TX symbols from theprocessor chip12 via theprocessor bus25 and buffers up to two TX symbols. Theprocessor chip12 is interrupted every other TX symbol time to write two more symbols into the TX symbol buffer.
• The TX symbol buffer in the[0044]TX FIR filter42 receives a write signal via the internal bus48 from the internaladdress decoding module34.
• After each TX clock interrupt (TXCLKINT) signal at 8 KHz on[0045]line26a, theprocessor chip12 writes out two 5-bit TX symbols. The data is in a DPSK gray code format. The TX symbol buffer outputs a symbol every 16 KHz for processing by theTX FIR filter42. This data is double buffered due to an asynchronism between theFIR chip16 and theprocessor chip12. The last data value is repeated until new data is written. Null data can be repeated in this manner. The TX symbol buffer is cleared during a reset.
• During training, a fixed sequence of symbols is sent to the[0046]FIR chip16 by theprocessor chip12. TheFIR chip16 performs FIR filtering on these symbols and outputs I,Q pairs to theDIF chip17.
• The[0047]radio20 loops the data back to theAID converter19. The samples are read by theprocessor chip12 as in the on-line mode and the coefficients of the processor RX filter implemented in theprocessor chip12 are adjusted. The only timing critical for training is generated by the RX andTX timing modules39,40.
• The[0048]RX timing module39 generates all reference clocks and strobes for processing the RX symbols. The timing is adjusted by theprocessor chip12 so that processing can be synchronized to the RX samples received vialine27afrom the base station. TheRX timing module39 includes an RX clock fractional timing circuit and an RX Slot timing circuit. The purpose of these two circuits is to synchronize the modem receive timing within theprocessor chip12 to the RX samples received online27afrom the base station, and via the A/D converter19, and also to regulate theTX timing module40 and thecodec timing module44.
• The[0049]RX timing module39 is clocked at a 3.2 MHz rate and receives the following control signal inputs from theprocessor chip12 via the processor bus25: an AM Strobe signal, an RX Slot Clock Write signal, and an RX Bit Tracking signal.
• Several outputs are generated by the[0050]RX timing module39. A 64 KHz write strobe is provided online49 to control writing to theRX sample buffer35. A 64 KHz A/DSYNC strobe signal is provided online27bto the A/D converter19 to synchronize the operation thereof. A 8 KHZ strobe signal also is provided to thecodec timing module44 vialine52. A 16 KHZ RX clock interrupt (RXCLKINT) signal online26cand RX start-of-slot interrupt (RXSOSINT) signal online26bare output to theprocessor chip12. A pre-RX slot timing strobe is provided online54 to control theTX timing module40.
• The fractional timing circuit in the[0051]RX timing module39 is set by theprocessor chip12 to generate the RX start of slot interrupt signal online26b. Theprocessor chip12 determines the location of an AM hole (strobe signal) transmitted by the base station during acquisition. When theprocessor chip12 detects the AM strobe signal, the slot timing circuit in theRX timing module39 is reset by a reset signal from theprocessor chip12. This aligns the frame and slot markers to the AM strobe signal. The frame marker is a 62.5 μsec pulse occurring every 45 milliseconds. The slot marker is a 62.5 μsec pulse repeating every 11.25 millisecond, or 22.5 milliseconds when in a QPSK mode.
• The incoming RX symbols are demodulated by the[0052]processor chip12 and timing is further adjusted if necessary. To adjust the 16 KHz RX symbol clock the processor chip forces the fractional timing (bit tracking) circuit to shorten or lengthen the 64 KHz strobe by up to fifty 3.2 MHz cycles.
• The[0053]processor chip12 monitors the relationship of the RX symbols to the frame timing and makes adjustments to the 16 KHz RX clock accordingly. When the RX clock is adjusted the slot and frame markers are changed also because they are a derivative of the RX clock.
• To keep the number of Pulse Code Modulated (PCM) samples provided to and from the SLIC and[0054]codec circuit11 synchronized to the frame timing, theRX timing module39 controls thecodec timing module44.
• The[0055]TX timing module40 includes a TX delay circuit and a TX control timing circuit. These circuits generate a TX clock interrupt (TXCLKINT) signal which is provided to theprocessor chip12 vialine26a. TheTX timing module40 is synchronized to theRX timing module39 by the pre-RX slot timing strobe, which is provided to the TX timing module by theRX timing module39 online54 and used to reset the TX delay circuit, which in turn generates the TX slot marker. Timing of the TX clock is based on the internal 3.2 MHz clock.
• The[0056]processor chip12 also controls the TX delay and TX timing circuits by providing TX data write control signals over theprocessor bus25. TheTX timing module40 provides a T/R control signal online30 to theradio20. This signal determines whether the radio is transmitting or receiving data.
• The[0057]TX timing module40 also controls TX symbol shifting, ROM addressing, accumulation timing, and I,Q product storage for output to theDIF chip17.
• The[0058]TX timing module40 provides control signals online56 for keeping theTX FIR filter42 synchronized to the TX symbol and slot timing. Such synchronization is accomplished in accordance with the TX slot timing marker. After a reset, theTX timing module40 actively generates control signals ontoline56 once a TX slot begins.
• The[0059]TX FIR filter42 module includes a ROM, which implements a FIR filter by providing I and Q data products in response to the ROM being addressed for lookup by a combination of TX symbols received from theprocessor chip12 via theprocessor bus25 and SINE and COSINE coefficient counts provided by a counter within the TXFIR filter module42. TheTX FIR filter42 accumulates six sequential I and Q data produces and stores results for output to theDIF chip17 vialine24a.
• The minimum frequency required for operation of the[0060]TX FIR filter42 is determined by the symbol rate (16 KHz) times the number of I and Q samples (2) times the number of coefficients (10) times the number of taps (6)=1.92 MHz. The master clock of 3.2 MHz meets this minimum frequency requirement. Wait periods are added to compensate for the faster execution time.
• The[0061]TX timing module40 is clocked at a 3.2 MHz clock rate, which defines one state period. Because this clock rate is greater than the required minimum of 1.92 MHz theTX FIR filter42 generates signals for the first six out of ten state periods.
• Each new TX symbol must be loaded into a circular buffer in the[0062]TX FIR filter42 at the rate of 16 KHz. The new TX symbol and the previous five TX symbols are stored in the circular buffer. The oldest TX symbol is dropped when a new TX symbol is shifted in. TheTX FIR filter42 output rate is 320 KHz. From each TX symbol, ten I data values are generated and ten Q data values are generated. Table 1 below shows how I, Q and null information can be derived from each 5-bit value.

  	TABLE 1
  	BIT 1	BIT 2	BIT 3	BIT 4	BIT 5
  	I & Q LSB	I & Q	I MSB	Q MSB	NULL

• The data in the circular buffer is rotated every 6 out of 10 states. One new TX symbol and the five previous TX symbols reside in the circular buffer for twenty of these ten state periods. The coefficient portion of the ROM address is also increased every six out of ten state periods. An accumulator in the[0063]TX FIR filter42 adds the results of each I-data product provided from the ROM for each of the six state periods. Therefore the accumulator register is cleared for the first addition, and each successive addition result is clocked into a feed back register of the accumulator so it can be added to the newly looked-up product. Once six additions occur the result is clocked into an output shift register. The same process occurs for the same coefficients and the Q-data products provided from the ROM for each TX symbol.
• The ROM address lines allow sixty COS coefficient and sixty SIN coefficient lookups for four possible I,Q data indexes. This requires seven address lines for coefficients and two address lines for I,Q data. The output of the FIR filter requires 10 bits. Two extra bits are required to maintain accuracy of the fractional portion of the lookup value. This makes the ROM size 512.times.12. The MSB of the I,Q data index is passed around the ROM to a 1's complement circuit which forces the output of the ROM to be inverted or not inverted.[0064]
• If the symbol addressing the ROM is a null symbol the null bit controls four of the seven coefficient address lines. Since seven address lines are used for coefficient lookup this provides 128 locations. Only 120 coefficients are needed. This leaves eight unused locations. Zero values are stored in these locations so null information can be easily output from the ROM.[0065]
• A 2's complement function is implemented by using a 1's complement and carrying in a logic 1 in the succeeding adder. The output of the adder is wrapped around to the input of the adder for successive additions or output through a MUX to an output shift register. The output is rounded off by using only the ten upper bits.[0066]
• The circular buffer outputs of the TX FIR filter are set to zero after a reset. This allows null information to be processed until new TX symbol values are loaded. I data is first processed followed by Q data.[0067]
• The TX Clock interrupt signal only occurs during a TX slot. The processor does not know when a TX slot begins or ends except by responding to this interrupt. The signal has an active low duration of one 3.2 MHz clock cycle to guarantee that the interrupt is not active once it has been serviced. The TX Clock interrupt occurs every other symbol time (16 KHz/2).[0068]
• The RX Clock interrupt occurs for a full frame. The[0069]processor chip12 masks out this interrupt by using the RX Slot marker as a mask. The RX Clock interrupt has an active low duration of one 3.2 MHz clock cycle.
• The RX Start of Slot interrupt occurs every 11.25 milliseconds, and has an active low duration of one 3.2 MHz clock cycle.[0070]
• Each interrupt signal is forced to an inactive high state upon reset.[0071]
• The[0072]codec timing module44 generates timing strobes and sends the necessary clock signal vialines29 to the SLIC andcodec circuit11 to cause 8 bits of data to be transferred between the codec and processor at an 8 KHz rate. Thecodec11 receives and transmits 8 bits of data every 8 KHz. Thecodec timing module44 sends a codec clock signal online29aand a codec sync signal online29b. The codec clock signal online29ais generated at a rate of 1.6 MHz by dividing the advanced 3.2 Mhz clock by two. An 8 KHz pulse of one 3.2 MHz period is received from theRX timing circuit39 and is reclocked to occur for one 1.6 MHz period, and thus is guaranteed to occur with respect to the 1.6 MHz clock rising edges. With these two signals, transfer of PCM data between thecodec11 and theprocessor chip12 is accomplished. This allows the subscriber PCM data to be synchronized to the base station PCM data.
• The[0073]ringer control module45 responds to a ring enable control signal originating in theprocessor chip12 and provided from the control and status register36 on internal bus48 by generating a 20 Hz square wave signal online31aand two 80 KHz phase control signals, PHASEA online31band PHASEB online31cand sending these signals to theringer circuit21. The 20 Hz square wave signal online31acontrols the polarity of the ringer voltage provided by theringer circuit21 to thetelephone interface circuit10. The 80 KHz phase signals onlines31band31ccontrol the pulse width modulated power source in theringer circuit21.
• A reset or a SLIC ring command signal on[0074]line29cfrom the SLIC portion of the SLIC andcodec circuit11 turns off or overrides these signals onlines31a,31b, and31cafter the ring enable signal originating in theprocessor chip12 has turned them on. This ensures that the ringer is off if a reset occurs or the telephone hand set is taken off hook.
• Since the[0075]ringer circuit21 generates a high voltage and dissipates much power, this voltage is not generated except when requested by theprocessor chip12.
• The external[0076]address decoding module37 generates chip selects onto theprocessor bus25 that are used by theprocessor chip12 to access theDIF chip17, the UART hardware, and theslow memory EPROMs14 in separate distinct address segments. Theprocessor chip12 provides eight MSB address lines, data space, and program space signals. These are decoded to generate the appropriate chip selects.
• The[0077]watchdog timer module38 generates a 50 millisecond hardware reset pulse online51, which resets allFIR chip16 modules and all subscriber unit modules in FIG. 1. Thewatchdog timer module38 generates a pulse if it is not reset within a 512 millisecond period by the Watchdog strobe signal provided on bus48 by the control and status registers36.
• The[0078]DIF chip17 is interfaced to theprocessor chip12 by theprocessor bus25, to theFIR chip16 bylines23 and24, to theDAC18 byline71 and to an oscillator in theradio20 byline72.
• The oscillator in the[0079]radio20 provides a 21.76 MHz master clock signal online72 to theDIF chip17.
• Referring to FIG. 3, the[0080]DIF chip17 includes aclock generator60, aprocessor decoding module61, a FIRchip interface module62, aninterpolator63, acontrol register64, tuning registers65, aDDS phase accumulator66, a DDS SIN andCOS generation module67, amodulator68 and anoise shaper69. In combination theDDS phase accumulator66 and the DDS SIN,COS generator67 constitute a direct digital synthesizer (DDS) for digitally synthesizing a digital intermediate frequency signal.
• The[0081]DIF chip17 is an ASIC chip, which is mapped as processor data memory.
• The[0082]DIF chip17 operates in one of two operating modes, a modulated carrier generation mode, and a pure carrier mode. In the modulated carrier generation mode, baseband data is input in the I,Q domain and this data is used to modulate the pure carrier generated by the DDS function of theDIF chip17. In the Pure Carrier Generation mode, the baseband data inputs are ignored and an unmodulated carrier from the DDS is provided to theDAC18.
• The[0083]clock generator60 generates all timing and clocks within theDIF chip17 and also generates the 3.2 MHz clock signal and the advanced 3.2 MHz clock signal that are provided to theFIR chip16 onlines23aand23b. The two primary timing signals used within theDIF chip17 are a 21.76 MHz clock and a 2.56 MHz interpolation gate signal. The 3.2 MHz clock is used internally to shift I and Q data online24afrom theFIR chip16 into theFIR interface module62.
• The[0084]clock generator60 buffers the 21.76 MHz clock received online72 from the oscillator in theradio20 and provides a buffered 21.76 clock signal online71a. Such buffering is done to provide sufficient drive capability for internal functions and to minimize clock skew. The buffered 21.76 MHz clock also provides a clock for theDAC18 and other external circuitry.
• The[0085]clock generator60 provides the 3.2 MHz clock signal by dividing the 21.76 MHz clock by 6 and by 8 in the following sequence: 6-8-6-8-6, which thereby results in an average divisor of 6.8 (21.76÷6.8=3.2). The effect of this per cycle variation is a minimum period of 276 ns and a maximum period of 368 ns. An advanced version of the 3.2 MHz clock signal is also generated as the advanced 3.2 MHz clock signal online23b. Both clocks are identical with the exception that the ROM deselect signal online23bleads the 3.2 MHz clock signal online23aby one 21.76 MHz clock cycle.
• The[0086]clock generator60 provides the 2.56 MHz gate signal oninternal line74 by dividing the 21.76 MHz clock by 8 and 9 in an even sequence (8-9-8-9- . . . ), which thereby results in an average divisor of 8.5 (21.76÷8.5=2.56 MHz). This signal is used by theinterpolator63 and themodulator68.
• The[0087]processor decoding module61 allows the processor to control all internal functions of theDIF chip17. Theprocessor decoding module61 decodes processor addresses and processor strobes received from data space on theprocessor bus25 to provide internal write strobes, which are provided oninternal bus76 to thecontrol register64 and the tuning registers65 to enable theprocessor chip12 to write control and configuration data. Only one output from theprocessor decoding module61 is active at any given time. The processor addresses determine which output is generated. If a function within theDIF chip17 address space is chosen, a chip select signal online24cfrom theFIR chip16 becomes active.
• The[0088]FIR interface module62 receives the I and Q samples from theFIR chip16 online24ain a serial format and converts them into 10-bit parallel format in which they are provided to the interpolator module online77. The I,Q gate signal online24bfrom theFIR chip16 is used to distinguish the I data from the Q data. TheFIR interface module62 also subtracts previous I and Q samples from current samples to form a ΔI and ΔQ samples which are then shifted right 4 places (÷16) to form the correct increment for the interpolator module online78. Since theFIR interface module62 supplies data to theinterpolator63, a sync signal is sent by theFIR interface module62 to theclock generator60 to synchronize the 2.56 MHz gate pulse provided online74.
• The[0089]interpolator63 accumulates the ΔI,Q at a 160 KHz×16=2.56 MHz rate and provides interpolated I and Q samples to themodulator68 on lines80 and81 respectively. Theinterpolator63 performs a ×16 linear interpolation in order to reduce the 160 KHz sampling spurs present in the baseband data received from theFIR chip16.
• The[0090]interpolator63 successively accumulates the ΔI and ΔQ samples to generate an output at a 2.56 MHz rate. At the end of an accumulation cycle (16 iterations), the output of the interpolator should be equal to the current I and Q samples. This is critical since the next accumulation cycle starts its cycle with the current data. To ensure that the data is correct, during the last accumulation cycle the current I and Q data are input directly to the interpolator output register in place of the output of the adder (which should have the same data).
• The control registers[0091]64 are used to control and configure theDIF chip17 and to select the operating modes. All of the control registers64 are loaded by theprocessor chip12 via theprocessor bus25.
• There are three control registers[0092]64. The first control register registers a CW MODE signal, an AUTO TUNE H-L signal, and an AUTO TUNE L-H signal. The second control register registers a SIGN SELECT signal, an OUTPUT CLOCK PHASE SELECT signal, an INTERPOLATOR ENABLE signal, a SERIAL PORT CLOCK SELECT signal, a SERIAL/PARALLEL MODE SELECT signal and a QUADRATURE ENABLE signal. The control functions associated with these signals are described later at the conclusion of the description of the other modules of theDIF chip17.
• The third control register enables and specifies the coefficients for the[0093]noise shaper69.
• There are three 8-bit tuning registers[0094]65 for storing 24 bits of phase increment data to specify the frequency of the DDS. This provides a 24-bit tuning word which allows a frequency resolution of (sample rate)/2₂₄=21.76 MHz/2₂₄=1.297 Hz. The output frequency of the DDS is equal to the resolution multiplied by the 24-bit tuning word.
• The tuning registers[0095]65 are loaded by theprocessor chip12 via theprocessor bus25. The tuning word is double buffered by the tuning registers65 so that theprocessor chip12 can write data to these registers freely without affecting the current DDS operation.
• The tuning word in loaded from buffer tuning registers into output tuning registers whenever a TUNE command is issued. The TUNE command is synchronized to the 21.76 MHz clock to provide a synchronous transition.[0096]
• The[0097]DDS phase accumulator66 performs a modulo2₂₄accumulation of the phase increment provided online82 by the tuning registers65. The output of thephase accumulator66 represents a digitized phase value which is provided online83 to the DDS SIN andCOS generator67. The DDS SIN andCOS generator67 generates a sinusoidal function. A DDS works on the principle that a digitized waveform may be generated by accumulating phase chances at a higher rate.
• The tuning word, which will be different for different subscriber units, represents a phase change to the[0098]phase accumulator66. The output of theaccumulator66 can range from 0 to 2₂₄−1. This interval represents a 360 degree phase change. Although theaccumulator66 works in standard binary, this digitized phase representation can be input to a waveform generator to produce any arbitrary waveform. In theDIF chip17, the DDS SIN andCOS generators67 produce SIN and COS functions onlines84 and85 respectively.
• The period of the waveform function is based on the time required to perform a summation to the accumulator upper limit (2[0099]₂₄−1). This means that if a large phase increment is provided, then this limit will be reached sooner. Conversely, if a small increment is given then a longer time is required. Thephase accumulator66 performs a simple summation of the input phase increment and can be represented by the following equation:$\begin{array}{cc}{\phi }_T=\sum _{i=1}^n{\phi }_{\mathrm{inc}}& \mathrm{Equation}1\end{array}$

  
• Where n is the number of iterations, and Φ[0100]_incis simply the data provided online82 from the tuning registers65.
• In the embodiment of the[0101]DIF chip17 described herein, the value of .phi..sub.T is constrained by the accumulator length to be a maximum of 2.sup.24. Therefore the current phase may be described as:
• Φ_t=(Φ_t−1+Φ_inc)modulo 2²⁴  Equation 2
• Since the accumulation clock is fixed to be the master 21.76 MHz input clock this results in a complete cycle taking 2.sup.24/.phi..sub.inc iterations at a per iteration period of 1/21.76 MHz. So the entire cycle takes the following amount of time:[0102]$\frac{2^{24}}{21.76{\mathrm{MH}}_Z•{\phi }_{\mathrm{inc}}}$

  
• Since this period represents a 360 degree cycle, the reciprocal of this expression represents a frequency. The DDS frequency is therefore[0103]$\begin{array}{cc}{f}_{\mathrm{DDS}}=\frac{21.76{\mathrm{MH}}_Z•{\phi }_{\mathrm{INC}}}{2^{24}}& \mathrm{Equation}3\end{array}$

  
• In the DDS SIN,[0104]COS generation module67, the SIN and COS waveforms are generated so a complex mixing may be performed in the modulator. Each is generated by two lookup tables representing a coarse and fine estimate of the waveform. The two values are added to form composite 12-bit signed 2's complement SIN and COS data output signals onlines84 and85. The lookup tables are implemented in ROM's that are addressed by the fourteen most significant bits of the signal online83 from theDDS phase accumulator66.
• It is desired to have as much phase and amplitude resolution as is practical. In the[0105]DIF chip17 design, 14 bits of phase input and 12 bits of amplitude data output are provided in the waveform generation section. If a “brute-force” approach were taken to generate this data then very large tables would be needed to generate all possible phase and amplitude values (e.g. 16K words×12 bits each). To minimize the table size, theDIF chip17 makes use of quadrant symmetry and trigonometric decomposition of the output data.
• Since SIN and COS waveforms have quadrant symmetry, the two most significant bits of the phase data are used to mirror the single quadrant data around the X and Y axis. For the SIN function the amplitude of the wave in the π to 2 π interval is just the negative of the amplitude in the 0 to or π interval. For the COS function the amplitude of the wave in the π/2 to 3 π/2 interval is just the negative of the amplitude in the 3 π/2 to π/2 interval. The two MSBs of the phase accumulator specify the quadrant (00→1, 01→2, 10→3, 11→4). For the SIN function, the MSB of the phase data is used to negate the positive data generated for the first two quadrants. For the COS function, an XOR of the two phase data MSBs is used to negate the positive data generated for quadrants 1 and 4.[0106]
• The above technique reduces memory requirements by a factor of 4. This still results in a memory requirement of 4K words×12 bits To reduce the table sizes further, a trigonometric decomposition is performed on the angles. The following trigonometric identity is used:[0107]
• sin Θ.=sin (Φ₁+Φ)=sin Φ₁cos Φ₂+sin Φ₁cos Φ₁  Equation 4
• Letting Φ[0108]₂<<._Φ1leads to the complete approximation as follows:
• sin θ=sin Φ₁+sin Φ₂cos Φ₁  Equation 5
• It is not necessary to use all bits of Φ[0109]₁when computing the second term of the equation so Φ₁is a subset of Φ₁. To generate the COS function, the same approximation may be used since
• cos θ=sin (θ+π/2)  Equation 6
• This results in a modification of the Φ[0110]₁& {circumflex over (Φ)}₁variables when computing the COS function. The data stored in the COS ROMs will incorporate this angle modification so no changes to the phase data are required.
• The[0111]modulator68 mixes the interpolated I and Q samples on lines80 and81 with the digital intermediate frequency signal represented by the complex SIN and COS function data onlines84 and85 to produce a modulated digital intermediate frequency signal online87.
• The interpolated I,Q samples and DDS output are digitally mixed by two 10×12 multipliers. The outputs of the mixing process are then summed by a 12 bit adder to form a modulated carrier. It is possible to alter the operation of the[0112]modulator68 by forcing the I input to all zeroes and the Q input to all ones. The effect of this is that one multiplier will output all zeroes and the other will output the signal from the DDS SIN,COS generator67 only. The sum of these two signals yields an unmodulated digital intermediate frequency signal.
• The[0113]modulator68 creates a modulated digital intermediate frequency signal online87 according to the following equation:
• f(t)=I·COS(Φ(t))+Q·SIN(Φ(t))  Equation 7
• The 12-bit output of the DDS SIN and[0114]COS generator67 is multiplied by the 10 bit interpolated I and Q samples from theinterpolator63 to generate two 12-bit products. The two products are then added (combined) to generated a 12-bit modulated output online87.
• Since both the I multiplier and the Q multiplier generate 12-bit products, it is possible that an overflow could occur when their outputs are combined. Therefore it is necessary to ensure that the magnitude of the vector generated by I and Q never exceeds[0115]1 (assuming |I|, |Q| are fractional numbers ≦1). If this is not ensured then an overflow of the modulator adder is possible.
• The[0116]noise shaper69 provides a filtered modulated or unmodulated digital intermediate frequency signal online71 to theDAC18. Thenoise shaper69 is designed to decrease the amount of noise power in the output spectrum caused by amplitude quantization error.
• The[0117]noise filter69 works on the fact that the quantization noise is a normal random process, and the power spectral density of the process is flat across the frequency band. The desired output signal is overlayed on top of this quantization noise floor. The noise shaping device is a simple multitap finite impulse response (FIR) filter. The filter creates a null which decreases the quantization noise power in a certain part of the frequency band. When the desired signal is overlayed on the filtered noise spectrum, the effective SQNR increases.
• The FIR filter transfer function is given by[0118]
• H(z)=1+bz⁻¹−z⁻²  Equation 8
• A two adder stage creates a second tap value of b in the range of +1.75 to −1.75 (in binary weights of 0, 0.25, 0.50, 1.0) that will move the zero of the filter across the output frequency band, so that it may be placed as near as possible to the desired output frequency for maximum SQNR performance.[0119]
• The null frequency can be computed by solving for the roots of the above equation in the z-plane. The roots are a complex conjugate pair that reside on the unit circle. The null frequency is given by the relation:[0120]$\begin{array}{cc}{f}_{\mathrm{null}}=\frac{\Theta }{360°}•f_{\mathrm{sampling}}& \mathrm{Equation}9\end{array}$

  
• where Θ is the angle of the root in the upper half plane. The conjugate root will provide a null reflected around the Nyquist frequency.[0121]
• Table 2 lists null frequencies generated by the binary weighted second tap. Let b[0122]3, b2, and b1 correspond to the weights 1.0 0.5 0.25, a “+” symbol means the tap is equal to its weight, a “−” symbol means that the tap is equal to the negative of its weight, and ‘0’ means that the tap has no weight. Some of the null frequencies are equal to those of other combinations, simply because the possible combinations sometimes overlap (e.g. 1.0+0.5-0.25=1.0+0.0+0.25). f_sampleis 1.00.

  TABLE 2
  b3	b2	b1	f(null)	f(alias)
  0	0	0	0.250	0.750
  0	0	−	0.269	0.731
  0	0	+	0.230	0.770
  0	+	0	0.210	0.790
  0	+	+	0.188	0.812
  0	+	−	0.230	0.770
  0	−	0	0.290	0.710
  0	−	+	0.269	0.731
  0	−	−	0.312	0.688
  +	0	0	0.167	0.833
  +	0	−	0.188	0.812
  +	0	+	0.143	0.857
  +	+	0	0.115	0.885
  +	+	+	0.080	0.420
  +	+	−	0.143	0.857
  +	−	0	0.210	0.790
  +	−	+	0.188	0.812
  +	−	−	0.230	0.770
  −	0	0	0.333	0.667
  −	0	−	0.357	0.643
  −	0	+	0.312	0.688
  −	+	0	0.290	0.710
  −	+	+	0.269	0.731
  −	+	−	0.312	0.688
  −	−	0	0.385	0.615
  −	−	+	0.357	0.643
  −	−	−	0.420	0.580

• All timing is derived from the 21.76 MHz clock signal on[0123]line71a.
• The functions associated with the signals in the control registers[0124]64 are now described.
• When the CW MODE signal is set, the I input to the respective multiplier in the[0125]modulator68 is forced to all zeroes, and the corresponding Q input forced is to all ones. The net effect is that an unmodulated carrier will be generated. This function is double buffered and the loaded data will not become active until a TUNE command is issued.
• The INTERPOLATOR ENABLE signal enables the ×16 interpolator on the I,Q samples. If the INTERPOLATOR ENABLE signal is not set then the I,Q data is input directly to the multiplier.[0126]
• External memory required for the operation of the[0127]processor chip12 is provided by afast memory13 and aslow memory14. Thefast memory13 is accessed by anaddress decoder15. Thefast memory13 is a cache memory implemented in a RAM having zero wait states. Theslow memory14 is a bulk memory that is implemented in an EPROM, having two wait states. Theslow memory14 is coupled to theprocessor chip12 for storing processing codes used by theprocessor chip12 when said codes need not be operated with zero wait states; and the fast memory is coupled to theprocessor chip12 for temporarily storing processing codes used by theprocessor chip12 when said codes are operated with zero wait states. When procedures must be run with zero wait states, the code can be uploaded from theslow memory14 to thefast memory15 and run from there. Such procedures include the interrupt service routines, symbol demodulation, RCC acquisition, BPSK demodulation, and voice and data processing.
• The[0128]processor chip12 includes a single model TMS320C25 digital signal processor, which performs four main tasks, a subscriber control task (SCT)91, channel control task (CCT)92, a signal processing task (SPT)93, and a modem processing task (MPT)94, as shown in FIG. 4. These four tasks are controlled by asupervisor module95. The SCT deals with the telephone interface and the high-level call processing. The CCT controls the modem and RELP operation and timing, and performs power-level and TX timing adjustments according to requests from the base station. The SPT performs the RELP, echo cancellation and tone generation functions. The supervisor calls these four tasks sequentially and communicates with them via control words.
• The[0129]SCT91 provides the high level control function within the subscriber unit and has three fundamental modes of operation: idle, voice and abort.
• The SCT enters Idle Mode after power up and remains in that state until an actual voice connection is made. While in the Idle Mode, the SCT monitors the subscriber telephone interface for activity and responds to base station requests received over the radio Control channel (RCC).[0130]
• The primary function of the SCT is to lead the Subscriber Unit through the setup and teardown of voice connections on a radio channel. Before the unit can set up any kind of call, however, it must find the correct base station. The SCT determines which RCC frequency to use, and sends the frequency information to the CCT. A description of the initialization of a communication channel between the subscriber unit and the base station is contained in U.S. patent application Ser. No. 07/070,970 filed Jul. 8, 1987 now U.S. Pat. No. 4,811,420.[0131]
• Once the subscriber unit has gained RCC sync, it can set up a call by exchanging messages over the RCC with the base station, and by monitoring and setting hardware signals on the telephone interface. The following walk through briefly describe the events that take place during call setup.[0132]
• Normal call setup for call origination begins with the subscriber taking the handset off hook to initiate a service request. The SCT sends a CALL REQUEST message to the base station. The SCT receives a CALL CONNECT message. The SCT signals the CCT to attempt sync on the voice channel assigned via the CALL CONNECT message. The CCT attains sync on the voice channel. The subscriber receives a dial tone from the central office. Call setup is complete. The central office provides the remaining call termination support.[0133]
• Normal call setup for call termination takes place as follows: The SCT receives a PAGE message from the base station. The SCT replies with a CALL ACCEPT. The SCT receives a CALL CONNECT message. The SCT signals the CCT to attempt sync on the voice channel assigned via the CALL CONNECT message. The CCT attains sync on the voice channel. The SCT starts the Ring Generator to apply ring to the local loop. The subscriber takes the hand set off hook. The ringing is stopped. The voice connection is complete.[0134]
• The SCT implements the call setup and teardown operations as a finite state machine.[0135]
• If a voice channel seizure is successfully completed, the SCT switches to the voice mode and performs a very limited set of support functions. SCT processor loading is kept to a minimum at this time to give the RELP speech compression, echo cancellation and modem processing algorithms maximum processor availability.[0136]
• The SCT enters the abort mode as a result of an unsuccessful call origination attempt or an unexpected call teardown sequence. During the abort mode, a reorder is sent to the handset. The SCT monitors the subscriber telephone interface for a disconnect (extended on-hook), at which time the subscriber unit enters the idle Mode. Base station requests received over the radio control channel (RCC) are rejected until the disconnect is detected.[0137]
• The[0138]CCT92 operates as a link level channel controller in the baseband software. The CCT has three fundamental states: RCC operation, refinement, and voice operation.
• At power up, the CCT enters the RCC operation state to search for and then support the RCC channel. The RCC operation includes the following functions: AM hole control; monitoring sync and modem task status; radio channel timing adjustment; RX RCC message filtering; TX RCC message formatting; monitoring the PCM buffer I/O; and link information processing.[0139]
• After a voice connection is established, the CCT enters the refinement state to fine tune the modem's fractional timing. Refinement includes the following functions; interpreting and responding to refinement bursts; creating and formatting TX refinement bursts; forwarding messages to the SCT as appropriate; monitoring the modem status; and monitoring the PCM buffer I/O.[0140]
• Following Refinement, the CCT begins voice operation, which includes the following functions: code word signalling support; dropout recovery; monitoring sync and modem status; and monitoring the PCM buffer I/O.[0141]
• The[0142]CCT92 has three fundamental states of operation: idle, refinement and voice. The following is a walk through of the state transitions involved in CCT operation.
• After a reset the CCT enters the idle state and remains inactive until given channel assignment instructions by the SCT. The SCT provides the CCT with a frequency upon which to search for the radio control channel (RCC). The CCT then instructs the MPT to synchronize the receiver to the given frequency and to search for an AM hole. Failure to detect an AM hole within a predetermined time period causes the CCT to request another frequency upon which to search from the SCT. This continues indefinitely until the AM hole detection is successful.[0143]
• Following a successful AM hole detection, the CCT begins to check received data for the unique word. A small window around the nominal unique word position is scanned since the AM hole detection process may be off by a few symbol times. Once the unique word is located and the CRC error detection word is verified correct, the exact receive symbol timing can be determined. The TDM framing markers are then adjusted to the correct alignment and normal RCC support begins. If the unique word cannot be located, the AM hole detection is considered false and the CCT requests a new frequency assignment from the SCT.[0144]
• During RCC operation the CCT filters received RCC messages. The majority of the base station's RCC messages are null patterns and these are discarded after link information is read from the link byte. RCC messages that contain real information are forwarded to the SCT for processing. If RCC synchronization is lost, the CCT again requests a new frequency from the SCT. The SCT will respond with the correct frequency according to the RCC frequency search algorithm.[0145]
• When the SCT initiates a voice call, the CCT is assigned a voice channel and time slot. The CCT brings the subscriber unit active according to this assignment and begins the refinement process. During refinement, the base and subscriber units transmit a BPSK signal specifically designed to assist the modem in fractional bit time acquisition. The base station CCU relays the bit timing offset back to the subscriber unit as a two's complement adjustment value. The CCT maintains a time average of these fedback offsets. Once the CCT determines that the fractional timing value is within a required tolerance, it adjusts the subscriber unit's transmit timing accordingly. The length of the time average is determined dynamically, depending upon the variance of the fractional time samples. After a timing adjustment, the time average is reset and the procedure is repeated.[0146]
• Once the base station detects that the subscriber unit is within an acceptable timing tolerance, it terminates the refinement process and voice operation begins. The length of the refinement process is determined dynamically, depending upon the success of the subscriber unit's timing adjustments. Power and integer symbol timing are also monitored and adjusted as necessary during the refinement process. If the subscriber fails to find the base station's refinement bursts after a period of time, or if the refinement process cannot yield acceptable timing, the connection is broken and the CCT returns to RCC operation.[0147]
• Following successful refinement, the CCT enters voice operation at the assigned modulation level. The voice operation tasks include controlling RELP and MPT operations, establishing voice synchronization and continuously monitoring the voice code words sent from the base station. Local loop control changes, signalled via the code words, are reported to the SCT as they occur. Power and fractional timing incremental changes are also determined from the code words. Transmitted voice code words are formulated by the CCT based upon the local loop control provided by the SCT and the channel link quality reported by the modem. The CCT returns to the RCC when the SCT executes a call teardown sequence.[0148]
• If voice synchronization is lost, the CCT initiates a fade recovery operation. After ten seconds of failure to reestablish a good voice connection, the CCT informs the SCT of the condition, initiating a call teardown. This returns the CCT to the Idle state.[0149]
• During a channel test operation, a voice burst is replaced with channel test data. When a burst has just been received, it is analyzed for bit errors. The bit error count is passed to the base station via the reverse channel bursts.[0150]
• The[0151]SPT93 performs all of the digital signal processing (DSP) tasks within the subscriber unit. The various DSP functions are invoked as required, under the control of thesupervisor module95.
• The SPT includes a RELP module. which is executed from a high speed RAM. The RELP module performs RELP Speech compression and expansion with echo cancellation. The RELP module transforms 180 byte blocks of 64 Kbps PCM voice data to and from 42 bytes of compressed voice data using the RELP algorithm.[0152]
• The SPT also includes a signal processing control (SPC) module, which determines if tone generation or RELP should be invoked. If RELP, SPC determines whether to call the synthesis or analysis routines. The synthesis routine returns a parity error count, which is handled by the SPTCTL routine. If tone generation is required, it determines whether to output silence or reorder.[0153]
• The SPT is controlled via commands from the SCT and the CCT. These commands invoke and control the operation of the various functions within the SPT as they are required by the subscriber unit. RELP and echo cancellation software, for example, are only executed when the subscriber unit is active on a voice call. Call progress tones are generated anytime the subscriber unit's receiver is off hook and RELP is not active. The tones include silence and reorder. Except for the IDLE mode, the interrupt service routine handling the PCM codec operates continuously as a foreground process, filling the circular PCM buffer.[0154]
• The control and modem functions are performed in between the analysis and synthesis processing.[0155]
• The[0156]MPT94 demodulation procedure is divided into two procedures: DEMODA & DEMODB, thus allowing the RELP synthesis to be executed on the RX data in buffer A right after the DEMODA procedure is completed. After DEMODA all internal RAM variables should be stored in external RAM, then reloaded to internal RAM before performing DEMODB. This is because RELP uses the internal RAM.
• When the RXCLK interrupt on[0157]line26cis received by theprocessor chip12, the MPT causes four received RX data samples to be read and then placed in a circular buffer, for processing by the demodulation procedure. This allows other tasks to be performed while receiving RX samples.
• The MPT receives the RXCLK interrupt signal on[0158]line26cfrom theFIR chip16 every 62.5 .mu.sec during the receive slot. The RXCLK interrupt signal is masked by the processor chip firmware during idle or transmit slots.
• The MPT receives the TXCLK interrupt signal on[0159]line26cfrom theFIR chip16 only during the transmit slot. The TXCLK interrupt signal tells theprocessor chip12 when to send a new TX symbol to the FIR chip.
• The MPT reads four samples from the[0160]RX sample buffer35 in theFIR chip16 during each RXCLK interrupt online26c. The MPT resets the input and output address counters to the buffer at the start of the receive slot.
• The MPT sends TX symbols to the[0161]TX symbol buffer36 in theFIR chip16.
• The MPT provides the data to the fractional timing circuit in the[0162]RX timing module39 in theFIR chip16 that is used to align the RXCLK interrupt signal online26cwith the base station transmission.
• The MPT also synchronizes the DDS frequency to the base station transmit frequency.[0163]
• Referring to FIG. 5 the MPT includes the following modules: a[0164]supervisor module101, atraining module102, afrequency acquisition module103, abit synchronization module104, avoice demodulation module105, a symbol receivemodule106, and a transmitmodule107.
• The[0165]supervisor module101 is the MPT task supervisor. It reads the MPT control word (CTRL0) from the RAM, and calls other routines according to the control word.
• The[0166]training module102 computes a vector of 28 complex FIR filter coefficients. It is activated in the idle mode after power up and about every three hours. A training transmitter implemented by the MPT is activated in a loopback mode to send a certain sequence of symbols. This sequence is looped back to a training receiver implemented by the MPT, in a normal mode, in advanced and delayed timing modes, and in upper and lower adjacent channels.
• The training receiver uses the samples of the input waveform to create a positive definite symmetric matrix A of order[0167]28. Also a 28-word vector V is created from the input samples. The coefficients vector C is given by:
• C=A⁻¹V  {Eq. 10}
• The B coefficient is then calculated according to the algorithm: B=A[0168]⁻¹given A.
• The training transmitter is activated in the loopback mode to transmit five similar pairs of sequences. Each pair consists of the following two sequences:[0169]
• I sequence: 9 null symbols, “i”, 22 null symbols[0170]
• Q sequence: 9 null symbols, “j”, 22 null symbols[0171]
• The “i” can be any symbol. The “j” is a symbol that differs from “i” by 90 degrees.[0172]
• The receiver processing tasks are:[0173]
• Adjust the AGC so that the signal peak in the normal mode is 50 to 70% of the maximum. The AGC is increased by 23 db for the 4th and 5th modes.[0174]
• Read and store the input samples. The first 32 samples are discarded and the next 64 samples are stored, for each sequence.[0175]
• Build the matrix A([0176]28,28). The following process is done in the normal mode:
• A(I,J)=A(I,J)+ΣX(4N−I)·X(4N−J)  Equation 11
• The addition is for all N that satisfy:[0177]
• 0<=4N−I<64 & 0<=4N−J<64  Equation 12
• For the advanced and delayed sequences, the same process is performed except that the term resulting from N=8 is not added. In the upper and lower adjacent channel channel sequences, the following process is performed:[0178]
• A(I,J)=A(I,J)+ΣX(2N−I)·X(2N−J)  Equation 13
• The addition is for all N that satisfy:[0179]
• 0<=2N−I<64 & 0<=2N−J<64  Equation 14
• Create the vector V(1:28) from the samples of the first pair of sequences:[0180]
• Re{V(I)}=X(32−I); where X are samples of the first (I) sequence.[0181]
• Im{V(I)}=X(32−I); where X are samples of the second (Q) sequence.[0182]
• Find the coefficients vector C by solving the equation:[0183]
• A×C−V=0  Equation 15
• These processing steps are more fully described in U.S. Pat. No. 4,644,561 issued Feb. 17, 1987 to Eric Paneth, David N. Critchlow and Moshe Yehushua.[0184]
• The[0185]frequency acquisition module103 is run when receiving the control channel, in order to synchronize the subscriber unit RX frequency to the base station transmit frequency. This is done by adjusting the DDS CW output until the energies of the received signal's two sidebands are equal. Afterwards, the DDS TX frequencies are adjusted according to the computed frequency deviation.
• If the procedure fails to achieve frequency sync, an appropriate error code is placed in the status word.[0186]
• The[0187]bit synchronization module104 is run when receiving the RCC and after completing the frequency acquisition. A certain pattern is transmitted in the first 44 symbols in the RCC transmission from the base station, and this is used by this module to compute the RXCLK deviation from the correct sampling time. This deviation is used to adjust the RXCLK timing.
• The[0188]voice demodulation module105 is activated to demodulate a voice slot. It is resident in the slow EPROM and its functions are divided between two procedures DEMODA and DEMODB.
• The DEMODA functions include initializing parameters for the symbol receive[0189]module106; calling the symbol receive module to process the received symbols for buffer A; and storing the variables in external RAM before exiting.
• The DEMOOB functions include loading the variables from external RAM to internal RAM; calling the symbol receive module to process the received symbols for buffer B; and determining link quality and other information after receiving all the symbols in the slot.[0190]
• The symbol receive[0191]module106 is uploaded to the RAM when the CCT goes to the voice mode. It is called by DEMODA or DEMODB to perform the following: (1) read I and Q samples from the circular buffer; (2) FIR filtering of the I&Q samples; (3) determine the transmitted symbols and and put them in a buffer; (4) execute a phase-lock-loop to synchronize the DDS to the incoming signal; (5) execute the bit tracking algorithm; (6) AGC calculation; and (7) accumulate data for link quality calculation.
• The transmit[0192]module107 includes the interrupt service routine for the TXCLK interrupt signal received online26cfrom theFIR chip16, which occurs once per two symbols during a transmit slot. The functions of the transmitmodule107 include: (1) unpacking the transmit symbol from the RELP buffer; (2) performing an inverse GRAY coding on it; (3) adding it to the previous transmitted phase (because of the DPSK transmission); and (4) sending it to the TX buffer in theFIR chip16.
• The interface of the MPT to the baseband tasks is accomplished via control and status words and data buffers in the shared memory. Procedures requiring fast execution are uploaded into the cache memory when needed. These include the interrupt service routines. symbol demodulation, RCC acquisition; and BPSK demodulation.[0193]
• The MPT supervisor will not wait for RXSOS to read and decode the control word, but will do that immediately when it is called.[0194]
• The TMS320C25 goes to a powerdown mode when executing the IDLE instruction. In order to conserve power the firmware will be in the idle mode most of the time, when there is no phone call in progress. So after a reset the supervisor will acquire RCC sync then go to idle mode until a predetermined interrupt causes a corresponding service routine to be executed. When operated in the powerdown mode, the TMS320C25 enters a dormant state and requires only a fracion of the power normally needed to supply the device. While in powerdown mode, all of the internal contents of the processor are maintained to allow operation to continue unaltered when the powerdown mode is terminated. Upon receipt of an interrupt the[0195]processor chip12 terminates the powerdown mode temporally and resumes normal operation for a minimum time of one main loop cycle. The requirements of the powerdown mode are checked at end of main loop every time to determine whether or not the subscriber unit to return to the powerdown mode.
• The slot clock is based on the hardware generated slot timing. When a slot marker triggers an interrupt, the routine increments the clock by one tick. Each clock tick represents 11.25 ms in time.[0196]
• The receive and transmit functions of the UART are not interrupt driven, but are controlled by the background software (this controls processor loading and prevents runaway interrupt conditions). The. processing code supports the XON/XOFF protocol by intercepting these characters directly and immediately enabling or disabling UART transmission as appropriate. The rate of the receive and transmit operation is designed to be selective by an external DIP switch device. The typical data reception rate is at 9600 baud. A circular buffer is used to control the UART's transmission. The background software periodically checks the queue and initiates transmission if it is not empty. It does this by sending bytes to the UART one byte at a time until the queue is empty.[0197]
• The switch hook is sampled with the TMS320C25 internal timer interrupt routine. To simulate DC signalling, a 1.5 ms sample period is used. This interrupt is aligned to frame timing at the beginning of each frame therefore its frequency is phase locked to the base station to prevent underrun or overflow of the switch hook buffer. For each interrupt, a bit representing the switch hook detect signal (from the SLIC) is entered in the 60-bit Switch Hook Sample buffer (SSB). The SSB is examined by the SCT once every 45 ms during normal operation. This interrupt is enabled by the software at all times.[0198]

Claims (21)

What is claimed is:

1. A system for processing communication signals which include a first information signal and a first communication signal which carries information contained in digital input symbols, and a second communication signal provides a second information signal, the communication signals provided over a plurality of radio frequency channels within a selected band of radio frequencies, the system comprising:

a circuit for generating a base radio frequency signal;

a circuit for generating a digital intermediate frequency signal such that the combination of the digital intermediate frequency signal with the base radio frequency signal produces a communication signal having a frequency within one of said radio frequency channels selected for communication;

a circuit for modulating the digital intermediate frequency signal with the digital input symbols to produce a modulated digital intermediate frequency signal, and combining the modulated digital intermediate frequency signal with the base radio frequency signal to produce the first communication signal; and

a circuit for combining the modulated digital intermediate frequency signal with the base radio frequency signal to produce the first communication signal.

2. The system ofclaim 1 further comprising a circuit for accumulating phase increment data to produce digitized phase values and generating the digital intermediate frequency signal based on the digitized phase values.

3. The system ofclaim 1 further comprising a demodulation circuit, the demodulation circuit using the digital intermediate frequency signal to demodulate the second communication signal.

4. The system ofclaim 3 further comprising a filter, including a noise shaping circuit, for filtering the digital intermediate frequency prior to demodulating the second communication signal received.

5. The system ofclaim 1 further comprising:

a memory; and

the circuit for generating the digital intermediate frequency signal based on the digitized phase values by using predefined values stored in the memory.

6. The system ofclaim 1 comprising:

a lookup table comprising a memory having a set of predefined stored values pertaining to the amplitude of a signal for a single quadrant;

a signal generator receiving an input signal and generating sine and cosine waveforms utilizing amplitude values obtained from said lookup tables, wherein the input signal includes phase data and specifies the quadrant and the algebraic sign of the phase data;

said signal generator accessing said lookup table differently depending upon the quadrant and sign of the phase data such that the lookup table provides an amplitude value from said set of values based upon the phase data; and

a circuit combining the sine and cosine waveforms to produce the digital frequency.

7. The system ofclaim 6 wherein the lookup table includes a first table having stored values pertaining to large angle approximations for coarse resolution frequency adjustment of the digital frequency and a second table for stored values pertaining to small angle approximations for fine resolution frequency adjustment.

8. The system ofclaim 6 wherein trigonometric decomposition is utilized to further reduce table size.

9. The system ofclaim 1 further comprising the circuit for modulating the digital intermediate frequency signal transcoding the first information signal into digital input symbols and modulating the digital intermediate frequency signal with the digital input symbols to produce the modulated digital intermediate frequency signal.

10. The system ofclaim 1 further comprising the circuit for generating the digital intermediate frequency signal using predefined sine and cosine waveform values stored in the memory to provide digitized phase values.

11. The system ofclaim 10 further comprising the circuit for generating the digital intermediate frequency using coarse and fine resolution frequency approximations stored in the memory.

12. The system ofclaim 11 further comprising the memory providing two lookup tables.

13. The system ofclaim 12 further comprising the circuit for generating the digital intermediate frequency signal reducing the table sizes by utilizing quadrant symmetry of the sine and cosine waveform values.

14. The system ofclaim 13 further comprising the circuit for generating the digital intermediate frequency signal utilizing trigonometric decomposition, thereby reducing required table sizes.

15. The system ofclaim 1 comprising:

a lookup table comprising a memory having a set of predefined stored values pertaining to the amplitude of a signal for a single quadrant;

a signal generator receiving an input signal and generating sine and cosine waveforms utilizing amplitude values obtained from said lookup tables, wherein the input signal includes phase data and specifies the quadrant and the algebraic sign of the phase data;

the lookup table providing an amplitude value from said set of values based upon the phase data; and

a circuit combining the sine and cosine waveforms to produce the digital frequency.

16. The system ofclaim 15 comprising:

the lookup table including a first table having stored values pertaining to large angle approximations for coarse resolution frequency adjustment of the digital frequency and a second table for stored values pertaining to small angle approximations for fine resolution frequency adjustment; and

the circuit for generating the digital intermediate frequency signal utilizing trigonometric decomposition, thereby reducing required table sizes.

17. The system for processing communication signals which include a first information signal and a first communication signal which carries information contained in digital input symbols, and a second communication signal provides a second information signal, the communication signals provided over a plurality of radio frequency channels within a selected band of radio frequencies, the system comprising:

means for generating a base radio frequency signal;

means for generating a digital intermediate frequency signal such that the combination of the digital intermediate frequency signal with the base radio frequency signal produces a communication signal having a frequency within one of said radio frequency channels selected for communication;

means for modulating the digital intermediate frequency signal with the digital input symbols to produce a modulated digital intermediate frequency signal; and

circuit means for combining the modulated digital intermediate frequency signal with the base radio frequency signal to produce the first communication signal.

18. The system ofclaim 17 further comprising:

demodulation means, responsive to the digital intermediate frequency signal to demodulate the second communication signal; and

means for filtering the digital intermediate frequency by noise shaping prior to demodulating the second communication signal.

19. The system ofclaim 17 further comprising:

the means for generating a digital intermediate frequency signal generating the digital intermediate frequency signal based on predefined values stored in a memory store;

demodulation means, responsive to the digital intermediate frequency signal to demodulate the second communication signal;

the means for generating the digital intermediate frequency signal generating the signal based on the digitized phase values by using by utilizing quadrant symmetry of sine and cosine waveform values stored in the memory store;

the means for generating the digital intermediate frequency using coarse and fine resolution frequency approximations stored in the memory store.

20. The system ofclaim 17 comprising:

a lookup table comprising a memory having a set of predefined stored values pertaining to the amplitude of a signal for a single quadrant, the lookup table including a first table having stored values pertaining to large angle approximations for coarse resolution frequency adjustment of the digital frequency and a second table for stored values pertaining to small angle approximations for fine resolution frequency adjustment;

means for generating sine and cosine waveforms utilizing amplitude values and phase data obtained from said lookup tables; and

means for combining the sine and cosine waveforms to produce the digital frequency.

21. The system ofclaim 20 wherein trigonometric decomposition is utilized, thereby reducing table size.

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