本發明是有關於一種轉換器，特別是指一種低電壓應力直流轉換器。This invention relates to a converter, and more particularly to a low voltage stress DC converter.
在論文「B.R.Lin and H.K.Chiang,“Analysis and Implementation of a Soft Switching Interleaved Forward Converter with Current Double Rectifier,”IET Electr.Power Appl.,Vol.1,No.5,pp.697-704,2007.」提出一種習知的電源轉換器。In the paper "BRLin and HKChiang, "Analysis and Implementation of a Soft Switching Interleaved Forward Converter with Current Double Rectifier," IET Electr. Power Appl., Vol. 1, No. 5, pp. 697-704, 2007." A conventional power converter is proposed.
但是習知的電源轉換器的缺點為：However, the disadvantages of conventional power converters are:
1.所使用的開關應力是vin/1-D，其中vin為輸入電壓，D為功率開關導通比(duty ratio)，當D=0.5，開關應力為2vin，不適合高輸入電壓應用。1. The switching stress used is vin /1-D, where vin is the input voltage and D is the power switch duty ratio. When D = 0.5, the switching stress is 2vin , which is not suitable for high input voltage applications.
2.使用四個開關，增加硬體成本。2. Use four switches to increase hardware cost.
因此，本發明之目的，即在提供一種低電壓應力直流轉換器。Accordingly, it is an object of the present invention to provide a low voltage stress DC converter.
該低電壓應力直流轉換器，包含：第一及第二變壓器，每一變壓器具有一個一次側繞組和一個二次側繞組，且每一側繞組皆具有一第一端及一第二端，其中，該第二變壓器的一次側繞組的第一端電連接於該第一變壓器的一次側繞組的第二端，該第二變壓器的二次側繞組的第二端電連接於該第一變壓器的二次側繞組的第二端；一第一電容，具有一接收一呈直流的輸入電壓的第一端，及一第二端；一第二電容，具有一電連接於該第一電容之第二端的第一端，及一接收該輸入電壓的負極的第二端；一共振電感，電連接於該第一電容的第二端與該第一變壓器的一次側繞組的第二端之間；一第一開關，具有一電連接於該第一電容之第一端的第一端，及一電連接於該第一變壓器的一次側繞組的第一端的第二端，且該第一開關受控制以切換於導通狀態和不導通狀態間；一第二開關，具有一電連接於該第二變壓器的一次側繞組的第二端的第一端，及一電連接於該第二電容之第二端的第二端，且該第二開關受控制以切換於導通狀態和不導通狀態間；一初級側二極體，具有一電連接於該第二變壓器的一次側繞組的第二端的陽極及一電連接於該第一變壓器的一次側繞組的第一端的陰極；一第一二極體，具有一電連接於該第一變壓器的二次側繞組的第一端的陰極，及一陽極；一第二二極體，具有一電連接於該第二變壓器的二次側繞組的第一端的陰極，及一電連接於該第一二極體的陽極的陽極；一第一輸出電感，具有一電連接於該第一二極體之陰極的第一端，及一第二端；一第二輸出電感，具有一電連接於該第二二極體之陰極的第一端，及一第二端；及一輸出電容，電連接於該第一輸出電感的第二端與該第一二極體的陽極之間，用於提供一輸出電壓。The low voltage stress DC converter comprises: first and second transformers, each transformer having a primary side winding and a secondary side winding, and each side winding has a first end and a second end, wherein a first end of the primary winding of the second transformer is electrically connected to a second end of the primary winding of the first transformer, and a second end of the secondary winding of the second transformer is electrically connected to the first transformer Secondary windinga second end; a first capacitor having a first end receiving a DC input voltage and a second end; a second capacitor having a first electrical connection to the second end of the first capacitor And a second end of the negative electrode receiving the input voltage; a resonant inductor electrically connected between the second end of the first capacitor and the second end of the primary winding of the first transformer; a first switch a first end electrically connected to the first end of the first capacitor, and a second end electrically connected to the first end of the primary side winding of the first transformer, and the first switch is controlled to switch a second switch having a first end electrically connected to the second end of the primary winding of the second transformer, and a second electrically connected to the second end of the second capacitor And the second switch is controlled to switch between the conducting state and the non-conducting state; a primary side diode having an anode electrically connected to the second end of the primary winding of the second transformer and an electrical connection First end of the primary winding of the first transformer a cathode; a first diode having a cathode electrically connected to the first end of the secondary winding of the first transformer; and an anode; a second diode having an electrical connection to the second transformer a cathode of the first end of the secondary winding, and an anode electrically connected to the anode of the first diode; a first output inductor having an electrical connection to the cathode of the first diodea first end of the pole, and a second end; a second output inductor having a first end electrically connected to the cathode of the second diode, and a second end; and an output capacitor electrically connected The second end of the first output inductor is coupled to the anode of the first diode for providing an output voltage.
有關本發明之前述及其他技術內容、特點與功效，在以下配合參考圖式之一個較佳實施例的詳細說明中，將可清楚的呈現。The above and other technical contents, features and advantages of the present invention will be apparent from the following detailed description of the preferred embodiments.
如圖1所示，本發明低電壓應力直流轉換器之較佳實施例，包含：第一及第二變壓器T1、T2、第一及第二電容C1、C2、一共振電感Lr、第一及第二開關Q1、Q2、一初級側二極體Dr、第一及第二二極體D1、D2、第一及第二輸出電感L1、L2，及一輸出電容CO。As shown in FIG. 1, a preferred embodiment of the low voltage stress DC converter of the present invention comprises: first and second transformers T1 , T2 , first and second capacitors C1 , C2 , and a resonant inductor Lr , first and second switches Q1 , Q2 , a primary side diode Dr , first and second diodes D1 , D2 , first and second output inductors L1 , L2 , And an output capacitor CO .
每一變壓器T1、T2具有一個一次側繞組Lp1、Lp2和一個二次側繞組Ls1、Ls2，且每一側繞組Ls1、Ls2、Lp1、Lp2皆具有一第一端及一第二端，其中，該第二變壓器T2的一次側繞組Lp2的第一端電連接於該第一變壓器T1的一次側繞組Lp1的第二端，該第二變壓器T2的二次側繞組Ls2的第二端電連接於該第一變壓器T1的二次側繞組Ls1的第二端。該第一及第二變壓器T1、T2的匝數比相等，且每一個一次側繞組Lp1、Lp2的第一端是打點端，每一個一次側繞組Lp1、Lp2的第二端是非打點端。每一個二次側繞組Ls1、Ls2的第一端是打點端，每一個二次側繞組Ls1、Ls2的第二端是非打點端。Each of the transformers T1 and T2 has a primary side winding Lp1 , Lp2 and a secondary side winding Ls1 , Ls2 , and each of the side windings Ls1 , Ls2 , Lp1 , Lp2 has a first One end and a second end, wherein the first end of the primary side winding Lp2 of the second transformer T2 is electrically connected to the second end of the primary side winding Lp1 of the first transformer T1 , the second transformer The second end of the secondary winding Ls2 of T2 is electrically connected to the second end of the secondary winding Ls1 of the first transformer T1 . The first and second transformers T1, T2 ratio is equal to the number of turns, and each of the primary winding Lp1, Lp2 of the dot end is the first end, each of the primary winding Lp1, Lp2 of the second The end is the non-dip end. The first end of each of the secondary side windings Ls1 , Ls2 is a striking end, and the second end of each of the secondary side windings Ls1 , Ls2 is a non-tapping end.
第一電容C1具有一接收一呈直流的輸入電壓的正極的第一端，及一第二端。The first capacitor C1 has a first end that receives a positive input voltage and a second end.
第二電容C2具有一電連接於該第一電容C1之第二端的第一端，及一接收該輸入電壓vin的負極的第二端。The second capacitor C2 has a first end electrically connected to the second end of the first capacitor C1 and a second end receiving the input voltage vin the negative pole.
共振電感Lr電連接於該第一電容C1的第二端與該第一變壓器T1的一次側繞組Lp1的第二端之間。The resonant inductor Lr is electrically connected between the second end of the first capacitor C1 and the second end of the primary side winding Lp1 of the first transformer T1 .
第一開關Q1具有一電連接於該第一電容C1之第一端的第一端，及一電連接於該第一變壓器T1的一次側繞組Lp1的第一端的第二端，且該第一開關Q1受控制以切換於導通狀態和不導通狀態間。該第一開關Q1是一N型功率半導體電晶體，且該第一開關Q1的第一端是汲極，該第一開關Q1的第二端是源極。The first switch Q1 has a first end electrically connected to the first end of the first capacitor C1 and a second end electrically connected to the first end of the primary side winding Lp1 of the first transformer T1 And the first switch Q1 is controlled to switch between a conducting state and a non-conducting state. The first switch Q1 is an N-type power semiconductor transistor, and a first terminal of the first switch Q1 is a drain, the second terminal of the first switch Q1 is a source electrode.
第二開關Q2具有一電連接於該第二變壓器T2的一次側繞組Lp2的第二端的第一端，及一電連接於該第二電容C2之第二端的第二端，且該第二開關Q2受控制以切換於導通狀態和不導通狀態間。該第二開關Q2是一N型功率半導體電晶體，且該第二開關Q2的第一端是汲極，該第二開關Q2的第二端是源極。The second switch Q2 has a first end electrically connected to the second end of the primary side winding Lp2 of the second transformer T2 , and a second end electrically connected to the second end of the second capacitor C2 , and The second switch Q2 is controlled to switch between a conducting state and a non-conducting state. The second switch Q2 is an N-type power semiconductor transistor, and a first end of the second switch Q2 is a drain, the second terminal of the second switch Q2 is a source electrode.
初級側二極體Dr具有一電連接於該第二變壓器T2的一次側繞組Lp2的第二端的陽極及一電連接於該第一變壓器T1的一次側繞組Lp1的第一端的陰極。The primary side diode Dr has an anode electrically connected to the second end of the primary side winding Lp2 of the second transformer T2 and a first end electrically connected to the primary side winding Lp1 of the first transformer T1 Cathode.
第一二極體D1具有一電連接於該第一變壓器T1的二次側繞組Ls1的第一端的陰極，及一陽極。The first diode D1 has a cathode electrically connected to the first end of the secondary winding Ls1 of the first transformer T1 , and an anode.
第二二極體D2具有一電連接於該第二變壓器T2的二次側繞組Ls2的第一端的陰極，及一電連接於該第一二極體D1的陽極的陽極。The second diode D2 has a cathode electrically connected to the first end of the secondary winding Ls2 of the second transformer T2 , and an anode electrically connected to the anode of the first diode D1 .
第一輸出電感L1具有一電連接於該第一二極體D1之陰極的第一端，及一第二端。The first output inductor L1 has a first end electrically connected to the cathode of the first diode D1 and a second end.
第二輸出電感L2具有一電連接於該第二二極體D2之陰極的第一端，及一第二端。The second output inductor L2 has a first end electrically connected to the cathode of the second diode D2 and a second end.
輸出電容CO電連接於該第一輸出電感L1的第二端與該第一二極體D1的陽極之間，用於提供一輸出電壓VO。The output capacitor CO is electrically connected between the second end of the first output inductor L1 and the anode of the first diode D1 for providing an output voltage VO .
參閱圖2，為本實施例的操作時序圖，其中，參數vg1、vg2分別代表控制該第一及第二開關Q1、Q2是否導通的電壓，參數vCr1、vCr2分別代表該第一及第二開關Q1、Q2的寄生電容Cr1、Cr2的跨壓，參數iLm1、iLm2分別代表流經該二變壓器T1、T2的磁化電感Lm1、Lm2之電流，參數iLr代表流經該共振電感Lr之電流，參數iD1~iD2分別代表流過第一至第二二極體D1~D2的電流，參數iL1、iL2分別代表流過該第一輸出電感L1的電流、流過該第二輸出電感L2的電流，參數iO代表總輸出電流。依據該二開關Q1、Q2的切換，本實施例會在八種模式下操作，且在以下模式中會於圖示中畫出該二變壓器T1、T2的磁化電感Lm1、Lm2，且導通的元件以實線表示，不導通的元件以虛線表示，以下分別針對每一模式進行說明且令該二開關Q1、Q2的責任導通週期D<0.5。Referring to FIG. 2, it is an operation timing diagram of the embodiment, wherein the parameters vg1 and vg2 respectively represent voltages for controlling whether the first and second switches Q1 and Q2 are turned on, and the parameters vCr1 and vCr2 respectively represent the The voltages of the parasitic capacitances Cr1 and Cr2 of the first and second switches Q1 and Q2 , the parameters iLm1 and iLm2 respectively represent the magnetizing inductances Lm1 and Lm2 flowing through the two transformers T1 and T2 . Current, the parameter iLr represents the current flowing through the resonant inductor Lr , and the parameters iD1 ~iD2 represent the current flowing through the first to second diodes D1 -D2 , respectively, and the parameters iL1 and iL2 respectively represent output current of the first inductor of L1 flows, flows through the output inductor L current second parameter representative of the output current iO2. According to the switching of the two switches Q1 and Q2 , the present embodiment operates in eight modes, and in the following modes, the magnetizing inductances Lm1 and Lm2 of the two transformers T1 and T2 are drawn in the figure. The components that are turned on are indicated by solid lines, and the components that are not turned on are indicated by broken lines. Hereinafter, each mode is described for each mode and the duty-on period D<0.5 of the two switches Q1 and Q2 is made.
且以下分析，假設條件為：And the following analysis, the assumptions are:
1.第一及第二變壓器T1、T2的匝數比相等且磁化電感值相等(Lm1=Lm2=Lm)，且其漏電感相等。1. The first and second transformers T1 and T2 have equal turns ratios and equal magnetization inductance values (Lm1 = Lm2 = Lm ), and their leakage inductances are equal.
2.磁化電感Lm遠大於共振電感Lr及漏電感。2. The magnetizing inductanceLm is much larger than the resonant inductor Lr and the leakage inductance.
3.第一及第二電容C1、C2的電容值遠大於第一及第二開關Q1、Q2的寄生電容Cr1、Cr2。3. The capacitance values of the first and second capacitors C1 and C2 are much larger than the parasitic capacitances Cr1 and Cr2 of the first and second switches Q1 and Q2 .
4.第一及第二輸出電感L1、L2的電感值相等，即L1=L2。4. The inductance values of the first and second output inductors L1 and L2 are equal, that is,L1 =L2 .
5.輸出電容CO很大，輸出電壓vo可視為常數。5. The output capacitor CO is large, and the output voltagevo can be regarded as a constant.
6.操作在連續導通模式(CCM)。6. Operate in continuous conduction mode (CCM).
7.儲存於共振電感Lr及漏電感的能量大於寄生電容Cr1、Cr2的能量，以達成零電壓切換(Zero voltage switching，ZVS)操作。7. The energy stored in the resonant inductor Lr and the leakage inductance is greater than the energy of the parasitic capacitancesCr 1 ,Cr 2 to achieve a zero voltage switching (ZVS) operation.
模式一(時間：tMode one (time: t00~t~t11)：):
參閱圖2及圖3a，第一開關Q1導通，而第二開關Q2不導通。Referring to Figures 2 and 3a, the first switch Q1 is turned on and the second switch Q2 is turned off.
第一開關Q1處於導通狀態，使儲存於第一電容C1的能量藉由第一變壓器T1傳遞至負載，其詳細操作為：第一開關Q1跨壓vCr1=0，第一變壓器T1的一次側電壓vP1vC1>0，磁化電感電流iLm1線性上升，變壓器T1的二次側電壓，且第二開關Q2跨壓vCr2=vin，第二變壓器T2經由初級側二極體Dr作去磁重置，所以vP2-vP1<0，而使第二二極體D2導通且第一二極體D1為不導通，第一輸出電感電壓vL1=2nvC1-vo>0，其電流iL1線性上升，此時儲存於第一電容C1的能量藉由第一變壓器T1傳遞至負載。且第二輸出電感電壓vL2=-vo<0，其電流iL2線性下降，因此總輸出電流i0=iL1+iL2會有漣波相消的效果The first switch Q1 is in an on state, so that the energy stored in the first capacitor C1 is transmitted to the load through the first transformer T1 , and the detailed operation is as follows: the first switch Q1 crosses the voltage vCr1 =0, the first transformer Primary side voltage of V1 vP1 vC1 >0, the magnetizing inductor current iLm1 rises linearly, and the secondary side voltage of the transformerT1 And the second switch Q2 crosses the voltage vCr2 =vin , and the second transformer T2 is demagnetized reset via the primary side diode Dr , so vP2 -vP1 <0, and the second diode D2 is turned on and the first diode D1 is non-conducting, the first output inductor voltage vL1 =2nvC1 -vo >0, and the currentiL 1 is linear Ascending, the energy stored in the first capacitor C1 is transferred to the load by the first transformer T1 . And the second output inductor voltage vL2 = -vo <0, the current iL2 linearly decreases, so the total output current i0 = iL1 + iL2 will have the effect of chopping cancellation
模式二(時間：tMode two (time: t11~t~t22)：):
參閱圖2及圖3b，第一及第二開關Q1、Q2皆不導通。Referring to FIG. 2 and FIG. 3b, the first and second switches Q1 and Q2 are not turned on.
流經第一開關Q1的電流iQ1為正值且對其寄生電容Cr1充電，而使寄生電容Cr1的電壓vCr1上升，初級側二極體Dr導通，而使二寄生電容Cr1、Cr2的電壓vCr1和vCr2滿足vCr1+vCr2=vin，所以寄生電容Cr2放電，其電壓vCr2下降。由於二寄生電容Cr1和Cr2非常小，因此vCr1上升和vCr2下降非常快，因此模式二歷時很短，磁化電感電流iLm可視為常數，同時iQ1=niL1，因此寄生電容Cr1受電流iQ1快速充電。Current flowing through the first switch Q1 iQ1 and its positive charge parasitic capacitance CR1, the stray capacitance of the voltage v Cr1Cr1 rises, the primary-side diode Dr is turned on, the two parasitic capacitance C The voltages vCr1 and vCr2 ofr1 and Cr2 satisfy vCr1 +vCr2 =vin , so the parasitic capacitance Cr2 is discharged, and the voltagevCr 2 thereof is lowered. Since the two parasitic capacitances Cr1 and Cr2 are very small, vCr1 rises and vCr2 drops very fast, so mode 2 lasts for a short time, and magnetizing inductor current iLm can be regarded as a constant At the same time, iQ1 =niL1 , so the parasitic capacitance Cr1 is quickly charged by the current iQ1 .
當t=t2時，第一開關跨壓vCr1上升至vC1時，此時vCr2也下降至vC2，則第一變壓器T1的一次側電壓vP1=0，第二變壓器T2的一次側電壓vP2=0，因此vS1=0，而且vS2=0，使第一二極體D1開始導通，進入電流換向(commutation)，而進入模式三。Whent =t2 , when the first switch voltage vCr1 rises to vC1 , at this time vCr2 also drops to vC2 , the primary side voltage vP1 =0 of the first transformer T1 , the second transformer T2 The primary side voltage vP2 =0, so vS1 =0, and vS2 =0, causes the first diode D1 to start conducting, enters current commutation, and enters mode three.
模式三(時間：tMode three (time: t22~t~t33)：):
參閱圖2及圖3c，第一及第二開關Q1、Q2皆不導通。Referring to FIG. 2 and FIG. 3c, the first and second switches Q1 and Q2 are not turned on.
第一及第二開關Q1、Q2的跨壓vCr1=vC1、vCr2=vC2，第一及第二變壓器T1和T2的一次側電壓箝位在零，iLm1和iLm2保持常數，第一及第二變壓器T1和T2的二次側電壓vS1=vS2=0，進行換向過程。The voltage across the first and second switches Q1 , Q2 is vCr1 = vC1 , vCr2 = vC2 , and the primary side voltages of the first and second transformers T1 and T2 are clamped at zero, iLm1 and iLm2 is kept constant, and the secondary side voltages VS1 =vS2 =0 of the first and second transformers T1 and T2 are subjected to a commutation process.
共振電感Lr、寄生電容Cr1和Cr2形成共振電路，第一開關Q1跨壓vCr1持續上升，第二開關Q2跨壓vCr2持續下降，共振電感Lr跨負電壓，其電流iLr下降，而使第二二極體電流iD2遞減，第一二極體電流iD1遞增。The resonant inductor Lr and the parasitic capacitances Cr1 and Cr2 form a resonant circuit. The first switch Q1 continues to rise across the voltage vCr1 , the second switch Q2 continues to fall across the voltage vCr2 , and the resonant inductor Lr crosses the negative voltage. iLr drops, causing the second diode current iD2 to decrease, and the first diode current iD1 to increase.
模式三的共振電感Lr的初始儲能必須大於寄生電容Cr1和Cr2的初始儲能，方能使第二開關Q2跨壓vCr2下降至零，達到零電壓切換(ZVS)的條件。The initial energy storage of the resonant inductor Lr of mode 3 must be greater than the initial energy storage of the parasitic capacitances Cr1 and Cr2 to enable the second switch Q2 to fall to zero across the voltage vCr2 to achieve zero voltage switching (ZVS) conditions. .
當第二開關Q2跨壓vCr2下降至0，第二開關Q2的本體二極體(body diode)導通，模式三結束。When the second switch Q2 falls to zero across the voltage vCr2 , the body diode of the second switch Q2 is turned on, and the mode three ends.
模式四(時間：tMode four (time:t33~t~T44)：):
參閱圖2及圖3d，第一及第二開關Q1、Q2皆不導通。Referring to Figures 2 and 3d, the first and second switchesQ1 andQ2 are not turned on.
第二開關Q2跨壓vCr2箝位在0，而且vCr1=vin。第二開關Q2之本體二極體導通，第二開關Q2之跨壓為零，在流經第二開關電流iQ2變成正值之前，必須將第二開關Q2切換為導通，達成ZVS操作。The second switch Q2 is clamped at 0 across voltage vCr2 and vCr1 = vin . Before the second switch Q2 of the body diode is turned on, the voltage across the second switch Q2 is zero, the current flowing through the second switch iQ2 becomes positive, the second switch Q2 must be switched on, to achieve ZVS operating.
第一變壓器T1的一次側電壓vP1=0，而且vP2=0，共振電感電壓vLr-vC2，共振電感電流iLr線性下降。The primary side voltage vP1 =0 of the first transformer T1 and vP2 =0, the resonant inductor voltage vLr -vC2 , the resonant inductor current iLr decreases linearly.
當第一二極體電流iD1上升至iL1，且第二二極體電流iD2下升至0，換向完成，第二二極體D2不導通，模式四結束。When the first diode current iD1 rises to iL1 and the second diode current iD2 rises to 0, the commutation is completed, the second diode D2 is not turned on, and the mode four ends.
模式五(時間：tMode five (time: t44~t~t55)：):
參閱圖2及圖3e，第一開關Q1不導通，而第二開關Q2導通。Referring to FIG. 2 and FIG. 3E, a first switchQ1 is not turned on, the second switchQ2 is turned on.
第一二極體電流iD1=iL2，且第二二極體電流iD2=0，vP2=vC2，iLm2線性上升，斜率為vC2/Lm，第二變壓器T2的二次側電壓vS2=nvp2>0，此時儲存在第二電容C2之能量經由第二變壓器T2傳遞至輸出負載側。此時第一變壓器T1經由初級側二極體Dr作磁通重置(flux resetting)，且vP1=-vP2，因此磁化電流iLm1線性下降。在輸出電感電流方面，因為vL2=2nvC2-vo>0，iL2線性上升；vL1=-vo，iL1線性下降，所以總輸出電流iO=iL1+iL2會有漣波相消的效果。The first diode current iD1 =iL2 , and the second diode current iD2 =0, vP2 =vC2 , iLm2 linearly rises, the slope is vC2 /Lm , and the second transformer T2 The secondary side voltage vS2 =nvp2 >0, at which time the energy stored in the second capacitor C2 is transferred to the output load side via the second transformer T2 . At this time, the first transformer T1 is flux resetted via the primary side diode Dr , and vP1 =−vP2 , so the magnetizing current iLm1 linearly decreases. In terms of output inductor current, since vL2 = 2nvC2 - vo > 0, iL2 rises linearly; vL1 = -vo , iL1 decreases linearly, so the total output current iO = iL1 + iL2 will be 涟The effect of wave cancellation.
模式六(時間：tMode six (time: t55~t~t66)：):
參閱圖2及圖3f，第一開關Q1不導通，而第二開關Q2不導通。Referring to Figures 2 and 3f, the first switchQ1 is not conducting, and the second switchQ2 is not conducting.
流經第二開關Q2的電流iQ2為正值且對其寄生電容Cr2充電，電壓vCr2上升，由於初級側二極體Dr導通，電壓vCr1和vCr2滿足vin=vCr1+vCr2，所以電容Cr1放電，電壓vCr1下降。由於第一及第二開關Q1、Q2的寄生電容Cr1和Cr2非常小，vCr2上升和vCr1下降非常快，因此模式六歷經的時間很短。The current iQ2 flowing through the second switchQ2 is positive and charges the parasitic capacitance Cr2 , and the voltage vCr2 rises. Since the primary side diode Dr is turned on, the voltages vCr1 and vCr2 satisfy vin = vCr1 . +vCr2 , so the capacitor Cr1 discharges and the voltage vCr1 drops. Since the parasitic capacitances Cr1 and Cr2 of the first and second switchesQ1 ,Q2 are very small, vCr2 rises and vCr1 drops very fast, so the mode 6 has a very short time.
當第二開關Q2跨壓vCr2上升至vC2，此時vCr1下降至vC1，第二變壓器T2的一次側電壓vP2=0而且vP1=0。當第二二極體D2開始導通，模式六結束。When the second switch Q2 rises to vC2 across the voltage vCr2 , vCr1 drops to vC1 at this time, and the primary side voltage vP2 =0 of the second transformer T2 and vP1 =0. When the second diode D2 begins to conduct, mode six ends.
模式七(時間：tMode seven (time:t66~t~T77)：):
參閱圖2及圖3g，第一及第二開關Q1、Q2皆不導通。Referring to FIG. 2 and FIG. 3g, the first and second switches Q1 and Q2 are not turned on.
第一及第二變壓器T1、T2的一次側電壓vP1和vP2箝位於零，第一及第二二極體D1、D2進行換向，模式七相似於模式三，磁化電感電壓箝位於零，iLm1和iLm2保持常數。共振電感Lr、寄生電容Cr1和Cr2形成共振電路，vCr2持續上升(vCr2>vC2)，vCr1持續下降(vCr2<vC1)。共振電感Lr跨正電壓，共振電流iLr上升，電流iD2遞增，iD1遞減。The primary side voltages vP1 and vP2 of the first and second transformers T1 and T2 are clamped at zero, the first and second diodes D1 and D2 are commutated, and the mode seven is similar to the mode three, the magnetizing inductance The voltage clamp is at zero and iLm1 and iLm2 remain constant. The resonant inductor Lr , the parasitic capacitances Cr1 and Cr2 form a resonant circuit, vCr2 continues to rise (vCr2 >vC2 ), and vCr1 continues to decrease (vCr2 <vC1 ). The resonant inductor Lr crosses the positive voltage, the resonant current iLr rises, the current iD2 increases, and iD1 decreases.
模式七的共振電感Lr及漏電感的初始儲能必須大於該二寄生電容Cr1和Cr2的初始儲能，方能使第一開關跨壓vCr1下降至零，達到零電壓切換(ZVS)的條件。The initial energy storage of the resonant inductor Lr and the leakage inductance of mode 7 must be greater than the initial energy storage of the two parasitic capacitances Cr1 and Cr2 in order to reduce the first switching voltage vCr1 to zero and achieve zero voltage switching (ZVS). )conditions of.
當第一開關電壓vCr1下降至零，第一開關Q1的本體二極體導通，模式七結束。When the first switching voltage vCr1 drops to zero, the body diode of the first switch Q1 is turned on, and the mode seven ends.
模式八(時間：tMode eight (time: t77~T~TSS+t+t00)：):
參閱圖2及圖3h，第一及第二開關Q1、Q2皆不導通。Referring to FIG. 2 and FIG. 3h, the first and second switches Q1 and Q2 are not turned on.
電流流經第一開關Q1之本體二極體，第一開關Q1之跨壓為零，第一開關跨壓vCr1箝位在零，且vCr2=vin。因為vCr1=0，當流經第一開關電流iQ1變成正值之前，必須將第一開關Q1切換為導通，達成ZVS操作。A current flowing through the first switch Q1 of the body diode, the first switch Q1 is zero voltage across the first switch voltage across vCr1 clamped to zero, and vCr2 = vin. Since vCr1 =0, before the first switching current iQ1 flows to a positive value, the first switch Q1 must be switched to be turned on to achieve a ZVS operation.
共振電感電壓vLrvC1，共振電感電流iLr線性上升，該二二極體D1、D2持續進行換向過程。Resonant inductor voltage vLr vC1 , the resonant inductor current iLr rises linearly, and thedipoles D1 , D2 continue to perform the commutation process.
當第二二極體電流iD2=iL1，且第一二極體電流iD1下降至0，第一二極體D1轉為不導通而換向完成，進入下一切換週期。When the second diode current iD2 =iL1 and the first diode current iD1 falls to 0, the first diode D1 turns non-conductive and the commutation is completed, and enters the next switching period.
實驗模擬Experimental simulation
由圖4可知，在vin=400V時，第一開關Q1之跨壓vQ1,ds都下降至零後，驅動信號vg1才切換為導通，達到ZVS性能，而第二開關Q2之跨壓vQ2,d都下降至零後，驅動信號vg2才切換為導通，達到ZVS性能。從圖中可知其電壓應力皆為vin，量測結果符合第一及第二開關Q1、Q2具有低電壓應力。When seen from the rear in FIG. 4, the vin = 400V, the voltage across the first switch Q vQ1, ds of1 has fallen to zero, the drive signal vg1 was switched on to achieve ZVS performance, the second switch Q2 After the voltage vQ2, d drops to zero, the drive signal vg2 is switched to conduct to achieve ZVS performance. It can be seen from the figure that the voltage stress is vin , and the measurement result is consistent with the first and second switches Q1 , Q2 having low voltage stress.
如圖5為輸出電感電流iL1、iL2及總輸出電流iO的波形量測圖，由模擬波形可知：在穩態操作下，iL1和iL2漣波反相，確實使漣波△iO(=0.3A)降低許多(△iL1△iL23.9A)，可選用較小的輸出濾波電容元件，可使得轉換器體積減小，提高功率密度。另外，iL1=iL2=7.5A確實分擔總輸出電流(15A)，可分散磁性元件的功率損失及熱應力，且具有高輸出電流且低輸出電流漣波的性能。Figure 5 shows the waveform measurement of the output inductor currents iL1 , iL2 and the total output current iO . It can be seen from the analog waveform that under steady state operation, iL1 and iL2 are chopped in opposite directions, which indeed makes the ripple △ iO (=0.3A) is reduced a lot (△iL1 △iL2 3.9A), a smaller output filter capacitor can be used to reduce the converter's size and increase the power density. In addition, iL1 =iL2 =7.5A does share the total output current (15A), disperses the power loss and thermal stress of the magnetic element, and has high output current and low output current chopping performance.
如圖6為第一及第二二極體D1~D2的電流量測圖，從圖中可看出該二開關Q1、Q2的驅動信號vg1、vg2與第一及第二二極體D1~D2的電流換向的波形。6 is a current measurement diagram of the first and second diodes D1 to D2 , and the driving signals vg1 and vg2 of the two switches Q1 and Q2 can be seen from the figure. The current commutation waveform of the diodes D1 to D2 .
綜上所述，上述實施例具有以下優點：In summary, the above embodiment has the following advantages:
1.每一開關Q1、Q2有較低的電壓應力，其開關應力等同於輸入電壓vin，適用於高輸入電壓的應用。1. Each switch Q1 , Q2 has a lower voltage stress, and its switching stress is equivalent to the input voltage vin , which is suitable for high input voltage applications.
2.只包含二個開關Q1、Q2，能降低硬體成本。2. Only two switches Q1 and Q2 are included , which can reduce the hardware cost.
3.第一及第二開關Q1、Q2都能達到零電壓切換(ZVS)操作，減少切換損失，能提高功率轉換效率。3. The first and second switches Q1 and Q2 can achieve zero voltage switching (ZVS) operation, reduce switching losses, and improve power conversion efficiency.
4.利用第一及第二輸出電感L1、L2來分擔輸出電流，適合高輸出電流應用。4. The first and second output inductors L1 and L2 are used to share the output current, which is suitable for high output current applications.
5.流經第一及第二輸出電感L1、L2的電流的漣波為反相，可進行漣波互消作用，所以具有低輸出電流漣波。5. The chopping of the current flowing through the first and second output inductors L1 and L2 is inverted, and the chopping canceling action can be performed, so that the output current is chopped with low output current.
6.輸出電容CO的操作頻率為開關Q1、Q2切換頻率兩倍，在相同的輸出漣波規格下，可選擇較小之濾波元件。6. The operating frequency of the output capacitor CO is twice the switching frequency of the switches Q1 and Q2 . Under the same output chopping specification, a smaller filter component can be selected.
惟以上所述者，僅為本發明之較佳實施例而已，當不能以此限定本發明實施之範圍，即大凡依本發明申請專利範圍及發明說明內容所作之簡單的等效變化與修飾，皆仍屬本發明專利涵蓋之範圍內。However, the above is only the preferred embodiment of the present invention, and the scope of the present invention cannot be limited thereto, that is, the patent application according to the present inventionThe scope of the invention and the equivalent equivalents and modifications of the invention are still within the scope of the invention.
T1‧‧‧第一變壓器T1 ‧‧‧First Transformer
D2‧‧‧第二二極體D2 ‧‧‧Secondary
T2‧‧‧第二變壓器T2 ‧‧‧second transformer
L1‧‧‧第一輸出電感L1 ‧‧‧first output inductor
C1‧‧‧第一電容C1 ‧‧‧first capacitor
L2‧‧‧第二輸出電感L2 ‧‧‧second output inductor
C2‧‧‧第二電容C2 ‧‧‧second capacitor
CO‧‧‧輸出電容CO ‧‧‧ output capacitor
Lr‧‧‧共振電感Lr ‧‧‧Resonance inductance
Lp1、Lp2‧‧‧一次側繞組Lp1 , Lp2 ‧‧‧ primary winding
Q1‧‧‧第一開關Q1 ‧‧‧First switch
Ls1、Ls2‧‧‧二次側繞組Ls1 , Ls2 ‧‧‧ secondary winding
Q2‧‧‧第二開關Q2 ‧‧‧Second switch
Lm1~Lm2‧‧‧磁化電感Lm1 ~Lm2 ‧‧‧ Magnetizing inductance
Dr‧‧‧初級側二極體Dr ‧‧‧Primary side diode
vin‧‧‧輸入電壓vin ‧‧‧Input voltage
D1‧‧‧第一二極體D1 ‧‧‧First Diode
VO‧‧‧輸出電壓VO ‧‧‧Output voltage
圖1是本發明低電壓應力直流轉換器之較佳實施例的一電路圖；圖2是該較佳實施例的一時序圖；圖3a是該較佳實施例於模式一的一電路圖；圖3b是該較佳實施例於模式二的一電路圖；圖3c是該較佳實施例於模式三的一電路圖；圖3d是該較佳實施例於模式四的一電路圖；圖3e是該較佳實施例於模式五的一電路圖；圖3f是該較佳實施例於模式六的一電路圖；圖3g是該較佳實施例於模式七的一電路圖；圖3h是該較佳實施例於模式八的一電路圖；圖4是該較佳實施例的第一種量測圖；圖5是該較佳實施例的第二種量測圖；及圖6是該較佳實施例的第三種量測圖。1 is a circuit diagram of a preferred embodiment of a low voltage stress DC converter of the present invention; FIG. 2 is a timing diagram of the preferred embodiment; FIG. 3a is a circuit diagram of the preferred embodiment in mode one; Figure 3c is a circuit diagram of the preferred embodiment in mode three; Figure 3d is a circuit diagram of the preferred embodiment in mode four; Figure 3e is a preferred embodiment of the preferred embodiment Figure 3f is a circuit diagram of the preferred embodiment in mode six; Figure 3g is a circuit diagram of the preferred embodiment in mode seven; Figure 3h is a preferred embodiment of mode eight FIG. 4 is a first measurement diagram of the preferred embodiment; FIG. 5 is a second measurement diagram of the preferred embodiment; and FIG. 6 is a third measurement of the preferred embodiment. Figure.
T1‧‧‧第一變壓器T1 ‧‧‧First Transformer
L1‧‧‧第一輸出電感L1 ‧‧‧first output inductor
T2‧‧‧第二變壓器T2 ‧‧‧second transformer
L2‧‧‧第二輸出電感L2 ‧‧‧second output inductor
C1‧‧‧第一電容C1 ‧‧‧first capacitor
CO‧‧‧輸出電容CO ‧‧‧ output capacitor
C2‧‧‧第二電容C2 ‧‧‧second capacitor
Lp1、Lp2‧‧‧一次側繞組Lp1 , Lp2 ‧‧‧ primary winding
Lr‧‧‧共振電感Lr ‧‧‧Resonance inductance
Ls1、Ls2‧‧‧二次側繞組Ls1 , Ls2 ‧‧‧ secondary winding
Q1‧‧‧第一開關Q1 ‧‧‧First switch
vin‧‧‧輸入電壓vin ‧‧‧Input voltage
Q2‧‧‧第二開關Q2 ‧‧‧Second switch
VO‧‧‧輸出電壓VO ‧‧‧Output voltage
Dr‧‧‧初級側二極體Dr ‧‧‧Primary side diode
D1‧‧‧第一二極體D1 ‧‧‧First Diode
D2‧‧‧第二二極體D2 ‧‧‧Secondary
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| TW101122815ATWI441435B (en) | 2012-06-26 | 2012-06-26 | Low voltage stress DC converter |
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| TW101122815ATWI441435B (en) | 2012-06-26 | 2012-06-26 | Low voltage stress DC converter |
| Publication Number | Publication Date |
|---|---|
| TW201401746A TW201401746A (en) | 2014-01-01 |
| TWI441435Btrue TWI441435B (en) | 2014-06-11 |
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| TW101122815ATWI441435B (en) | 2012-06-26 | 2012-06-26 | Low voltage stress DC converter |
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| TW (1) | TWI441435B (en) |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| TWI607622B (en)* | 2016-09-23 | 2017-12-01 | 亞力電機股份有限公司 | Step-up direct current converter |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| TWI607622B (en)* | 2016-09-23 | 2017-12-01 | 亞力電機股份有限公司 | Step-up direct current converter |
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| TW201401746A (en) | 2014-01-01 |
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