【発明の詳細な説明】［概要］ＬＩＮＣ方式の定振幅波合成形増幅器に関し。[Detailed description of the invention][overview]Regarding LINC type constant amplitude wave synthesis amplifier.
定振幅波合成形増幅器の低消費電力化、小型化、および
高速動作化を図ることを目的とし。The purpose of this project is to reduce power consumption, downsize, and operate at high speed in constant-amplitude wave synthesis amplifiers.
制御信号に応じて発振周波数が変化される周波数可変発
振器であってその出力信号が第１の定振幅波となるもの
と、この周波数可変発振器からの第１の定振幅波と入力
信号波とを合成して第２の定振幅波を生成する第１の合
成回路と、第１．第２の定振幅波をそれぞれ別々に増幅
する二つの増幅器と、二つの増幅器でそれぞれ増幅され
た定振幅波を合成して出力信号波を生成する第２の合成
回路と、第１．第２の定振幅波を比較してそれらの振幅
差に応じた値を周波数可変発振器に位相誤差を修正する
制御信号として与える制御回路とを具備してなる。A variable frequency oscillator whose oscillation frequency is changed according to a control signal and whose output signal is a first constant amplitude wave, and a first constant amplitude wave from this variable frequency oscillator and an input signal wave. a first combining circuit that combines to generate a second constant amplitude wave; two amplifiers that separately amplify the second constant amplitude waves; a second synthesis circuit that generates an output signal wave by synthesizing the constant amplitude waves respectively amplified by the two amplifiers; The control circuit includes a control circuit that compares the second constant amplitude waves and provides a value corresponding to their amplitude difference to the variable frequency oscillator as a control signal for correcting a phase error.
［産業上の利用分野］本発明は、 Ｌ Ｉ Ｎ Ｃ（Ｌｉｎｅａｒ Ａａｍｐ
ｌｉｆｉｃａｔｉｏｎｗｉｔｈ Ｎｏｎ１ｉｎｅａｒ
Ｃｏｍｐｏｒ＋ｅｎｔｓ）方式の定振幅波合成形増幅器
に関する。[Industrial Field of Application] The present invention is based on LIN C (Linear Amp).
ification with Non1inear
The present invention relates to a constant-amplitude wave-synthesizing amplifier using the Compor+ents method.
定振幅波合成形増幅器は２例えば移動通信用無線装置、
多重無線装置、衛星通信無線装置、あるいは放送機器等
の各種通信装置に適用されている低歪かつ高効率電力の
増幅器である。この定振幅波合成形増幅器は低消費電力
で小規模な回路構成で実現できることが必要とされてい
る。The constant amplitude wave synthesis amplifier is used in two applications, for example, mobile communication radio equipment,
This is a low-distortion, high-efficiency power amplifier that is applied to various communication devices such as multiplex radio equipment, satellite communication radio equipment, and broadcasting equipment. There is a need for this constant amplitude wave synthesis amplifier to be realized with low power consumption and a small-scale circuit configuration.
［従来の技術］従来、低歪かつ高効率を達成できる電力増幅器としては
ＬＩＮＣ方式を用いた定振幅波合成形増幅器が知られて
いる。この定振幅波合成形増幅器の原理はベル研究所の
り、Ｃ，ＣＯＸによって提案されたもので、ＩＥＥＥ、
Ｃ０Ｍ−２２，ＰＬ９４２等に開示されている。また応
用例として。[Prior Art] Conventionally, a constant amplitude wave synthesis amplifier using the LINC method is known as a power amplifier that can achieve low distortion and high efficiency. The principle of this constant amplitude wave-synthesizing amplifier was proposed by Nori, C., and COX at Bell Laboratories, and was published in the IEEE.
It is disclosed in C0M-22, PL942, etc. Also as an application example.
１里等の方式が、特開平１−２８４１０６号公報等に開
示されている。A method such as Ichiri is disclosed in Japanese Patent Application Laid-Open No. 1-284106.
第１５図には、従来の定振幅波合成形増幅器の構成例が
示される。同図において、６１は定振幅波演算回路であ
り、ディジタル信号処理回路で構成されており、入力さ
れた信号波Ｘとの合成波が定振幅となるような直交波Ｙ
を演算して、この入力信号波Ｘと直交波Ｙを合成するこ
とで、定包絡線の等振幅な二つの定振幅波Ａ、Ｂを生成
し出力する。なお、ここで英大文字のＸ、Ｙ、Ａ、Ｂは
位相を含むベクＩ〜ル量を表すものとする。FIG. 15 shows an example of the configuration of a conventional constant amplitude wave synthesis amplifier. In the figure, 61 is a constant amplitude wave calculation circuit, which is composed of a digital signal processing circuit, and generates an orthogonal wave Y such that the composite wave with the input signal wave X has a constant amplitude.
By calculating the input signal wave X and the orthogonal wave Y, two constant amplitude waves A and B of equal amplitude with a constant envelope are generated and output. Note that the capital letters X, Y, A, and B represent vector quantities including phase.
６２．６３は定振幅波演算回路６１の二つの定振幅波Ａ
、Ｂをそれぞれ別々に増幅する増幅器であり、０級等の
高効率な非線形増幅器が使用されている。62.63 are two constant amplitude waves A of the constant amplitude wave calculation circuit 61
, B separately, and a highly efficient nonlinear amplifier such as a class 0 amplifier is used.
６４は合成回路であり、増幅器６２．６３で増幅後の定
振幅波ｋＡ、ｋＢを合成して入力波Ｘの増幅出力として
出力波ｋＸを生成する。Reference numeral 64 denotes a synthesis circuit, which synthesizes the constant amplitude waves kA and kB amplified by amplifiers 62 and 63 to generate an output wave kX as the amplified output of the input wave X.
この従来の定振幅波合成形増幅器は、入力波Ｘを定振幅
波演算回路６１で二つの定振幅波Ａ、 Ｈに変換して、
増幅器６２．６３でそれぞれ増幅し、その増幅出力ｋＡ
、ｋＢを合成回路６４で合成することで９元の人力波Ｘ
の増幅出力である出力波ｋＸを得るものである。This conventional constant amplitude wave synthesis amplifier converts an input wave X into two constant amplitude waves A and H using a constant amplitude wave calculation circuit 61.
Amplified by amplifiers 62 and 63, and the amplified output kA
, kB in the synthesis circuit 64, the human power wave X of 9 yuan is generated.
This is to obtain an output wave kX which is the amplified output of .
この定振幅波合成形増幅器においては、増幅器６２．６
３は定包絡線の定振幅波Ａ、Ｂを増幅するだけのもので
あるから、線形増幅器を用いる必要はな（，６級等の高
効率な非線形増幅器を使用することができ、しかも出力
側の合成回路６４で元の信号が復元できるから、非線形
増幅器によりながらも歪が少な（直線性の良い増幅が可
能である。このように定振幅波合成形増幅器は非線形増
幅器を用いて高効率、低歪の増幅器を実現できるもので
ある。In this constant amplitude wave synthesis amplifier, the amplifier 62.6
3 only amplifies the constant amplitude waves A and B with a constant envelope, so there is no need to use a linear amplifier (a high-efficiency nonlinear amplifier such as class 6 can be used, and the output side Since the original signal can be restored by the combining circuit 64, it is possible to perform amplification with little distortion (good linearity) even though it is a nonlinear amplifier.In this way, the constant amplitude wave combining amplifier uses a nonlinear amplifier to achieve high efficiency, This makes it possible to realize a low distortion amplifier.
〔発明が解決しようとする課題］一般に従来の定振幅波合成形増幅器では、入力波Ｘから
二つの定振幅波Ａ、Ｂを生成する定振幅波演算回路６１
は、ディジタル信号処理回路を用いて構成されている。[Problems to be Solved by the Invention] Generally, in a conventional constant amplitude wave synthesis amplifier, a constant amplitude wave calculation circuit 61 generates two constant amplitude waves A and B from an input wave X.
is constructed using a digital signal processing circuit.
しかしながらディジタル信号処理回路を用いた場合、定
振幅波演算回路の消費電力が大きくなり９また高速信号
処理に追従できなく、さらに回路規模が大きくなるとい
問題点がある。However, when a digital signal processing circuit is used, there are problems in that the power consumption of the constant amplitude wave calculation circuit becomes large,9 and it is not possible to follow high-speed signal processing, and furthermore, the circuit scale increases.
このため、従来の定振幅波合成形増幅器は、増幅器消費
電力がディジタル信号処理回路のそれに比べて十分に大
きいためディジタル信号処理回路での消費電力が無視で
きる場合、あるいは大きさの制限がなく低速信号を伝送
する装置に使用する場合などに使途が限定される。以上
の問題の解決には、半導体の低消費電力化、高速化、ｍ
ＬｓＩ化等の半導体技術の進歩をまたなければならない
。For this reason, conventional constant-amplitude wave synthesis amplifiers are used in cases where the power consumption of the amplifier is sufficiently large compared to that of the digital signal processing circuit and the power consumption of the digital signal processing circuit can be ignored, or when there is no size restriction and the power consumption is low speed. Its use is limited to devices that transmit signals. To solve the above problems, it is necessary to reduce the power consumption of semiconductors, increase their speed,
We must continue to advance in semiconductor technology, such as LSI technology.
本発明はかかる技術的諸問題に鑑みなされたものであり
、その目的とするところは、定振幅波合成形増幅器の低
消費電力化９回路規模の小型化。The present invention has been made in view of these technical problems, and its purpose is to reduce the power consumption of a constant amplitude wave synthesis amplifier and to reduce the size of nine circuits.
および高速動作化を図ることにある。and to achieve high-speed operation.
［課題を解決するための手段］第１図〜第４図はそれぞれ本発明に係る原理説明図であ
る。[Means for Solving the Problems] FIGS. 1 to 4 are diagrams each illustrating the principle of the present invention.
本発明に係る定振幅波合成形増幅器は、一つの形態とし
て、第１図に示されるように、制御信号に応じて発振周
波数が変化される周波数可変発振器４１であってその出
力信号が第１の定振幅波となるものと９周波数可変発振
器４１からの第１の定振幅波と入力信号波とを合成して
第２の定振幅波を生成する第１の合成回路４２と、第１
．第２の定振幅波をそれぞれ別々に増幅する二つの増幅
器４３．４４と、二つの増幅器４３．４４でそれぞれ増
幅された定振幅波を合成して出力信号波を生成する第２
の合成回路４５と、第１．第２の定振幅波を比較してそ
れらの振幅差に応じた値を周波数可変発振器４１に位相
誤差を修正する制御信号として与える制御回路４６とを
具備してなる。One form of the constant amplitude wave synthesis amplifier according to the present invention is a variable frequency oscillator 41 whose oscillation frequency is changed according to a control signal, as shown in FIG. a first synthesis circuit 42 that generates a second constant amplitude wave by synthesizing the first constant amplitude wave from the nine frequency variable oscillator 41 and the input signal wave to generate a second constant amplitude wave;
．． Two amplifiers 43 and 44 separately amplify the second constant amplitude waves, and a second amplifier 43 and 44 that synthesizes the constant amplitude waves respectively amplified by the two amplifiers 43 and 44 to generate an output signal wave.
a first combining circuit 45 and a first combining circuit 45; The control circuit 46 compares the second constant amplitude waves and provides a value corresponding to their amplitude difference to the variable frequency oscillator 41 as a control signal for correcting a phase error.
本発明に係る定振幅波合成形増幅器は、他の形態として
、第２図に示されるように、制御信号に応じて発振周波
数が変化される周波数可変発振器４１であってその出力
信号が第１の定振幅波となるものと９周波数可変発振器
４１からの第１の定振幅波と入力信号波とを合成して第
２の定振幅波を生成する第１の合成回路４２と、第１．
第２の定振幅波をそれぞれ別々に増幅する二つの増幅器
４３．４４と、二つの増幅器４３．４４でそれぞれ増幅
された定振幅波を合成して出力信号波を生成する第２の
合成回路４５と、第２の合成回路４５の出力信号波と入
力信号波とを比較して出力信号波に含まれる歪成分を抽
出し、この歪成分を周波数可変発振器４１に位相誤差を
修正する制御信号として与える制御回路４７とを具備し
てなる。As another form of the constant amplitude wave synthesis amplifier according to the present invention, as shown in FIG. A first synthesis circuit 42 generates a second constant amplitude wave by synthesizing the first constant amplitude wave from the nine frequency variable oscillator 41 and the input signal wave to generate a constant amplitude wave;
Two amplifiers 43, 44 that separately amplify the second constant amplitude waves, and a second synthesis circuit 45 that synthesizes the constant amplitude waves respectively amplified by the two amplifiers 43, 44 to generate an output signal wave. The output signal wave of the second synthesis circuit 45 is compared with the input signal wave to extract the distortion component contained in the output signal wave, and this distortion component is sent to the variable frequency oscillator 41 as a control signal for correcting the phase error. The control circuit 47 is provided with a control circuit 47 for giving
本発明に係る定振幅波合成形増幅器は、他の形態として
、第３図または第４図に示されるように、上述の二つの
形態の定振幅波合成形増幅器において、前記二つの増幅
器４３．４４の少なくとも一方の前段に、その増幅器に
入力される定振幅波の振幅を制御信号に応じて変化させ
る利得可変増幅器４８が更に備えられる。As another form of the constant amplitude wave combining amplifier according to the present invention, as shown in FIG. 3 or 4, in the above two forms of the constant amplitude wave combining amplifier, the two amplifiers 43. A variable gain amplifier 48 is further provided in front of at least one of the amplifiers 44 and 48 for changing the amplitude of a constant amplitude wave input to the amplifier according to a control signal.
この利得可変増幅器４８は前記第２の合成回路４５の出
力信号波と入力信号波とを比較して抽出した出力信号波
の歪成分を、振幅誤差を修正するための制御信号とする
ように構成することができる。This variable gain amplifier 48 is configured to compare the output signal wave of the second synthesis circuit 45 with the input signal wave and use the extracted distortion component of the output signal wave as a control signal for correcting the amplitude error. can do.
またこの利得可変増幅器は自身の出力信号を所定の一定
値と比較し、その差分に応じた値を振幅誤差を修正する
ための制御信号とするように構成することができる。Further, this variable gain amplifier can be configured to compare its output signal with a predetermined constant value and use a value corresponding to the difference as a control signal for correcting the amplitude error.
［作用］第１図の形態の定振幅波合成形増幅器では、制御回路４
６により周波数可変発振器４１から出力される第１の定
振幅波と第１の合成回路から出力される第２の定振幅波
との振幅差を検出してこの振幅差がなくなるように９周
波数可変発振器４１の周波数を制御する。これにより第
１．第２の定振幅波は包絡線振幅が一定の波となり、こ
れを増幅する増幅器４３．４４としては、高効率な非線
形増幅器を利用することが可能となり、かつ、かかる非
線形増幅器を使用しつつも入力信号波に対して線形で低
歪な増幅が可能となる。[Function] In the constant amplitude wave synthesis amplifier of the form shown in FIG.
6 detects the amplitude difference between the first constant amplitude wave output from the variable frequency oscillator 41 and the second constant amplitude wave output from the first synthesis circuit, and adjusts the frequency 9 to eliminate this amplitude difference. Controls the frequency of the oscillator 41. This leads to the first. The second constant amplitude wave is a wave with a constant envelope amplitude, and as the amplifiers 43 and 44 for amplifying this, it is possible to use a highly efficient nonlinear amplifier, and even though such a nonlinear amplifier is used, It becomes possible to amplify the input signal wave linearly and with low distortion.
第２図の形態の定振幅波合成形増幅器では、制御回路４
７により出力信号波に含まれる歪成分を抽出し、この歪
成分が低減されるように周波数可変発振器４１の周波数
を制御する。この制御の結果、出力信号波の歪成分が低
減された時には、増幅器４３．４４に入力される定振幅
波が定包絡線の波になっていると考えることができる。In the constant amplitude wave synthesis amplifier of the form shown in FIG.
7 extracts the distortion component contained in the output signal wave, and controls the frequency of the variable frequency oscillator 41 so that this distortion component is reduced. As a result of this control, when the distortion component of the output signal wave is reduced, it can be considered that the constant amplitude waves input to the amplifiers 43 and 44 have become constant envelope waves.
第３図、第４図の形態の定振幅波合成形増幅器では、上
記二つの形態の定振幅波合成形増幅器により定振幅波の
位相誤差の制御を行ったにもかかわらず、まだ第１．第
２の定振幅波に振幅誤差がある場合に、これら第１．第
２の定振幅波の振幅を利得可変増幅器４８で調整して振
幅誤差を低減させている。In the constant amplitude wave combining amplifiers of the configurations shown in FIGS. 3 and 4, although the phase error of the constant amplitude wave is controlled by the two types of constant amplitude wave combining amplifiers described above, the first. When there is an amplitude error in the second constant amplitude wave, these first. The amplitude of the second constant amplitude wave is adjusted by the variable gain amplifier 48 to reduce amplitude errors.
利得可変増幅器４８の利得制御の仕方としては、第２の
合成回路４５の出力信号波に含まれる歪成分を低減する
ように利得を調整する方法、あるいは利得可変増幅器４
８の出力信号の増幅が所定の一定値になるようにその利
得を調整する方法などが可能である。The gain of the variable gain amplifier 48 can be controlled by adjusting the gain so as to reduce the distortion component included in the output signal wave of the second combining circuit 45, or by adjusting the gain of the variable gain amplifier 48.
It is possible to adjust the gain so that the amplification of the output signal of No. 8 becomes a predetermined constant value.
［実施例］以下１図面を参照して本発明の詳細な説明する。なお、
以下の説明では、同一の参照番号が付されたものは各図
を通じて同じ回路構成要素を表しているものとする。[Example] The present invention will be described in detail below with reference to one drawing. In addition,
In the following description, it is assumed that the same reference numerals represent the same circuit components throughout the figures.
本発明の一実施例としての定振幅波合成形増幅器が第５
図に示される。第５図において、入力端子１０には入力
信号波として入力波Ｘが入力され、この入力波Ｘは更に
加算器ｌに入力されて。A constant amplitude wave synthesis amplifier as an embodiment of the present invention is a fifth embodiment of the present invention.
As shown in the figure. In FIG. 5, an input wave X is input as an input signal wave to an input terminal 10, and this input wave X is further input to an adder l.
ここで電圧制御発振器２から出力される定振幅波Ａと加
算されて定振幅波Ｂが合成され出力される。なお、ここ
でＸ、Ａ、Ｂはそれぞれ位相を含むベクトル量を表して
いるものとする。Here, it is added to the constant amplitude wave A output from the voltage controlled oscillator 2 to synthesize the constant amplitude wave B and output. It is assumed here that X, A, and B each represent a vector quantity including a phase.
この電圧制御発振器２の定振幅波Ａと加算器１の定振幅
波Ｂは演算回路５に入力される。演算回路５は入力され
た信号に対して Ａ ２ ８２の演算を行うことにより
、定振幅波ＡとＢの振幅差に応じた値を出力信号として
生成し、この差分値を低域フィルタ６を介して電圧制御
発振器２に位相誤差を修正するための制御信号として与
えるようになっている。The constant amplitude wave A of the voltage controlled oscillator 2 and the constant amplitude wave B of the adder 1 are input to an arithmetic circuit 5. The arithmetic circuit 5 performs the calculation of A 2 82 on the input signal to generate a value corresponding to the amplitude difference between the constant amplitude waves A and B as an output signal, and this difference value is passed through the low-pass filter 6. The signal is supplied to the voltage controlled oscillator 2 as a control signal for correcting phase errors.
また定振幅波ＡとＢはそれぞれ別々に増幅器３と４に入
力される。増幅器３，４は増幅度にの増幅器であり、入
力信号が包絡線の振幅が一定の定振幅波であるので、Ｃ
級増幅器などの高効率な非線形増幅器を用いることがで
きる。Further, the constant amplitude waves A and B are separately input to amplifiers 3 and 4, respectively. Amplifiers 3 and 4 are amplifiers with amplification degree, and since the input signal is a constant amplitude wave with a constant amplitude of the envelope, C
Highly efficient nonlinear amplifiers such as class amplifiers can be used.
増幅器３，４の増幅出力ｋＡ、ｋＢはそれぞれ逆相ハイ
ブリッド回路７の入力端子７１．７２に人力される。逆
相ハイブリッド回路７は各入力端子に入力された信号を
２分岐して二つの出力端子に出力する回路であり、同方
向出力端子（端子７１と７３間、あるいは端子７２と７
４間）に同相成分を、また交差方向出力端子（端子７１
と７４間、あるいは端子７２と７３間）に逆相成分をそ
れぞれ出力し、各出力端子７３．７４において。The amplified outputs kA and kB of the amplifiers 3 and 4 are input to input terminals 71 and 72 of the anti-phase hybrid circuit 7, respectively. The anti-phase hybrid circuit 7 is a circuit that branches a signal input to each input terminal into two and outputs it to two output terminals.
4), and the cross-direction output terminal (terminal 71).
and 74 or between terminals 72 and 73), and at each output terminal 73 and 74.
二つの入力端子７１．７２にそれぞれ入力された信号を
、その一方を逆相にしてベクトル合成し出力するように
なっている。The signals inputted to the two input terminals 71 and 72 are vector-combined with one of them in reverse phase and output.
この逆相ハイブリッド回路７の出力端子７４には９人力
波Ｘの増幅出力である出力波ｋＸが出力され、一方、出
力端子７３に出力される信号は終端抵抗器８で終端され
るようになっている。Output wave kX, which is the amplified output of nine human power waves ing.
この実施例回路の動作が図面を参照しつつ以下に説明さ
れる。The operation of this embodiment circuit will be explained below with reference to the drawings.
入力端子１０に入力された入力波Ｘは加算器１によって
電圧制御発振器２からの定振幅波Ａとベクトル的に加算
されて合成波Ｂが出力される。この場合、何等の位相制
御も行わない場合には、任意の入力波Ｘに対して９人力
波Ｘと定振幅波Ａのベクトル合成波Ｂは第６図に示され
るような関係にあり、定振幅波Ａと合成波Ｂの振幅は等
しくなく１合成波Ｂの振幅は入力波Ｘに応じて変動する
ので２合成波Ｂは定振幅波とはならない。An input wave X input to an input terminal 10 is vectorially added to a constant amplitude wave A from a voltage controlled oscillator 2 by an adder 1, and a composite wave B is output. In this case, if no phase control is performed, the vector composite wave B of the nine human power waves X and the constant amplitude wave A for any input wave The amplitudes of the amplitude wave A and the composite wave B are not equal, and the amplitude of one composite wave B varies depending on the input wave X, so the two composite waves B do not become a constant amplitude wave.
ここで演算回路５により定振幅波への振幅と合成波Ｂの
振幅の差分に応じた値Ａ ２−８２を演算し、この値Ａ
２ Ｂ ２を電圧制御発振器２に制御電圧として与
え、この電圧制御発振器２の発振周波数（あるいは位相
）を変化させて。Here, the arithmetic circuit 5 calculates a value A2-82 corresponding to the difference between the amplitude of the constant amplitude wave and the amplitude of the composite wave B, and this value A
2 B 2 is applied to the voltage controlled oscillator 2 as a control voltage, and the oscillation frequency (or phase) of this voltage controlled oscillator 2 is changed.
Ａ”−８２＝０となるように制御を行う。すなわち、定振幅波Ａと合成
波Ｂの振幅が等しくなるように、それらの間の位相角θ
を調整するものである。A"-82=0. In other words, the phase angle θ between the constant amplitude wave A and the composite wave B is adjusted so that the amplitudes of the constant amplitude wave A and the composite wave B are equal.
This is to adjust the
定振幅波Ａと合成波Ｂの振幅が等しくなった場合、Ａ”
−８２＝Ｏが成り立って回路は安定となり、第７図に示
されるように、定振幅波Ａと合成波Ｂのベクトル軌跡は
円上にのり９合成波Ｂは定振幅波となる。When the amplitudes of constant amplitude wave A and composite wave B become equal, A”
-82=O holds, the circuit becomes stable, and as shown in FIG. 7, the vector loci of the constant amplitude wave A and the composite wave B lie on a circle, and the composite wave B becomes a constant amplitude wave.
この結果、増幅器３．４に人力される信号は共に振幅が
一定の信号となるので、増幅器３，４としては線形増幅
器を用いる必要がなくなり、Ｃ級増幅器などの効率のよ
い非線形増幅器を利用することが可能となる。As a result, the signals input to the amplifiers 3 and 4 both have constant amplitudes, so it is no longer necessary to use linear amplifiers as the amplifiers 3 and 4, and efficient nonlinear amplifiers such as class C amplifiers can be used. becomes possible.
増幅器３．４の増幅出力ｋＡ、ｋＢは逆相ハイブリッド
回路７に入力され、ここで合成される。The amplified outputs kA and kB of the amplifier 3.4 are input to the anti-phase hybrid circuit 7, where they are combined.
第８図にはこの増幅出力ｋＡ、ｋＢの位相関係が示され
る。ここで逆相ハイブリッド回路７の出力端子７４には
、ｋ （Ｂ−Ａ）の出力波が出力され、この出力波は入
力波Ｘの増幅出力となる。FIG. 8 shows the phase relationship between the amplified outputs kA and kB. Here, an output wave of k (B-A) is outputted to the output terminal 74 of the anti-phase hybrid circuit 7, and this output wave becomes an amplified output of the input wave X.
すなわち。Namely.
ｋＸ＝ｋ （Ｂ−Ａ）である。kX=k (B-A)It is.
本発明の実施にあたっては種々の変形形態が可能である
。第９図にはかかる本発明の他の実施例が示される。こ
の実施例が第５図の実施例と相違する点は演算回路５の
構成であり、この実施例の場合には、電圧制御発振器２
からの定振幅波Ａと加算器ｌからの定振幅波Ｂとを合成
回路１１に入力させて、Ａ＋ＢとＡ−Ｂの合成波を作り
、これらを乗算器１２で乗算することで Ａ２Ｂ２の近
似出力を生成し、この近似出力を低域フィルタ６を介し
て電圧制御発振器２への制御電圧としている。この実施
例回路の他の構成とその動作原理は前述の第５図の実施
例と同じである。Various modifications are possible in implementing the invention. FIG. 9 shows another embodiment of the present invention. This embodiment differs from the embodiment shown in FIG. 5 in the configuration of the arithmetic circuit 5.
The constant amplitude wave A from the adder 1 and the constant amplitude wave B from the adder l are input to the synthesis circuit 11 to create composite waves of A+B and A-B, and these are multiplied by the multiplier 12 to approximate A2B2. An output is generated, and this approximate output is passed through a low-pass filter 6 as a control voltage to the voltage controlled oscillator 2. The other structure and operating principle of this embodiment circuit are the same as the embodiment shown in FIG. 5 described above.
また第１０図には本発明の更に他の実施例が示される。Further, FIG. 10 shows still another embodiment of the present invention.
第１Ｏ図において、加算器１，１！圧制御発振器２．増
幅器３，４．低域フィルタ６、逆相ハイブリッド回路７
は前述したものと同じである。この実施例が前述の二つ
の実施例と相違する点は電圧制御発振器２の制御電圧の
作り方にある。すなわちこの実施例では、逆相ハイブリ
ッド回路７からの出力波ｋＸを１／に減衰器１３とリミ
ッタ増幅器１４を介して乗算器１６に導き。In FIG. 1O, adders 1,1! Pressure controlled oscillator 2. Amplifiers 3, 4. Low-pass filter 6, anti-phase hybrid circuit 7
is the same as described above. This embodiment differs from the previous two embodiments in how the control voltage of the voltage controlled oscillator 2 is generated. That is, in this embodiment, the output wave kX from the anti-phase hybrid circuit 7 is guided to a multiplier 16 via an attenuator 13 and a limiter amplifier 14 to 1/1.
方、入力波Ｘをリミッタ増幅器１５を介して乗算器１６
に導き、この乗算器１６で両方の入力を乗算することで
、その出力信号として出力波ｋＸに含まれる歪成分△を
抽出し、この歪成分△を低域フィルタ６を介して電圧制
御発振器２に制御電圧として与えている。On the other hand, the input wave X is passed through the limiter amplifier 15 to the multiplier 16
By multiplying both inputs by this multiplier 16, the distortion component △ contained in the output wave kX is extracted as the output signal, and this distortion component △ is passed through the low-pass filter 6 to the voltage controlled oscillator 2. is given as a control voltage.
この実施例の動作が以下に説明される。いま。The operation of this embodiment will be explained below. now.
加算器ｌからの合成波Ｂが定振幅波となっていなかった
場合、増幅器４は非線形増幅器であるため、その増幅出
力ｋＢは歪むことが予想され、この結果、最終的な逆相
ハイブリッド回路７からの出力波ｋＡも歪成分を含むこ
とになる。これは言い換えれば、電圧制御発振器２によ
る位相調整が的確に行われて合成波Ｂが定振幅波になっ
ている場合には、逆相ハイブリッド回路７から出力され
る出力波ｋＸに含まれる歪成分を最小化できるというこ
とである。If the composite wave B from the adder l is not a constant amplitude wave, since the amplifier 4 is a nonlinear amplifier, its amplified output kB is expected to be distorted, and as a result, the final anti-phase hybrid circuit 7 The output wave kA from will also include distortion components. In other words, if the phase adjustment by the voltage controlled oscillator 2 is performed accurately and the composite wave B is a constant amplitude wave, the distortion component included in the output wave kX output from the anti-phase hybrid circuit 7 This means that it can be minimized.
そこで、この出力波ｋ Ｘの歪成分を前述の減衰器１３
．リミッタ増幅器１４，１５．乗算器１６からなる回路
で抽出し、これを電圧制御発振器２に制御電圧として与
え、この歪成分が無くなる方向に電圧制御発振器２の周
波数制御（位相側＃）を行う。これにより出力波ｋＸに
おける歪成分が居小化された時には、定振幅波Ａと合成
波Ｂの位相ａ？Ｉが最適な状態にあり合成波Ｂは定振幅
波となっていると推定することができる。Therefore, the distortion component of this output wave k
．． Limiter amplifiers 14, 15. This is extracted by a circuit consisting of a multiplier 16, and applied as a control voltage to the voltage controlled oscillator 2, and the frequency of the voltage controlled oscillator 2 is controlled (on the phase side #) in a direction in which this distortion component is eliminated. As a result, when the distortion component in the output wave kX is reduced, the phase a? of the constant amplitude wave A and the composite wave B? It can be estimated that I is in an optimal state and the composite wave B is a constant amplitude wave.
なお、この実施例ではリミッタ増幅１４．１５を用いて
いるのは、リミッタ増幅器１５は入力波Ｘが小さくなっ
た時に回路動作が不安定とならないように入力信号をあ
る一定の大きさとするためであり、リミッタ増幅器１４
は帰還ループ側の位相関係をリミッタ増幅器１５に合わ
せて調整するためのものである。In this embodiment, limiter amplifiers 14 and 15 are used because the limiter amplifier 15 keeps the input signal at a certain level so that the circuit operation does not become unstable when the input wave X becomes small. Yes, limiter amplifier 14
is for adjusting the phase relationship on the feedback loop side in accordance with the limiter amplifier 15.
第１１図には本発明の更に他の実施例が示される。この
実施例は第１Ｏ図の実施例と同じ動作原理によるもので
あるが、相違点として、出力波ｋＸから歪成分を抽出す
る回路の構成が異なっている。FIG. 11 shows yet another embodiment of the present invention. This embodiment is based on the same operating principle as the embodiment shown in FIG. 1O, but the difference is in the configuration of the circuit for extracting the distortion component from the output wave kX.
すなわち、この実施例においては、入力波Ｘを二乗回路
１８を介して減算器１９に導き、一方。That is, in this embodiment, the input wave X is guided to the subtracter 19 via the squaring circuit 18;
逆相ハイブリッド回路７の出力波ｋＸをｌ／に減衰器１
３と二乗回路１７を介して減算器１９に導き、この減算
器１９で両人力信号の差分をとることで、出力波ｋＸに
含まれている歪成分を抽出し、これを低域フィルタ６を
介して電圧制御発振器２に制御電圧として与えている。Attenuator 1 converts the output wave kX of the anti-phase hybrid circuit 7 into l/
3 and a squaring circuit 17 to a subtracter 19, and this subtracter 19 takes the difference between the two human power signals to extract the distortion component contained in the output wave kX. It is applied to the voltage controlled oscillator 2 as a control voltage via the voltage controlled oscillator 2.
この実施例回路の動作原理は上記に説明した第１０図回
路のものと同じである。The operating principle of this embodiment circuit is the same as that of the circuit of FIG. 10 described above.
第１２図には本発明の更にまた他の実施例が示される。FIG. 12 shows yet another embodiment of the present invention.
この実施例は、増幅器３または増幅器４の前段にＡＧＣ
（自動利得調整）増幅器２１を設け、制御回路２２で出
力波ｋＸから歪成分を抽出し、この歪成分を電圧制御発
振器２の制御電圧とすると共に、ＡＧＣ増幅器２１の制
御電圧ともしている。In this embodiment, AGC is installed before amplifier 3 or amplifier 4.
(Automatic gain adjustment) An amplifier 21 is provided, a control circuit 22 extracts a distortion component from the output wave kX, and uses this distortion component as a control voltage for the voltage controlled oscillator 2 and also as a control voltage for the AGC amplifier 21.
すなわち、出力波ｋＸの歪成分を抽出する制御回路２２
の構成としては、第１０図に示されるようなリミッタ増
幅器１４，１５．ミキサ１６からなる回路、あるいは第
１１図に示されるような二乗回路１７．１８．減算器１
９からなる回路が利用できる。That is, the control circuit 22 extracts the distortion component of the output wave kX.
The configuration includes limiter amplifiers 14, 15 . . . as shown in FIG. A circuit consisting of a mixer 16, or a squaring circuit 17, 18, as shown in FIG. Subtractor 1
A circuit consisting of 9 is available.
この制御回路２２の歪抽出出力は二分岐されて、一方は
低域フィルタ６を介して電圧制御発振器２に位相誤差を
修正するための制御電圧として供給され、他方は低域フ
ィルタ２３を介してＡＧＣ増幅記２Ｉに振幅誤差を修正
するための制御電圧として供給される。The distortion extraction output of the control circuit 22 is branched into two branches, one of which is supplied to the voltage controlled oscillator 2 as a control voltage for correcting phase errors via the low-pass filter 6, and the other is supplied to the voltage-controlled oscillator 2 via the low-pass filter 23. It is supplied to the AGC amplifier 2I as a control voltage for correcting amplitude errors.
この第１２図の実施例回路は、前述の電圧制御発振器２
による位相調整だけでは定振幅波Ａ、 Ｂにおける振幅
誤差を除去しきれない場合に最終的な出力波ｋＸに歪成
分が生じるので、この振幅誤差をＡ Ｇ Ｃ増幅器２１
で取り除くものであり、出力波ｋＸに生じた歪成分を制
御回路２２で抽出して１この歪成分が無くなるように、
電圧制御発振器２の発振周波数だけでなく、ＡＧＣ増幅
器２１により定振幅波Ａ（または定振幅波Ｂ）の振幅も
調整するようにしたものである。This embodiment circuit of FIG. 12 is based on the voltage controlled oscillator 2 described above.
If the amplitude error in the constant amplitude waves A and B cannot be removed by phase adjustment alone, a distortion component will occur in the final output wave kX.
The distortion component generated in the output wave kX is extracted by the control circuit 22 so that this distortion component is eliminated.
Not only the oscillation frequency of the voltage controlled oscillator 2 but also the amplitude of the constant amplitude wave A (or the constant amplitude wave B) is adjusted by the AGC amplifier 21.
このように９位相誤差と振幅誤差の調整を共に行うこと
により出力波ｋＸに生じる歪を一層低減させることがで
きる。In this way, by adjusting both the phase error and the amplitude error, distortion occurring in the output wave kX can be further reduced.
第１３図には、上述の第１２図の実施例を原理を用いた
本発明の更に他の実施例が示される。すなわち、この実
施例では、前述の第５図および第９図の実施例と同様に
、制御回路２４により定振幅波Ａ、Ｂの振幅差に応じた
値Ａ ２ Ｂ ２を求め、これを電圧制御発振器２の
制ｍ電圧として用いており、一方、ＡＧＣ増幅器２１の
制御電圧としては、第１２図の実施例と同様に制御回路
２２で出力波ｋＸに含まれる歪成分を抽出して用いてい
る。このように構成することで、電圧制御発振器２によ
る位相誤差の調整とＡＧＣ増幅器２１による振幅差の調
整とを別々に行っている。FIG. 13 shows still another embodiment of the present invention using the principle of the embodiment of FIG. 12 described above. That is, in this embodiment, the control circuit 24 determines the value A 2 B 2 corresponding to the amplitude difference between the constant amplitude waves A and B, and converts this into the voltage. It is used as the control voltage of the controlled oscillator 2, and on the other hand, as the control voltage of the AGC amplifier 21, the distortion component contained in the output wave kX is extracted and used in the control circuit 22 as in the embodiment shown in FIG. There is. With this configuration, the phase error adjustment by the voltage controlled oscillator 2 and the amplitude difference adjustment by the AGC amplifier 21 are performed separately.
第１４図には、同じく第１２図の実施例の原理を用いた
本発明の更に他の実施例が示される。この実施例は、制
御回路２５によりこのＡ Ｇ Ｃ増幅器２１の出力信号
を所定の一定値ｖ０と比較し。FIG. 14 shows yet another embodiment of the invention, also using the principle of the embodiment of FIG. 12. In this embodiment, a control circuit 25 compares the output signal of the AGC amplifier 21 with a predetermined constant value v0.
その差分に応じた値を低域フィルタ２３を介してＡ、
Ｇ Ｃ増幅器の制御電圧に用いている。The value corresponding to the difference is passed through the low-pass filter 23 to A,
It is used for the control voltage of the GC amplifier.
この実施例の動作を説明すると９回路が正常に動作して
いる時にはＡＧＣ増幅器２１の出力はある一定振幅とな
るものであるから、この一定振幅値を予め定めておいて
これを一定値■。とじ、ＡＧＣ増幅器２１の出力がこの
一定値■。どなるようにＡＧＣ増幅器２１の利得を制御
することで。To explain the operation of this embodiment, when the 9 circuits are operating normally, the output of the AGC amplifier 21 has a certain constant amplitude, so this constant amplitude value is determined in advance and is set to the constant value (2). At the end, the output of the AGC amplifier 21 is at this constant value ■. By controlling the gain of the AGC amplifier 21.
振幅誤差をなくすものである。This eliminates amplitude errors.
［発明の効果］本発明によれば、定振幅波を発生するための回路を小型
、小電力、高速動作のアナログ回路で構成することがで
きるので、ディジタル信号処理回路を用いていた従来の
定振幅波合成形増幅器に比べて、定振幅波合成形増幅器
の一層の低消費電力化、小型化および高速動作化を図る
ことができる。[Effects of the Invention] According to the present invention, a circuit for generating a constant amplitude wave can be configured with a small, low power, high-speed operation analog circuit, so that the conventional constant amplitude wave generating circuit that uses a digital signal processing circuit can be used. Compared to amplitude wave synthesis amplifiers, the constant amplitude wave synthesis amplifier can achieve lower power consumption, smaller size, and faster operation.
第１図、第２区、第３図、第４図はそれぞれ本発明に係
る定振幅波合成形増幅器の原理説明図。第５図は本発明の一実施例としての定振幅波合成形増幅
器を示すブロック図。第６図、第７図、第８図は実施例回路の動作説明図。第９図、第ｉｏ図、第１１図、第１２図、第１３図、第
１４図はそれぞれ本発明の他の実施例を示すブロック図
、および第１５図は従来の定振幅波合成形増幅器を示すブロック
図である。図において。■・・・加算器２・・・電圧制御発振器３．４・・・非線形増幅器５・・・演算回路６．２３・・・低域フィルタ７・・・逆相ハイブリッド回路７８・・・終端抵抗器９・・・出力端子１０・・・入力端子１１・・合成回路１２．１６・・・乗算器（ミキサ）１３・・・ｉ／に減衰器１４．１５・・・リミッタ増幅器１７．１８・・・二乗回路１９・・・減算器２１・・・ＡＧＣ増幅器２２・・・制御回路本発明Ｆ係ゐ原理説明図第１図第２の定橡１ｓ浪、本発明（９係ろ厘、理説明口第２図り↓く瀉ギ丘 帆 の ９ン、 凭イダ１第５図＊うヤＬ佼ｉ の ルリイ勺＝ｓＬ日月吹党４列の動イ
乍瀞り明第６図第７図丈銅１イタ１１ のａ Ａ）丁名労−ロ目第８図４−Ｌ ａ’Ａ ’）ｌｔｅ−” ＊ ｅｉ、ｐ＋第９図ヰ唖た明の杷の実」瞥棹１第１Ｑ図ノド発明の杷の大を秤１第１２因本発明の杷カ実恵Δ夕１１FIG. 1, Section 2, FIG. 3, and FIG. 4 are explanatory diagrams of the principle of a constant amplitude wave synthesis amplifier according to the present invention, respectively. FIG. 5 is a block diagram showing a constant amplitude wave synthesis amplifier as an embodiment of the present invention. FIG. 6, FIG. 7, and FIG. 8 are operation explanatory diagrams of the embodiment circuit. FIG. 9, FIG. io, FIG. 11, FIG. 12, FIG. 13, and FIG. 14 are block diagrams showing other embodiments of the present invention, and FIG. 15 is a conventional constant amplitude wave synthesis amplifier. FIG. In fig. ■...Adder 2...Voltage controlled oscillator 3.4...Nonlinear amplifier 5...Arithmetic circuit 6.23...Low pass filter 7...Negative phase hybrid circuit 7 8...Terminal Resistor 9... Output terminal 10... Input terminal 11... Combining circuit 12.16... Multiplier (mixer) 13... Attenuator 14.15... Limiter amplifier 17.18 ...Squaring circuit 19...Subtractor 21...AGC amplifier 22...Control circuit Principle explanatory diagram according to the present invention Explanation mouth No. 2 Diagram ↓ Kusagi Hill Sails 9th Figure 5 7 Figure length bronze 1 Ita 11 a A) Ding Meiro - B Me 8 Figure 4 - L a'A ') lte -" * ei, p + 9th Figure 1 Figure 1Q Weighing the size of the loquat of the invention 1 Factor 12 The loquat of the invention ΔYu 11
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2027700AJP2758683B2 (en) | 1990-02-07 | 1990-02-07 | Constant amplitude wave synthesis type amplifier |
| CA 2035845CA2035845C (en) | 1990-02-07 | 1991-02-06 | Constant-amplitude wave combination type amplifier |
| EP19910101602EP0443368B1 (en) | 1990-02-07 | 1991-02-06 | Constant-amplitude wave combination type amplifier |
| EP19940117669EP0644650B1 (en) | 1990-02-07 | 1991-02-06 | Constant-amplitude wave combination type amplifier |
| EP94117670AEP0635934B1 (en) | 1990-02-07 | 1991-02-06 | Constant-amplitude wave combination type amplifier |
| DE1991630296DE69130296T2 (en) | 1990-02-07 | 1991-02-06 | Compound waves amplifier with constant amplitude |
| DE1991620312DE69120312T2 (en) | 1990-02-07 | 1991-02-06 | Combined waves amplifier with constant amplitude |
| DE1991630366DE69130366T2 (en) | 1990-02-07 | 1991-02-06 | Combined waves amplifier with constant amplitude |
| DE1991630181DE69130181T2 (en) | 1990-02-07 | 1991-02-06 | Combined waves amplifier with constant amplitude |
| EP94117668AEP0635933B1 (en) | 1990-02-07 | 1991-02-06 | Constant-amplitude wave combination type amplifier |
| US07/969,522US5287069A (en) | 1990-02-07 | 1992-10-30 | Constant-amplitude wave combination type amplifier |
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP2027700AJP2758683B2 (en) | 1990-02-07 | 1990-02-07 | Constant amplitude wave synthesis type amplifier |
| Publication Number | Publication Date |
|---|---|
| JPH03232307Atrue JPH03232307A (en) | 1991-10-16 |
| JP2758683B2 JP2758683B2 (en) | 1998-05-28 |
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP2027700AExpired - LifetimeJP2758683B2 (en) | 1990-02-07 | 1990-02-07 | Constant amplitude wave synthesis type amplifier |
| Country | Link |
|---|---|
| JP (1) | JP2758683B2 (en) |
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| Publication number | Publication date |
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| JP2758683B2 (en) | 1998-05-28 |
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