BACKGROUND OF THE INVENTIONI. Field of the InventionThe present invention relates to communications. More particularly, the present invention relates to a novel and improved method and apparatus for high data rate CDMA wireless communication.
II. Description of the Related ArtWireless communication systems including cellular, satellite and point to point communication systems use a wireless link comprised of a modulated radio frequency (RF) signal to transmit data between two systems. The use of a wireless link is desirable for a variety of reasons including increased mobility and reduced infrastructure requirements when compared to wire line communication systems. One drawback of using a wireless link is the limited amount of communication capacity that results from the limited amount of available RF bandwidth. This limited communication capacity is in contrast to wire based communication systems where additional capacity can be added by installing additional wire line connections.
Recognizing the limited nature of RF bandwidth, various signal processing techniques have been developed for increasing the efficiency with which wireless communication systems utilize the available RF bandwidth. One widely accepted example of such a bandwidth efficient signal processing technique is the IS-95 over the air interface standard and its derivatives such as IS-95-A (referred to hereafter collectively as the IS-95 standard) promulgated by the telecommunication industry association (TIA) and used primarily within cellular telecommunications systems. The 15-95 standard incorporates code division multiple access (CDMA) signal modulation techniques to conduct multiple communications simultaneously over the same RF bandwidth. When combined with comprehensive power control, conducting multiple communications over the same bandwidth increases the total number of calls and other communications that can be conducted in a wireless communication system by, among other things, increasing the frequency reuse in comparison to other wireless telecommunication technologies. The use of CDMA techniques in a multiple access communication system is disclosed in U.S. Patent No. 4,901,307, entitled "SPREAD SPECTRUM COMMUNICATION SYSTEM USING SATELLITE OR TERRESTRIAL REPEATERS", and U.S. Patent No. 5,103,459, entitled "SYSTEM AND METHOD FOR GENERATING SIGNAL WAVEFORMS IN A CDMA CELLULAR TELEPHONE SYSTEM", both of which are assigned to the assignee of the present invention and incorporated by reference herein.
Fig. 1 provides a highly simplified illustration of a cellular telephone system configured in accordance with the use of the IS-95 standard. During operation, a set of subscriber units10a - d conduct wireless communication by establishing one or more RF interfaces with one or more base stations12a - d using CDMA modulated RF signals. Each RF interface between a base station12 and a subscriber unit10 is comprised of a forward link signal transmitted from the base station12, and a reverse link signal transmitted from the subscriber unit. Using these RF interfaces, a communication with another user is generally conducted by way of mobile telephone switching office (MTSO)14 and public switch telephone network (PSTN)16. The links between base stations12, MTSO14 and PSTN16 are usually formed via wire line connections, although the use of additional RF or microwave links is also known.
In accordance with the IS-95 standard each subscriber unit 10 transmits user data via a single channel, non-coherent, reverse link signal at a maximum data rate of 9.6 or 14.4 kbits/sec depending on which rate set from a set of rate sets is selected. A non-coherent link is one in which phase information is not utilized by the received system. A coherent link is one in which the receiver exploits knowledge of the carrier signals phase during processing. The phase information typically takes the form of a pilot signal, but can also be estimated from the data transmitted. The IS-95 standard calls for a set of sixty four Walsh codes, each comprised of sixty four chips, to be used for the forward link.
The use of a single channel, non-coherent, reverse link signal having a maximum data rate of 9.6 of 14.4 kbits/sec as specified by IS-95 is well suited for a wireless cellular telephone system in which the typical communication involves the transmission of digitized voice or lower rate digital data such a facsimile. A non-coherent reverse link was selected because, in a system in which up to 80 subscriber units10 may communicate with a base station12 for each 1.2288 MHz of bandwidth allocated, providing the necessary pilot data in the transmission from each subscriber unit10 would substantially increase the degree to which a set of subscriber units10 interfere with one another. Also, at data rates of 9.6 or 14.4 kbits/sec, the ratio of the transmit power of any pilot data to the user data would be significant, and therefore also increase inter-subscriber unit interference. The use of a single channel reverse link signal was chosen because engaging in only one type of communication at a time is consistent with the use of wireline telephones, the paradigm on which current wireless cellular communications is based. Also, the complexity of processing a single channel is less than that associated with processing multiple channels.
As digital communications progress, the demand for wireless transmission of data for applications such as interactive file browsing and video teleconferencing is anticipated to increase substantially. This increase will transform the way in which wireless communications systems are used, and the conditions under which the associated RF interfaces are conducted. In particular, data will be transmitted at higher maximum rates and with a greater variety of possible rates. Also, more reliable transmission may become necessary as errors in the transmission of data are less tolerable than errors in the transmission of audio information. Additionally, the increased number of data types will create a need to transmit multiple types of data simultaneously. For example, it may be necessary to exchange a data file while maintaining an audio or video interface. Also, as the rate of transmission from a subscriber unit increases the number of subscriber units10 communicating with a base station12 per amount of RF bandwidth will decrease, as the higher data transmission rates will cause the data processing capacity of the base station to be reached with fewer subscriber units10.
Furthermore, attention is drawn to the document WO-95-03652, which describes a method and system for allocating a set of orthogonal pseudo noise code sequences of variable length among user channels operative at different data rates in a spread spectrum communication system. Pseudo noise code sequences are constructed to provide orthogonality between users to reduce mutual interference thereby allowing higher capacity and better link performance. The pseudo-noise code sequences assignments are made on the basis of channel data rates in a manner which results in improved utilization of the available frequency spectrum.
Attention is also drawn to the document WO-95-23464, which describes a communication system that time division multiplexes the use of spreading codes to provide increased or high data rate traffic. The communication system accepts information from at least two users and codes each user's information utilizing error correction coders. The coded information is then time-multiplexed by a multiplexer into timeslots.,The output of the multiplex is spread by a common spreading (Walsh) code scrambled with a pseudo-noise sequence, and conveyed to a modulator for transmission. In this manner, information from two users may be transmitted utilizing only a single spreading (Walsh) code.
In some instances, the current IS-95 reverse link may not be ideally suited for all these changes. Therefore, the present invention is related to providing a higher data rate, bandwidth efficient, COMA interface over which multiple types of communication can be performed.
In accordance with the present invention a method for generating modulated data, as set forth in claim 1, and an apparatus for generating modulated data for transmission, as set forth in claim 5, are provided. Preferred embodiments of the invention are claimed in the dependent claims.
The use of the orthogonal codes provides interference suppression while still allowing extensive error correction coding and repetition for time diversity to overcome the Raleigh fading commonly experienced in terrestrial wireless systems. In practise the orthogonal (sub-channel) codes are comprised of Walsh codes orthogonal to one another, and four chips in duration. The use of four sub-channels is preferred as it allows shorter orthogonal codes to be used, however, the use of a greater number of channels and therefore longer codes is consistent with the invention.
Additionally pilot data is transmitted via a first one of the transmit channels and a user data or power control data, is transmitted via a second transmit channel. The remaining transmit channels are used for transmitting non-specified digital data including user data or signaling, data, or both. Furthermore one of the two non-specified transmit channels is configured for BPSK modulation and transmission over the quadrature-phase term.
BRIEF DESCRIPTION OF THE DRAWINGSThe features, objects, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings in which like reference characters identify correspondingly throughout and wherein:
- Fig. 1 is a block diagram of cellular telephone system;
- Fig. 2 is a block diagram of a subscriber unit and base station configured in accordance with the exemplary embodiment of the invention;
- Fig. 3 is a block diagram of a BPSK channel encoder and a QPSK channel encoder configured in accordance with the exemplary embodiment of the invention;
- Fig. 4 is a block diagram of a transmit signal processing system configured in accordance with the exemplary embodiment of the invention;
- Fig. 5 is a block diagram of a receive processing system configured in accordance with the exemplary embodiment of the invention;
- Fig. 6 is a block diagram of a finger processing system configured in accordance with one embodiment of the invention;
- Fig. 7 is a block diagram of a BPSK channel decoder and a QPSK channel decoder configured in accordance with the exemplary embodiment of the invention; and
- Fig. 8 is a block diagram of a transmit signal processing system configured in accordance with a second exemplary embodiment of the invention; and
- Fig. 9 is a block diagram of a finger processing system configured in accordance with one embodiment of the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTSA novel and improved method and apparatus for high rate CDMA wireless communication is described in the context of the reverse link transmission portion of a cellular telecommunications system. While the invention is particularly adapted for use within the multipoint-to-point reverse link transmission of a cellular telephone system, the present invention is equally applicable to forward link transmissions. In addition, many other wireless communication systems will benefit by incorporation of the invention, including satellite based wireless communication systems, point to point wireless communication systems, and systems transmitting radio frequency signals via the use of co-axial or other broadband cables.
Fig. 2 is a block diagram of receive and transmit systems configured as a subscriber unit 100 and a base station120 in accordance with one embodiment of the invention. A first set of data (BPSK data) is received by BPSK channel encoder103, which generates a code symbol stream configured for performing BPSK modulation that is received by modulator104. A second set of data (QPSK data) is received by QPSK channel encoder102, which generates a code symbol stream configured for performing QPSK modulation that is also received by modulator104. Modulator104 also receives power control data and pilot data, which are modulated along with the BPSK and QPSK encoded data in accordance with code division multiple access (CDMA) techniques to generate a set of modulation symbols received by RF processing system106. RF processing system106 filters and upconverts the set of modulation symbols to a carrier frequency for transmission to the base station120 using antenna108. While only one subscriber unit100 is shown, multiple subscriber units communicate with base station120 in the preferred embodiment.
Within base station120, RF processing system122 receives the transmitted RF signals by way of antenna121 and performs bandpass filtering, downconversion to baseband, and digitization. Demodulator124 receives the digitized signals and performs demodulation in accordance with CDMA techniques to produce power control, BPSK, and QPSK soft decision data. BPSK channel decoder128 decodes the BPSK soft decision data received from demodulator124 to yield a best estimate of the BPSK data, and QPSK channel decoder126 decodes the QPSK soft decision data received by demodulator124 to produce a best estimate of the QPSK data. The best estimate of first and second set of data is then available for further processing or forwarding to a next destination, and the received power control data used either directly, or after decoding, to adjust the transmit power of the forward link channel used to transmit data to subscriber unit100.
Fig. 3 is a block diagram of BPSK channel encoder103 and QPSK channel encoder102 when configured in accordance with the exemplary embodiment of the invention. Within BPSK channel encoder103 the BPSK data is received by CRC check sum generator130 which generates a check sum for each 20 ms frame of the first set of data. The frame of data along with the CRC check sum is received by tail bit generator132 which appends tail bits comprised of eight logic zeros at the end of each frame to provide a known state at the end of the decoding process. The frame including the code tail bits and CRC check sum is then received by convolutional encoder134 which performs, constraint length (K) 9, rate (R) 1/4 convolutional encoding thereby generating code symbols at a rate four times the encoder input rate (ER). In the alternative embodiment of the invention, other encoding rates are performed including rate 1/2, but the use of rate 1/4 is preferred due to its optimal complexity-performance characteristics. Block interleaver136 performs bit interleaving on the code symbols to provide time diversity for more reliable transmission in fast fading environments. The resulting interleaved symbols are received by variable starting point repeater138, which repeats the interleaved symbol sequence a sufficient number of times NR to provide a constant rate symbol stream, which corresponds to outputting frames having a constant number of symbols. Repeating the symbol sequence also increases the time diversity of the data to overcome fading. In the exemplary embodiment, the constant number of symbols is equal to 6,144 symbols for each frame making the symbol rate 307.2 kilosymbols per second (ksps). Also, repeater138 uses a different starting point to begin the repetition for each symbol sequence. When the value of NR necessary to generate 6,144 symbols per frame is not an integer, the final repetition is only performed for a portion of the symbol sequence. The resulting set of repeated symbols are received by BPSK mapper139 which generates a BPSK code symbol stream (BPSK) of +1 and -1 values for performing BPSK modulation. In an alternative embodiment of the invention repeater138 is placed before block interleaver136 so that block interleaver136 receives the same number of symbols for each frame.
Within QPSK channel encoder102 the QPSK data is received by CRC check sum generator140 which generates a check sum for each 20 ms frame. The frame including the CRC check sum is received by code tail bits generator142 which appends a set of eight tail bits of logic zeros at the end of the frame. The frame, now including the code tail bits and CRC check sum, is received by convolutional encoder144 which performs K=9, R=1/4 convolutional encoding thereby generating symbols at a rate four times the encoder input rate (ER). Block interleaver146 performs bit interleaving on the symbols and the resulting interleaved symbols are received by variable starting point repeater148. Variable starting point repeater148 repeats the interleaved symbol sequence a sufficient number of times NR using a different starting point within the symbol sequence for each repetition to generate 12,288 symbols for each frame making the code symbol rate 614.4 kilosymbols per second (ksps). When NR is not an integer, the final repetition is performed for only a portion of the symbol sequence. The resulting repeated symbols are received by QPSK mapper149 which generates a QPSK code symbol stream configured for performing QPSK modulation comprised of an in-phase QPSK code symbol stream of +1 and -1 values (QPSKI), and a quadrature-phase QPSK code symbol stream of +1 and -1 values (QPSKQ). In an alternative embodiment of the invention repeater148 is placed before block interleaver146 so that block interleaver146 receives the same number of symbols for each frame.
Fig. 4 is a block diagram of modulator
104 of
Fig. 2 configured in accordance with the exemplary embodiment of the invention. The BPSK symbols from BPSK channel encoder
103 are each modulated by Walsh code W
2 using a multiplier 150b, and the QPSK
I and QPSK
Q symbols from QPSK channel encoder
102 are each modulated with Walsh code W
3 using multipliers
150c and
154d. The power control data (PC) is modulated by Walsh code W
1 using multiplier
150a. Gain adjust
152 receives pilot data (PILOT), which in the preferred embodiment of the invention is comprised of the logic level associated with positive voltage, and adjusts the amplitude according to a gain adjust factor A
0. The PILOT signal provides no user data but rather provides phase and amplitude information to the base station so that it can coherently demodulate the data carried on the remaining sub-channels, and scale the soft-decision output values for combining. Gain adjust
154 adjusts the amplitude of the Walsh code W
1 modulated power control data according to gain adjust factor A
1, and gain adjust
156 adjusts the amplitude of the Walsh code W
2 modulated BPSK channel data according amplification variable A
2. Gain adjusts
158a and
b adjust the amplitude of the in-phase and quadrature-phase Walsh code W
3 modulated QPSK symbols respectively according to gain adjust factor A
3. The four Walsh codes used in the preferred embodiment of the invention are shown in Table I.
Table I.| Walsh Code | Modulation Symbols |
| ++++ |
| +-+- |
| ++ - - |
| + - - + |
It will be apparent to one skilled in the art that the W0 code is effectively no modulation at all, which is consistent with processing of the pilot data shown. The power control data is modulated with the W1 code, the BPSK data with the W2 code, and the QPSK data with the W3 code. Once modulated with the appropriate Walsh code, the pilot, power control data, and BPSK data are transmitted in accordance with BPSK techniques, and the QPSK data (QPSKI and QPSKQ) in accordance with QPSK techniques as described below. It should also be understood that it is not necessary that every orthogonal channel be used, and that the use of only three of the four Walsh codes where only one user channel is provided is employed in an alternative embodiment of the invention.
The use of short orthogonal codes generates fewer chips per symbol, and therefore allows for more extensive coding and repetition when compared to systems incorporating the use of longer Walsh codes. This more extensive coding and repetition provides protection against Raleigh fading which is a major source of error in terrestrial communication systems. The use of other numbers of codes and code lengths is consistent with the present invention, however, the use of a larger set of longer Walsh codes reduces this enhanced protection against fading. The use of four chip codes is considered optimal because four channels provides substantial flexibility for the transmission of various types of data as illustrated below while also maintaining short code length.
Summer160 sums the resulting amplitude adjusted modulation symbols from gain adjusts152, 154, 156 and158a to generate summed modulation symbols161. PN spreading codes PNI and PNQ are spread via multiplication with long code180 using multipliers162a andb. The resulting pseudorandom code provided by multipliers162a and162b are used to modulate the summed modulation symbols161, and gain adjusted quadrature-phase symbols QPSKQ163, via complex multiplication using multipliers164a-d and summers166a andb. The resulting in-phase term XI and quadrature-phase term XQ are then filtered (filtering not shown), and upconverted to the carrier frequency within RF processing system106 shown in a highly simplified form using multipliers168 and an in-phase and a quadrature-phase sinusoid. An offset QPSK upconversion could also be used in an alternative embodiment of the invention. The resulting in-phase and quadrature-phase upconverted signals are summed using summer170 and amplified by master amplifier172 according to master gain adjust AM to generate signal s(t) which is transmitted to base station120. In the preferred embodiment of the invention, the signal is spread and filtered to a 1.2288 MHz bandwidth to remain compatible with the bandwidth of existing CDMA channels.
By providing multiple orthogonal channels over which data may be transmitted, as well as by using variable rate repeaters that reduce the amount of repeating N
R performed in response to high input data rates, the above described method and system of transmit signal processing allows a single subscriber unit or other transmit system to transmit data at a variety of data rates. In particular, by decreasing the rate of repetition N
R performed by variable starting point repeaters
138 or
148 of FIG. 3, an increasingly higher encoder input rate E
R can be sustained. In an alternative embodiment of the invention rate 1/2 convolution encoding is performed with the rate of repetition N
R increased by two. A set of exemplary encoder rates E
R supported by various rates of repetition N
R and encoding rates R equal to 1/4 and 1/2 for the BPSK channel and the QPSK channel are shown in Tables II and III respectively.
Table II. |
| Label | | Encoder Out R=1/4 (bits/frame) | | Encoder Out R=1/2 (bits/frame) | |
| High Rate-72 | 76,800 | 6,144 | 1 | 3,072 | 2 |
| High Rate-64 | 70,400 | 5,632 | 1 1/11 | 2,816 | 2 2/11 |
| 51,200 | 4,096 | 1 1/2 | 2,048 | 3 |
| High Rate-32 | 38,400 | 3,072 | 2 | 1,536 | 4 |
| 25,600 | 2,048 | 3 | 1,024 | 6 |
| RS2-Full Rate | 14,400 | 1,152 | 51/3 | 576 | 10 2/3 |
| RS1-Full Rate | 9,600 | 768 | 8 | 384 | 16 |
| NULL | 850 | 68 | 906/17 | 34 | 180 12/17 |
Table III. |
| Label | | Encoder Out R=1/4 (bits/frame) | | Encoder Out R=1/2 (bits/frame) | |
| 153,600 | 12,288 | 1 | 6,144 | 2 |
| High Rate-72 | 76,800 | 6,144 | 2 | 3,072 | 4 |
| High Rate-64 | 70,400 | 5,632 | 2 2/11 | 2,816 | 4 4/11 |
| 51,200 | 4,096 | 3 | 2,048 | 6 |
| High Rate-32 | 38,400 | 3,072 | 4 | 1,536 | 8 |
| 25,600 | 2,048 | 6 | 1,024 | 12 |
| RS2-Full Rate | 14,400 | 1,152 | 102/3 | 576 | 21 1/3 |
| RS1-Full Rate | 9,600 | 768 | 16 | 384 | 32 |
| NULL | 850 | 68 | 180 12/17 | 34 | 361 7/17 |
Tables II and III show that by adjusting the number of sequence repetitions NR, a wide variety of data rates can be supported including high data rates, as the encoder input rate ER corresponds to the data transmission rate minus a constant necessary for the transmission of CRC, code tail bits and any other overhead information. As also shown by tables II and III, QPSK modulation may also be used to increase the data transmission rate. Rates expected to be used commonly are provided labels such as "High Rate-72" and "High Rate-32." Those rates noted as High Rate-72, High Rate-64, and High Rate-32 have traffic rates of 72, 64 and 32 kbps respectively, plus multiplexed in signaling and other control data with rates of 3.6, 5.2, and 5.2 kbps respectively, in the exemplary embodiment of the invention. Rates RS1-Full Rate and RS2-Full Rate correspond to rates used in IS-95 compliant communication systems, and therefore are also expected to receive substantial use for purposes of compatibility. The null rate is the transmission of a single bit and is used to indicate a frame erasure, which is also part of the IS-95 standard.
The data transmission rate may also be increased by simultaneously transmitting data over two or more of the multiple orthogonal channels performed either in addition to, or instead of, increasing the transmission rate via reduction of the repetition rate NR. For example, a multiplexer (not shown) could split a single data source into a multiple data sources to be transmitted over multiple data sub-channels. Thus, the total transmit rate can be increased via either transmission over a particular channel at higher rates, or multiple transmission performed simultaneously over multiple channels, or both, until the signal processing capability of the receive system is exceeded and the error rate becomes unacceptable, or the maximum transmit power of the of the transmit system power is reached.
Providing multiple channels also enhances flexibility in the transmission of different types of data. For example, the BPSK channel may be designated for voice information and the QPSK channel designated for transmission of digital data. This embodiment could be more generalized by designating one channel for transmission of time sensitive data such as voice at a lower data rate, and designating the other channel for transmission of less time sensitive data such as digital files. In this embodiment interleaving could be performed in larger blocks for the less time sensitive data to further increase time diversity. In another embodiment of the invention, the BPSK channel performs the primary transmission of data, and the QPSK channel performs overflow transmission. The use of orthogonal Walsh codes eliminates or substantially reduces any interference among the set of channels transmitted from a subscriber unit, and thus minimizes the transmit energy necessary for their successful reception at the base station.
To increase the processing capability at the receive system, and therefore increase the extent to which the higher transmission capability of the subscriber unit may be utilized, pilot data is also transmitted via one of the orthogonal channels. Using the pilot data, coherent processing can be performed at the receive system by determining and removing the phase offset of the reverse link signal. Also, the pilot data can be used to optimally weigh multipath signals received with different time delays before being combined in a rake receiver. Once the phase offset is removed, and the multipath signals properly weighted, the multipath signals can be combined decreasing the power at which the reverse link signal must be received for proper processing. This decrease in the required receive power allows greater transmissions rates to be processed successfully, or conversely, the interference between a set of reverse link signals to be decreased. While some additional transmit power is necessary for the transmission of the pilot signal, in the context of higher transmission rates the ratio of pilot channel power to the total reverse link signal power is substantially lower than that associated with lower data rate digital voice data transmission cellular systems. Thus, within a high data rate CDMA system the Eb/N0 gains achieved by the use of a coherent reverse link outweigh the additional power necessary to transmit pilot data from each subscriber unit.
The use of gain adjusts152 - 158 as well as master amplifier172 further increases the degree to which the high transmission capability of the above described system can be utilized by allowing the transmit system to adapt to various radio channel conditions, transmission rates, and data types. In particular, the transmit power of a channel that is necessary for proper reception may change over time, and with changing conditions, in a manner that is independent of the other orthogonal channels. For example, during the initial acquisition of the reverse link signal the power of the pilot channel may need to be increased to facilitate detection and synchronization at the base station. Once the reverse link signal is acquired, however, the necessary transmit power of the pilot channel would substantially decrease, and would vary depending on various factors including the subscriber units rate of movement. Accordingly, the value of the gain adjust factor A0 would be increased during signal acquisition, and then reduced during an ongoing communication. In another example, when information more tolerable of error is being transmitted via the forward link, or the environment in which the forward link transmission is taking place is not prone to fade conditions, the gain adjust factor A1 may be reduced as the need to transmit power control data with a low error rate decreases. In one embodiment of the invention, whenever power control adjustment is not necessary the gain adjust factor A1 is reduced to zero.
In another embodiment of the invention, the ability to gain adjust each orthogonal channel or the entire reverse link signal is further exploited by allowing the base station120 or other receive system to alter the gain adjust of a channel, or of the entire reverse link signal, via the use of power control commands transmitted via the forward link signal. In particular, the base station may transmit power control information requesting the transmit power of a particular channel or the entire reverse link signal be adjusted. This is advantageous in many instances including when two types of data having different sensitivity to error, such as digitized voice and digital data, are being transmitted via the BPSK and QPSK channels. In this case, the base station120 would establish different target error rates for the two associated channels. If the actual error rate of a channel exceeded the target error rate, the base station would instruct the subscriber unit to reduce the gain adjust of that channel until the actual error rate reached the target error rate. This would eventually lead to the gain adjust factor of one channel being increased relative to the other. That is, the gain adjust factor associated with the more error sensitive data would be increased relative to the gain adjust factor associated with the less sensitive data. In other instances, the transmit power of the entire reverse link may require adjustment due to fade conditions or movement of the subscriber unit100. In these instances, the base station120 can do so via transmission of a single power control command.
Thus, by allowing the gain of the four orthogonal channels to be adjusted independently, as well as in conjunction with one another, the total transmit power of the reverse link signal can be kept at the minimum necessary for successful transmission of each data type, whether it is pilot data, power control data, signaling data, or different types of user data. Furthermore, successful transmission can be defined differently for each data type. Transmitting with the minimum amount of power necessary allows the greatest amount of data to be transmitted to the base station given the finite transmit power capability of a subscriber unit, and also reduces the interfere between subscriber units. This reduction in interference increases the total communication capacity of the entire CDMA wireless cellular system.
The power control channel used in the reverse link signal allows the subscriber unit to transmit power control information to the base station at a variety of rates including a rate of 800 power control bits per second. In the preferred embodiment of the invention, a power control bit instructs the base station to increase or decrease the transmit power of the forward link traffic channel being used to transmit information to the subscriber unit. While it is generally useful to have rapid power control within a CDMA system, it is especially useful in the context of higher data rate communications involving data transmission, because digital data is more sensitive to errors, and the high transmission causes substantial amounts of data to be lost during even brief fade conditions. Given that a high speed reverse link transmission is likely to be accompanied by a high speed forward link transmission, providing for the rapid transmission of power control over the reverse link further facilitates high speed communications within CDMA wireless telecommunications systems.
In an alternative exemplary embodiment of the invention a set of encoder input rates E
R defined by the particular N
R are used to transmit a particular type of data. That is, data may be transmitted at a maximum encoder input rate E
R or at a set of lower encoder input rates E
R, with the associated N
R adjusted accordingly. In the preferred implementation of this embodiment, the maximum rates corresponds to the maximum rates used in IS-95 compliant wireless communication system, referred to above with respect to Tables II and III as RS1-Ful Rate and RS2-Full Rate, and each lower rate is approximately one half the next higher rate, creating a set of rates comprised of a full rate, a half rate, a quarter rate, and an eighth rate. The lower data rates are preferable generated by increasing the symbol repetition rate N
R with value of N
R for rate set one and rate set two in a BPSK channel provided in Table IV.
Table IV. |
| Label | | Encoder Out R=1/4 (bits/frame) | | Encoder Out R=1/2 (bits/frame) | |
| RS2-Full Rate | 14,400 | 1,152 | 5 1/3 | 576 | 10 2/3 |
| RS2-Halt Rate | 7,200 | 576 | 10 2/3 | 288 | 21 1/3 |
| RS2-Quater Rate | 3,600 | 288 | 21 1/3 | 144 | 42 2/3 |
| RS2-Eigth Rate | 1,900 | 152 | 40 8/19 | 76 | 80 16/19 |
| RS1-Full Rate | 9,600 | 768 | 8 | 384 | 16 |
| RS1-Half Rate | 4,800 | 384 | 16 | 192 | 32 |
| RS1 -Quater Rate | 2,800 | 224 | 27 3/7 | 112 | 54 6/7 |
| RS1-Eigth Rate | 1,600 | 128 | 48 | 64 | 96 |
| NULL | 850 | 68 | 90 6/17 | 34 | 180 12/17 |
The repetition rates for a QPSK channel is twice that for the BPSK channel.
In accordance with the exemplary embodiment of the invention, when the data rate of a frame changes with respect to the previous frame the transmit power of the frame is adjusted according to the change in transmission rate. That is, when a lower rate frame is transmitted after a higher rate frame, the transmit power of the transmit channel over which the frame is being transmitted is reduced for the lower rate frame in proportion to the reduction in rate, and vice versa. For example, if the transmit power of a channel during the transmission of a full rate frame is transmit power T, the transmit power during the subsequent transmission of a half rate frame is transmit power T/2. The reduction is transmit power is preferably performed by reducing the transmit power for the entire duration of the frame, but may also be performed by reducing the transmit duty cycle such that some redundant information is "blanked out." In either case, the transmit power adjustment takes place in combination with a closed loop power control mechanism whereby the transmit power is further adjusted in response to power control data transmitted from the base station.
Fig. 5 is a block diagram of RF processing system122 and demodulator124 ofFig. 2 configured in accordance with the exemplary embodiment of the invention. Multipliers180a and180b downconvert the signals received from antenna121 with an in-phase sinusoid and a quadrature phase sinusoid producing in-phase receive samples RI and quadrature-phase receive samples RQ receptively. It should be understood that RF processing system122 is shown in a highly simplified form, and that the signals are also match filtered and digitized (not shown) in accordance with widely known techniques. Receive samples RI and RQ are then applied to finger demodulators182 within demodulator124. Each finger demodulator182 processes an instance of the reverse link signal transmitted by subscriber unit100, if such an instance is available, where each instance of the reverse link signal is generated via multipath phenomenon. While three finger demodulators are shown, the use of alternative numbers of finger processors are consistent with the invention including the use of a single finger demodulator182. Each finger demodulator182 produces a set of soft decision data comprised of power control data, BPSK data, and QPSKI data and QPSKQ data. Each set of soft decision data is also time adjusted within the corresponding finger demodulator182, although time adjustment could be performed within combiner184 in an alternative embodiment of the invention. Combiner184 then sums the sets of soft decision data received from finger demodulators182 to yield a single instance of power control, BPSK, QPSKI and QPSKQ soft decision data.
Fig. 6 is block diagram a finger demodulator182 ofFig. 5 configured in accordance with the exemplary embodiment of the invention. The RI and RQ receive samples are first time adjusted using time adjust190 in accordance with the amount of delay introduced by the transmission path of the particular instance of the reverse link signal being processed. Long code200 is mixed with pseudorandom spreading codes PNI and PNQ using multipliers201, and the complex conjugate of the resulting long code modulated PNI and PNQ spreading codes are complex multiplied with the time adjusted RI and RQ receive samples using multipliers202 and summers204 yielding terms XI and XQ. Three separate instances of the XI and XQ terms are then demodulated using the Walsh codes W1, W2 and W3 respectively, and the resulting Walsh demodulated data is summed over four demodulation chips using 4 to 1 summers212. A fourth instance of the XI and XQ data is summed over four demodulation chips using summers208, and then filtered using pilot filters214. In the preferred embodiment of the invention pilot filter214 performs averaging over a series of summations performed by summers208, but other filtering techniques will be apparent to one skilled in the art. The filtered in-phase and quadrature-phase pilot signals are used to phase rotate and scale the W1, and W2 Walsh code demodulated data in accordance with BPSK modulated data via complex conjugate multiplication using multipliers216 and adders217 yielding soft decision power control and BPSK data. The W3 Walsh code modulated data is phase rotated using the in-phase and quadrature-phase filtered pilot signals in accordance with QPSK modulated data using multipliers218 and adders220, yielding soft decision QPSK data. The soft decision power control data is summed over384 modulation symbols by384 to 1 summer 222 yielding power control soft decision data. The phase rotated W2 Walsh code modulated data, the W3 Walsh code modulated data, and the power control soft decision data are then made available for combining. In an alternative embodiment of the invention, encoding and decoding is performed on the power control data as well.
In addition to providing phase information the pilot may also be used within the receive system to facilitate time tracking. Time tracking is performed by also processing the received data at one sample time before (early), and one sample time after (late), the present receive sample being processed. To determine the time that most closely matches the actual arrival time, the amplitude of the pilot channel at the early and late sample time can be compared with the amplitude at the present sample time to determine that which is greatest. If the signal at one of the adjacent sample times is greater than that at the present sample time, the timing can be adjusted so that the best demodulation results are obtained.
FIG. 7 is a block diagram of BPSK channel decoder128 and QPSK channel decoder126 (Fig. 2) configured in accordance with the exemplary embodiment of the invention. BPSK soft decision data from combiner184 (Fig. 5) is received by accumulator 240 which stores the first sequence of 6,144/NR, demodulation symbols in the received frame where NR depends on the transmission rate of the BPSK soft decision data as described above, and adds each subsequent set of 6,144/NR demodulated symbols contained in the frame with the corresponding stored accumulated symbols. Block deinterleaver242 deinterleaves the accumulated soft decision data from variable starting point summer240, and Viterbi decoder244 decodes the deinterleaved soft decision data to produce hard decision data as well as CRC check sum results. Within QPSK decoder126 QPSKI and QPSKQ soft decision data from combiner184 (Fig. 5) are demultiplexed into a single soft decision data stream by demux246 and the single soft decision data stream is received by accumulator248 which accumulates every 6,144/NR demodulation symbols where NR depends on the transmission rate of the QPSK data. Block deinterleaver250 deinterleaves the soft decision data from variable starting point summer248, and Viterbi decoder252 decodes the deinterleaved modulation symbols to produce hard decision data as well as CRC check sum results. In the alternative exemplary embodiment described above with respect toFig. 3 in which symbol repetition is performed before interleaving, accumulators240 and248 are placed after block deinterleavers242 and250. In the embodiment of the invention incorporating the use of rate sets, and therefore in which the rate of particular frame is not known, multiple decoders are employed, each operating at a different transmission rate, and then the frame associated with the transmission rate most likely to have been used is selected based on the CRC checksum results. The use of other error checking methods is consistent with the practice of the present invention.
Fig. 8 is a block diagram of modulator104 (Fig. 2) configured in an alternative embodiment of the invention in which a single BPSK data channel is employed. Pilot data is gain adjusted by gain adjust452 in accordance with gain adjust factor A0. Power control data is modulated with Walsh code W1 by multiplier150a and gain adjusted by gain adjust454 in accordance with gain adjust factor A1. The gain adjusted pilot data and power control data are summed by summer460 producing summed data461. BPSK data is modulated with Walsh code W2 by multiplier150b and then gain adjusted using gain adjust456 in accordance with gain adjust factor A2.
In-phase pseudo random spreading code (PNI) and quadrature-phase pseudo random spreading code (PNQ) are both modulated with long code480. The resulting long code modulated PNI and PNQ codes are complex multiplied with the summed data461 and the gain adjusted BPSK data from gain adjust456 using multipliers464a - d and summers466a - b yielding terms XI and XQ. Terms XI and XQ are then upconverted with in-phase and quadrature-phase sinusoids suing multipliers468 and the resulting upconverted signals are summed by summers470 respectively, and amplified by amplifier472 in accordance with amplitude factor AM generating signal s(t).
The embodiment shown inFig. 8 differs from the other embodiments described herein in that the BPSK data is placed in the quadrature-phase channel while the pilot data and power control data are placed in the in-phase channel. In the other embodiments of the invention described herein the BPSK data is placed the in-phase channel along with the pilot data and power control data. Placing the BPSK data in the quadrature-phase channel and the pilot and power control data in the in-phase channel reduces the peak-to-average power ratio of the reverse link signal the phases of the channels are orthogonal causing the magnitude of the sum of the two channels to vary less in response to changing data. This reduces the peek power required to maintain a given average power, and thus reduces the peek-to-average power ratio characteristic of the reverse link signal. This reduction in the peek-to-average power ratio decreases the peek power at which a reverse link signal must be received at the base station in order to sustain a given transmission rate, and therefore increases the distance a subscriber unit having a maximum transmit power may be located from the base station before it is unable to transmit a signal that can received at base station with the necessary peek power. This increase in the distance a subscriber unit may be located from the base station while still being able to transmit a signal that is received with the necessary peek power increases the range at which the subscriber unit is able to successfully conduct communication at any given data rate.
Fig. 9 is a block diagram of finger demodulator182 when configured in accordance with the embodiment of the invention shown inFig. 8. Receive samples RI and RQ are time adjusted by timing adjust290 and the PNI and PNQ codes are multiplied by long code200 using multipliers301. The time adjusted receive samples are then multiplied by the complex conjugate of the PNI and PNQ codes using multipliers302 and summers304 yielding terms XI and XQ. A first and second instance of the XI and XQ terms are demodulated using Walsh code W1 and Walsh code W2 using multipliers310 and the resulting demodulation symbols are summed in sets of four using summers312. A third instance of the XI and XQ terms are summed over four demodulation symbols by summers308 to generate pilot reference data. The pilot reference data is filtered by pilot filters314 and used to phase rotate and scale the summed Walsh code modulated data using multipliers316 and adders320 producing BPSK soft decision data, and after being summed over 384 symbols by 384:1 summer322, soft decision power control data.
Thus, a multi-channel, high rate, CDMA wireless communication system has been described. The description is provided to enable any person skilled in the art to make or use the present invention. The various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without the use of the inventive faculty. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope defined by the claims.