技术领域Technical field
本发明涉及电子信息技术领域，具体涉及一种宽带零中频接收阵列的通道标校与均衡方法。The invention relates to the field of electronic information technology, and in particular to a channel calibration and equalization method for a wideband zero-IF receiving array.
背景技术Background technique
随着数字信号处理技术不断进步和相应处理能力的不断提高，宽带数字阵列以其覆盖频段宽、扫描波束多、设计灵活性高等特点，已经逐步取代模拟阵列天线，在通信、对抗、雷达等电子信息技术领域广泛应用。相比于超外差式中频采样接收阵列，由多通道零中频接收构建的宽带数字阵列具有低成本、低功耗、高集成等特点，是当前宽带数字阵列的主要研究方向。With the continuous advancement of digital signal processing technology and the continuous improvement of corresponding processing capabilities, broadband digital arrays have gradually replaced analog array antennas in electronics such as communications, countermeasures, and radar due to their wide coverage frequency bands, multiple scanning beams, and high design flexibility. Widely used in the field of information technology. Compared with superheterodyne IF sampling receiving arrays, broadband digital arrays constructed from multi-channel zero-IF reception have the characteristics of low cost, low power consumption, and high integration, and are currently the main research direction of broadband digital arrays.
在实际工程中，由于当前器件工艺水平的限制，宽带零中频接收阵列的通道误差主要由通道间频率响应误差和通道内同相(I)和正交(Q)支路不平衡误差组成。通道间频率响应误差主要来自于通道间模拟正交混频前的模拟器件(天线、预选带通滤波器、低噪声放大器等)，而IQ不平衡误差主要取决于模拟正交混频的同相本振与正交本振的非正交性，以及模拟正交混频后的同相支路与正交支路间的频率响应不一致性。相比于通道间频率响应误差，通道内IQ不平衡误差对数字阵列的波束合成性能影响更为严重，它使得零中频接收阵列波束合成后，不仅存在频域镜像分量，也会产生空域镜像分量。这些镜像分量存在将会极大影响接收阵列的整体性能。In actual engineering, due to the limitations of the current device technology level, the channel error of the broadband zero-IF receiving array is mainly composed of the inter-channel frequency response error and the intra-channel in-phase (I) and quadrature (Q) branch imbalance errors. The frequency response error between channels mainly comes from the analog devices (antennas, preselected bandpass filters, low noise amplifiers, etc.) before the analog quadrature mixing between channels, while the IQ imbalance error mainly depends on the in-phase nature of the analog quadrature mixing. The non-orthogonality of the oscillator and the quadrature local oscillator, as well as the frequency response inconsistency between the in-phase branch and the quadrature branch after simulating quadrature mixing. Compared with the inter-channel frequency response error, the intra-channel IQ imbalance error has a more serious impact on the beamforming performance of the digital array. It causes the zero-IF receiving array beamforming not only to have frequency domain image components, but also to produce spatial domain image components. . The existence of these image components will greatly affect the overall performance of the receiving array.
目前，关于宽带单通道零中频接收机的IQ不平衡估计与补偿方法的研究文献较多，例如中国专利公开号CN116684239A公开的一种零中频接收阵列IQ不平衡波束级补偿方法，但宽带零中频接收阵列中同时存在通道间频率响应误差和通道内IQ不平衡误差的标校和均衡方法却鲜有讨论。若宽带阵列仅存在通道间频率响应误差，以宽带信号作为标校源，通过采集各通道的宽带信号样本(标校信号样本)，可较容易的估计通道间频率响应误差，并对该误差进行补偿。然而在宽带零中频接收阵列中，若仍以宽带信号作为标校源，由于存在通道内IQ不平衡误差，使得各通道采集的信号样本中既包括宽带信号本身的样本，也包括它的镜像信号样本，这将导致标校信号样本失真，无法准确估计阵列各通道的频率响应误差。Currently, there are many research documents on the IQ imbalance estimation and compensation methods of wideband single-channel zero-IF receivers. For example, Chinese Patent Publication No. CN116684239A discloses a zero-IF receiving array IQ imbalance beam-level compensation method, but the wideband zero-IF Calibration and equalization methods for both inter-channel frequency response errors and intra-channel IQ imbalance errors in the receiving array are rarely discussed. If the broadband array only has inter-channel frequency response errors, use the broadband signal as the calibration source and collect the broadband signal samples (calibration signal samples) of each channel to easily estimate the inter-channel frequency response error and analyze the error. compensate. However, in the broadband zero-IF receiving array, if the broadband signal is still used as the calibration source, due to the IQ imbalance error within the channel, the signal samples collected by each channel include both the sample of the broadband signal itself and its image signal. samples, which will cause distortion of the calibration signal samples, making it impossible to accurately estimate the frequency response error of each channel of the array.
为了能够对宽带零中频接收通道的各种误差进行估计和补偿，同时简化宽带数字阵列中校正源设计的复杂度，有必要寻找一种既可以对宽带接收通道频率响应误差进行估计和补偿，也可以简化校正源设计复杂度，适应校正源无法产生调制信号场合，便于工程实现的多点频信号联合处理方法，解决宽带零中频接收数字阵列通道标校与均衡问题。In order to be able to estimate and compensate for various errors in the wideband zero-IF receive channel, and at the same time simplify the complexity of the correction source design in the wideband digital array, it is necessary to find a method that can not only estimate and compensate for the frequency response error of the wideband receive channel, but also It can simplify the design complexity of the correction source, adapt to situations where the correction source cannot generate modulated signals, facilitate engineering implementation of a multi-point frequency signal joint processing method, and solve the problem of broadband zero-IF receiving digital array channel calibration and equalization.
发明内容Contents of the invention
本发明所要解决的技术问题在于现有技术无法对宽带零中频接收通道的各种误差进行估计和补偿，从而无法实现宽带零中频接收数字阵列通道标校与均衡。The technical problem to be solved by the present invention is that the existing technology cannot estimate and compensate various errors of the wideband zero-IF receiving channel, and thus cannot realize the calibration and equalization of the wideband zero-IF receiving digital array channel.
本发明通过以下技术手段解决上述技术问题的：一种宽带零中频接收阵列的通道标校与均衡方法，包括以下步骤：The present invention solves the above technical problems through the following technical means: a channel calibration and equalization method for a wideband zero-IF receiving array, including the following steps:
步骤1：获取数字阵列各接收通道的基带数字复信号；Step 1: Obtain the baseband digital complex signal of each receiving channel of the digital array;
步骤2：获取第m个阵元接收的第p个频点的基带数字复单频信号；Step 2: Obtain the baseband digital complex single frequency signal of the pth frequency point received by the mth array element;
步骤3：计算各通道内每一基带数字复单频信号的离散时间傅里叶变换，得到各通道单频信号本身频谱值和对应镜像信号的频谱值；Step 3: Calculate the discrete time Fourier transform of each baseband digital complex single-frequency signal in each channel, and obtain the spectrum value of the single-frequency signal itself and the spectrum value of the corresponding image signal in each channel;
步骤4：计算各通道IQ不平衡频率响应第一补偿值和第二补偿值；Step 4: Calculate the first compensation value and the second compensation value of the IQ imbalance frequency response of each channel;
步骤5：估计IQ不平衡的第一FIR均衡器系数和第二FIR均衡器系数；Step 5: Estimate the first FIR equalizer coefficient and the second FIR equalizer coefficient of IQ imbalance;
步骤6：对各通道内所有频点的基带数字复单频信号进行补偿，得到IQ不平衡补偿后复信号；Step 6: Compensate the baseband digital complex single-frequency signals at all frequency points in each channel to obtain the IQ imbalance compensated complex signal;
步骤7：根据IQ不平衡补偿后复信号，以第一个接收通道为参考，计算其他通道相对于参考通道的通道间频率响应补偿值；Step 7: Based on the complex signal after IQ imbalance compensation, taking the first receiving channel as a reference, calculate the inter-channel frequency response compensation values of other channels relative to the reference channel;
步骤8：根据通道间频率响应补偿值，估计通道间的第三FIR均衡器系数；Step 8: Estimate the third FIR equalizer coefficient between channels according to the frequency response compensation value between channels;
步骤9：当阵列接收目标信号时，利用IQ不平衡的各个FIR均衡器系数，对阵列接收的基带数字复信号补偿，得到IQ不平衡误差补偿及通道频率响应误差补偿后的复信号。Step 9: When the array receives the target signal, use each FIR equalizer coefficient of IQ imbalance to compensate the baseband digital complex signal received by the array, and obtain the complex signal after IQ imbalance error compensation and channel frequency response error compensation.
进一步地，所述步骤1包括：Further, the step 1 includes:
数字阵列的第m个阵元接收目标信号为The mth array element of the digital array receives the target signal as
xm(t)＝A(t-τm)cos[(Ωc+Ωd)(t-τm)+φ(t-τm)]xm (t)=A(t-τm )cos[(Ωc +Ωd )(t-τm )+φ(t-τm )]
其中，φ(t)为信号t时刻的瞬时相位，Ωc为接收信号的载波模拟角频率，A(t)为信号t时刻的瞬时幅度，Ωd为信号的模拟角频率偏置，τm是空间时延差；Among them, φ(t) is the instantaneous phase of the signal at time t, Ωc is the simulated angular frequency of the carrier of the received signal, A(t) is the instantaneous amplitude of the signal at time t, Ωd is the simulated angular frequency offset of the signal, τm is the spatial delay difference;
第m个接收通道的基带数字复信号sm(n)，有The baseband digital complex signal sm (n) of the m-th receiving channel has
其中，符号代表线性卷积，/>Ts为A/D的采样频率，n为第n个采样点，符号e为自然常数，/>为信号um(n)的复共轭信号，符号*为复共轭运算，λ1,m(n)和λ2,m(n)分别为Among them, the symbol Represents linear convolution, /> Ts is the sampling frequency of A/D, n is the nth sampling point, symbol e is a natural constant,/> is the complex conjugate signal of signal um (n), the symbol * is the complex conjugate operation, λ1,m (n) and λ2,m (n) are respectively
g1,m(n)和g2,m(n)分别为g1,m (n) and g2,m (n) are respectively
其中，和/>分别为实低通滤波器/>和的离散采样序列，/>为/>相对于/>的剩余冲激响应，δ(n)为离散单位冲激序列，εm是由同相本振/>和正交本振/>之间的幅度差异带来的幅度误差，θm是由同相本振/>和正交本振/>之间的相位差异带来的相位误差，hL,m(n)是由第m个射频前端对应的等效低通滤波器冲激响应序列，/>为hL,m(n)的复共轭。in, and/> Respectively, they are real low-pass filters/> and discrete sampling sequence,/> for/> Relative to/> The residual impulse response of , δ(n) is the discrete unit impulse sequence, εm is the in-phase local oscillator/> and quadrature local oscillator/> The amplitude difference between the amplitude errors, θm is caused by the in-phase local oscillator/> and quadrature local oscillator/> The phase error caused by the phase difference between them, hL,m (n) is the equivalent low-pass filter impulse response sequence corresponding to the m-th RF front-end,/> is the complex conjugate of hL,m (n).
更进一步地，所述步骤2包括：Furthermore, the step 2 includes:
在数字阵列标校时，工作带宽B内，每个通道均接收P个频点的基带数字复单频信号，对应的基带频率分别为f1,f2,…,fP，且有-B/2≤f1<f2<…<fP≤B/2，这些频点等间隔分布在工作带宽B内，且P个基带频率f1,f2,…,fP均不等于零，其中，m＝1,2,3,…,M，p＝1,2,…,P；During digital array calibration, within the working bandwidth B, each channel receives baseband digital complex single-frequency signals at P frequency points. The corresponding baseband frequencies are f1 , f2 ,..., fP , and there is -B /2≤f1 <f2 <…<fP ≤B/2, these frequency points are equally spaced within the operating bandwidth B, and the P baseband frequencies f1 , f2 ,…, fP are not equal to zero, where , m＝1,2,3,…,M, p＝1,2,…,P;
第m个阵元接收的第p个频点的基带数字复单频信号为The baseband digital complex single frequency signal at the pth frequency point received by the mth array element is
其中，φ0为初始相位，A0为单频信号的幅度，ωp＝2πfpTs≠0为单频信号的数字角频率，λ1,m(ωp)和λ2,m(-ωp)分别为λ1,m(n)和λ2,m(n)的离散时间傅里叶变换，即in, φ0 is the initial phase, A0 is the amplitude of the single-frequency signal, ωp =2πfp Ts ≠0 is the digital angular frequency of the single-frequency signal, λ1,m (ωp ) and λ2,m (-ωp ) are the discrete time Fourier transforms of λ1,m (n) and λ2,m (n) respectively, that is
更进一步地，所述步骤3包括：Furthermore, the step 3 includes:
采集M个通道的标校信号，每个通道中包括P个基带数字复单频信号sm,p(n)，单个信号的采样点数为N(≥1)，并计算各通道内每一基带数字复单频信号sm,p(n)的离散时间傅里叶变换，得到各通道单频信号本身频谱值Sm,p(ωp)和对应镜像信号的频谱值Sm,p(-ωp)，分别表示为Collect the calibration signals of M channels. Each channel includes P baseband digital complex single-frequency signals sm,p (n). The number of sampling points of a single signal is N (≥1), and each baseband in each channel is calculated. The discrete time Fourier transform of the digital complex single frequency signal sm,p (n) obtains the spectrum value Sm,p (ωp ) of each channel single frequency signal itself and the spectrum value Sm,p (- of the corresponding image signal). ωp ), respectively expressed as
其中，n＝1,2,…,N。Among them, n=1,2,…,N.
更进一步地，所述步骤4包括：Furthermore, step 4 includes:
步骤4-1：根据通道单频信号本身频谱值Sm,p(ωp)和对应镜像信号的频谱值Sm,p(-ωp)，估计频点ωp处IQ不平衡的幅度误差ρp和相位误差ηp，即Step 4-1: Based on the spectrum value Sm,p (ωp ) of the channel single-frequency signal itself and the spectrum value Sm,p (-ωp ) of the corresponding image signal, estimate the amplitude error of the IQ imbalance at the frequency point ωp ρp and phase error ηp , that is
步骤4-2：根据频点ωp处IQ不平衡的幅度误差ρp和相位误差ηp，其中，p＝1,2,…,P，构造IQ不平衡误差频率响应第一估计值和第二估计值/>即Step 4-2: Construct the first estimate of the IQ imbalance error frequency response based on the amplitude error ρp and phase error ηp of the IQ imbalance at the frequency point ωp , where p = 1, 2,...,P and second estimate/> Right now
步骤4-3：由IQ不平衡误差频率响应第一估计值和第二估计值/>构造IQ不平衡频率响应第一补偿值W1,m(ωp)和第二补偿值W2,m(ωp)，即Step 4-3: First estimate of frequency response from IQ imbalance error and second estimate/> Construct the IQ unbalanced frequency response first compensation value W1,m (ωp ) and second compensation value W2,m (ωp ), that is
更进一步地，所述步骤5包括：Furthermore, the step 5 includes:
步骤5-1：给定FIR均衡器的阶数D1，通过求解如下最优化问题，获得IQ不平衡的第一FIR均衡器系数w1,m(n)，即Step 5-1: Given the order D1 of the FIR equalizer, obtain the first FIR equalizer coefficient w1,m (n) of IQ imbalance by solving the following optimization problem, that is
其中，P>D1；Among them, P>D1 ;
步骤5-2：给定FIR均衡器的阶数D2，通过求解如下最优化问题，获得IQ不平衡的第二FIR均衡器系数w2,m(n)，即Step 5-2: Given the order D2 of the FIR equalizer, obtain the second FIR equalizer coefficient w2,m (n) of IQ imbalance by solving the following optimization problem, that is
其中，P>D2。Among them, P>D2 .
更进一步地，所述步骤6包括：Furthermore, the step 6 includes:
利用IQ不平衡的第一FIR均衡器系数w1,m(n)和第二FIR均衡器系数w2,m(n)，对各通道内所有频点的基带数字复单频信号sm,p(n)进行补偿，得到IQ不平衡补偿后复信号表示为/>Using the IQ unbalanced first FIR equalizer coefficient w1,m (n) and the second FIR equalizer coefficient w2,m (n), for the baseband digital complex single frequency signal sm of all frequency points in each channel, p (n) is compensated to obtain the complex signal after IQ imbalance compensation Expressed as/>
更进一步地，所述步骤7包括：Furthermore, the step 7 includes:
根据IQ不平衡补偿后复信号以第一个接收通道为参考，计算其他通道相对于参考通道的通道间频率响应补偿值Gm(ωp)，表示为Compensated signal based on IQ imbalance Taking the first receiving channel as a reference, calculate the inter-channel frequency response compensation value Gm (ωp ) of other channels relative to the reference channel, expressed as
更进一步地，所述步骤8包括：Furthermore, the step 8 includes:
给定通道间FIR均衡器的阶数D3，根据通道间频率响应补偿值Gm(ωp)，p＝1,2,…,P，通过求解最优化问题，估计通道间的第三FIR均衡器系数cm(n)，通道间的第三FIR均衡器系数cm(n)满足的最优化问题为Given the order D3 of the inter-channel FIR equalizer, according to the inter-channel frequency response compensation value Gm (ωp ), p=1,2,...,P, the third FIR between channels is estimated by solving the optimization problem The optimization problem satisfied by the equalizer coefficient cm (n) and the third FIR equalizer coefficient cm (n) between channels is
其中，P>D3。Among them, P>D3 .
更进一步地，所述步骤9包括：Furthermore, the step 9 includes:
步骤9-1：利用IQ不平衡的第一FIR均衡器系数w1,m(n)和第二FIR均衡器系数w2,m(n)，对阵列接收的基带数字复信号进行补偿，得到IQ不平衡误差补偿后的复信号/>即Step 9-1: Use the IQ-unbalanced first FIR equalizer coefficient w1,m (n) and the second FIR equalizer coefficient w2,m (n) to analyze the baseband digital complex signal received by the array. Compensate and obtain the complex signal after IQ imbalance error compensation/> Right now
步骤9-2：利用通道间的第三FIR均衡器系数cm(n)，对IQ不平衡误差补偿后的复信号进行修正，得到通道频率响应误差补偿后的复信号/>即Step 9-2: Using the third FIR equalizer coefficient cm (n) between channels, the complex signal after IQ imbalance error compensation Make corrections to obtain the complex signal after channel frequency response error compensation/> Right now
本发明的优点在于：The advantages of the present invention are:
(1)本发明经过一系列计算分别获取IQ不平衡的第一FIR均衡器系数和第二FIR均衡器系数以及通道间的第三FIR均衡器系数，然后利用IQ不平衡的各个FIR均衡器系数，对阵列接收的基带数字复信号补偿，得到IQ不平衡误差补偿及通道频率响应误差补偿后的复信号，因而能够实现对宽带零中频接收通道的各种误差进行估计和补偿，实现宽带零中频接收数字阵列通道标校与均衡。(1) The present invention obtains the first FIR equalizer coefficient and the second FIR equalizer coefficient of IQ imbalance and the third FIR equalizer coefficient between channels through a series of calculations, and then uses each FIR equalizer coefficient of IQ imbalance , the baseband digital complex signal received by the array is compensated to obtain a complex signal after IQ imbalance error compensation and channel frequency response error compensation. Therefore, various errors of the wideband zero-IF receiving channel can be estimated and compensated to achieve wideband zero-IF Receive digital array channel calibration and equalization.
(2)本发明不需要发送复杂的宽带标校信号，不但可以仅用点频信号作为校正源，简化校正源设计的复杂度，而且可以将通道间频率响应不一致性和通道内IQ不平衡误差分别标定与校正，降低校正装置实现复杂度，便于工程实现。(2) The present invention does not need to send complex broadband calibration signals. Not only can it only use point frequency signals as the correction source, simplifying the complexity of the correction source design, but it can also reduce the frequency response inconsistency between channels and the IQ imbalance error within the channel. Separate calibration and correction reduce the complexity of the correction device and facilitate engineering implementation.
(3)本发明不需要事先存储校正源发送的参考信号，原理简单，运算量小，节省存储空间。本发明所估计出的IQ不平衡误差不是通道内的绝对误差，但经IQ不平衡误差补偿后，可有效抑制IQ不平衡带来的镜像分量，提高了阵列的整体镜像抑制比。(3) The present invention does not need to store the reference signal sent by the correction source in advance, has a simple principle, a small amount of calculation, and saves storage space. The IQ imbalance error estimated by the present invention is not an absolute error within the channel, but after IQ imbalance error compensation, the image component caused by the IQ imbalance can be effectively suppressed, thereby improving the overall image suppression ratio of the array.
(4)本发明可以将通道间频率响应误差和通道内IQ不平衡误差分别标定与校正，提高了校正性能，便于工程实现。整个方案涉及的方法不受阵列结构的限制，既适用于平面数字阵列，也可以用于共形数字阵列。(4) The present invention can separately calibrate and correct the frequency response error between channels and the IQ imbalance error within the channel, which improves the correction performance and facilitates engineering implementation. The method involved in the entire solution is not limited by the array structure and is applicable to both planar digital arrays and conformal digital arrays.
附图说明Description of the drawings
图1为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法的流程图；Figure 1 is a flow chart of a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention;
图2为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中IQ不平衡误差和通道间频率响应补偿的框图；Figure 2 is a block diagram of IQ imbalance error and inter-channel frequency response compensation in a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention;
图3为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中目标信号对应的通道频率响应示意图；其中，图3(a)为目标信号对应的通道幅频响应示意图；图3(b)为目标信号对应的通道相频响应示意图；Figure 3 is a schematic diagram of the channel frequency response corresponding to the target signal in the channel calibration and equalization method of the wideband zero-IF receiving array disclosed in the embodiment of the present invention; wherein, Figure 3(a) is the channel amplitude-frequency response corresponding to the target signal Schematic diagram; Figure 3(b) is a schematic diagram of the phase-frequency response of the channel corresponding to the target signal;
图4为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中镜像信号对应的通道频率响应示意图；其中，图4(a)为镜像信号对应的通道幅频响应示意图；图4(b)为镜像信号对应的通道相频响应示意图；Figure 4 is a schematic diagram of the channel frequency response corresponding to the image signal in the channel calibration and equalization method of the wideband zero-IF receiving array disclosed in the embodiment of the present invention; wherein, Figure 4(a) is the channel amplitude-frequency response corresponding to the image signal Schematic diagram; Figure 4(b) is a schematic diagram of the channel phase-frequency response corresponding to the image signal;
图5为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中W1,m(ωp)的估计和FIR拟合的幅频响应示意图；Figure 5 is a schematic diagram of the amplitude-frequency response of the estimation of W1,m (ωp ) and the FIR fitting in the channel calibration and equalization method of the wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图6为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中W1,m(ωp)的估计和FIR拟合的相频响应示意图；Figure 6 is a schematic diagram of the phase-frequency response of W1,m (ωp ) estimation and FIR fitting in the channel calibration and equalization method of a wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图7为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中W2,m(ωp)的估计和FIR拟合的幅频响应示意图；Figure 7 is a schematic diagram of the amplitude-frequency response of W2,m (ωp ) estimation and FIR fitting in the channel calibration and equalization method of a wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图8为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中W2,m(ωp)的估计和FIR拟合的相频响应示意图；Figure 8 is a schematic diagram of the phase-frequency response of W2,m (ωp ) estimation and FIR fitting in the channel calibration and equalization method of a wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图9为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中标校信号在IQ不平衡补偿前的频谱图；Figure 9 is a spectrum diagram of the calibration signal before IQ imbalance compensation in the channel calibration and equalization method of the wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图10为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中标校信号在IQ不平衡补偿后的频谱图；Figure 10 is a spectrum diagram of the calibration signal after IQ imbalance compensation in the channel calibration and equalization method of a wideband zero-IF receiving array disclosed in the embodiment of the present invention;
图11为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中单次仿真的通道间相对相频响应和幅频响应示意图；其中，图11(a)为单次仿真的通道间相对幅频响应示意图；图11(b)为单次仿真的通道间相对相频响应示意图；Figure 11 is a schematic diagram of the relative phase-frequency response and amplitude-frequency response between channels in a single simulation of a wideband zero-IF receiving array channel calibration and equalization method disclosed in an embodiment of the present invention; wherein, Figure 11(a) is a single simulation A schematic diagram of the relative amplitude-frequency response between channels for a single simulation; Figure 11(b) is a schematic diagram of the relative phase-frequency response between channels for a single simulation;
图12为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中通道间相对幅频响应估计值示意图；Figure 12 is a schematic diagram of the estimated relative amplitude-frequency response between channels in a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention;
图13为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中通道间相对相频响应估计值示意图；Figure 13 is a schematic diagram of the estimated relative phase-frequency response between channels in the channel calibration and equalization method of a wideband zero-IF receiving array disclosed in an embodiment of the present invention;
图14为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中射频频率8GHz，通道误差补偿前，目标与镜像信号对应的阵列法向波束图；Figure 14 is a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention. The radio frequency frequency is 8 GHz and before channel error compensation, the array normal beam diagram corresponding to the target and the image signal;
图15为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中射频频率8GHz，通道误差补偿后，目标与镜像信号对应的阵列法向波束图；Figure 15 is a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention. In the radio frequency frequency 8GHz, after channel error compensation, the array normal beam diagram corresponding to the target and the image signal;
图16为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中射频频率7.9GHz，通道误差补偿后，目标与镜像信号对应的阵列波束图；Figure 16 is a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention. In the radio frequency frequency 7.9GHz, after channel error compensation, the array beam diagram corresponding to the target and the image signal;
图17为本发明实施例所公开的一种宽带零中频接收阵列的通道标校与均衡方法中射频频率8.1GHz，通道误差补偿后，目标与镜像信号对应的阵列波束图。Figure 17 is a channel calibration and equalization method for a wideband zero-IF receiving array disclosed in an embodiment of the present invention. The RF frequency is 8.1 GHz. After channel error compensation, the array beam diagram corresponds to the target and the image signal.
具体实施方式Detailed ways
为使本发明实施例的目的、技术方案和优点更加清楚，下面将结合本发明实施例，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有作出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。In order to make the purpose, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are part of the present invention. Examples, not all examples. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without making creative efforts fall within the scope of protection of the present invention.
如图1所示，本发明提供一种宽带零中频接收阵列的通道标校与均衡方法，包括以下步骤：As shown in Figure 1, the present invention provides a channel calibration and equalization method for a wideband zero-IF receiving array, which includes the following steps:
步骤1：数字阵列共有M(≥2)个阵元，每个阵元对应接收通道的工作带宽均为B。在该工作带宽内，阵元接收的目标信号，经零中频接收阵列各通道预选滤波、低噪声放大、同相与正交混频、低通滤波和A/D采集后，得到M个接收通道的基带数字复信号具体过程为：Step 1: The digital array has a total of M (≥2) array elements, and the working bandwidth of the corresponding receiving channel of each array element is B. Within this working bandwidth, after the target signal received by the array element is pre-selected and filtered by each channel of the zero-IF receiving array, low-noise amplification, in-phase and quadrature mixing, low-pass filtering and A/D acquisition, the M receiving channels are obtained. baseband digital complex signal The specific process is:
在工作带宽B内，数字阵列的第m个阵元接收目标信号为Within the working bandwidth B, the m-th array element of the digital array receives the target signal as
xm(t)＝A(t-τm)cos[(Ωc+Ωd)(t-τm)+φ(t-τm)]xm (t)=A(t-τm )cos[(Ωc +Ωd )(t-τm )+φ(t-τm )]
其中，φ(t)为信号t时刻的瞬时相位，Ωc为接收信号的载波模拟角频率，A(t)为信号t时刻的瞬时幅度，Ωd为信号的模拟角频率偏置，τm是由阵元位置和信号入射角度决定的空间时延差；Among them, φ(t) is the instantaneous phase of the signal at time t, Ωc is the simulated angular frequency of the carrier of the received signal, A(t) is the instantaneous amplitude of the signal at time t, Ωd is the simulated angular frequency offset of the signal, τm It is the spatial delay difference determined by the array element position and signal incident angle;
经零中频接收阵列各通道预选滤波、低噪声放大、同相与正交混频、低通滤波和A/D采集后，得到第m个接收通道的基带数字复信号sm(n)，有After pre-selection filtering, low-noise amplification, in-phase and quadrature mixing, low-pass filtering and A/D acquisition of each channel of the zero-IF receiving array, the baseband digital complex signal sm (n) of the m-th receiving channel is obtained, as follows:
其中，符号代表线性卷积，/>Ts为A/D的采样频率，n为第n个采样点，符号e为自然常数，/>为信号um(n)的复共轭信号，符号*为复共轭运算，λ1,m(n)和λ2,m(n)分别为Among them, the symbol Represents linear convolution, /> Ts is the sampling frequency of A/D, n is the nth sampling point, symbol e is a natural constant,/> is the complex conjugate signal of signal um (n), the symbol * is the complex conjugate operation, λ1,m (n) and λ2,m (n) are respectively
g1,m(n)和g2,m(n)分别为g1,m (n) and g2,m (n) are respectively
其中，和/>分别为实低通滤波器/>和的离散采样序列，/>为/>相对于/>的剩余冲激响应，δ(n)为离散单位冲激序列，εm是由同相本振/>和正交本振/>之间的幅度差异带来的幅度误差，θm是由同相本振/>和正交本振/>之间的相位差异带来的相位误差，hL,m(n)是由第m个射频前端对应的等效低通滤波器冲激响应序列，/>为hL,m(n)的复共轭。in, and/> Respectively, they are real low-pass filters/> and discrete sampling sequence,/> for/> Relative to/> The residual impulse response of , δ(n) is the discrete unit impulse sequence, εm is the in-phase local oscillator/> and quadrature local oscillator/> The amplitude difference between the amplitude errors, θm is caused by the in-phase local oscillator/> and quadrature local oscillator/> The phase error caused by the phase difference between them, hL,m (n) is the equivalent low-pass filter impulse response sequence corresponding to the m-th RF front-end,/> is the complex conjugate of hL,m (n).
步骤2：在数字阵列标校时，工作带宽B内，每个通道均接收P(≥4)个频点的基带数字复单频信号sm,p(n)，对应的基带频率分别为f1,f2,…,fP，且有-B/2≤f1<f2<…<fP≤B/2，这些频点等间隔分布在工作带宽B内，且P个基带频率f1,f2,…,fP均不等于零，其中，m＝1,2,3,…,M，p＝1,2,…,P；Step 2: During digital array calibration, within the working bandwidth B, each channel receives P (≥4) frequency points of baseband digital complex single-frequency signals sm,p (n), and the corresponding baseband frequencies are f.1 ,f2 ,…,fP , and -B/2≤f1 <f2 <…<fP ≤B/2, these frequency points are equally spaced within the operating bandwidth B, and P baseband frequencies f1 , f2 ,…,fP are not equal to zero, where m=1,2,3,…,M, p=1,2,…,P;
第m个阵元接收的第p个频点的基带数字复单频信号为The baseband digital complex single frequency signal at the pth frequency point received by the mth array element is
其中，φ0为初始相位，A0为单频信号的幅度，ωp＝2πfpTs≠0为单频信号的数字角频率，λ1,m(ωp)和λ2,m(-ωp)分别为λ1,m(n)和λ2,m(n)的离散时间傅里叶变换，即in, φ0 is the initial phase, A0 is the amplitude of the single-frequency signal, ωp =2πfp Ts ≠0 is the digital angular frequency of the single-frequency signal, λ1,m (ωp ) and λ2,m (-ωp ) are the discrete time Fourier transforms of λ1,m (n) and λ2,m (n) respectively, that is
步骤3：采集M(≥2)个通道的标校信号，每个通道中包括P(≥4)个基带数字复单频信号sm,p(n)，单个信号的采样点数为N(≥1)，并计算各通道内每一基带数字复单频信号sm,p(n)的离散时间傅里叶变换，得到各通道单频信号本身频谱值Sm,p(ωp)和对应镜像信号的频谱值Sm,p(-ωp)，分别表示为Step 3: Collect the calibration signals of M (≥2) channels. Each channel includes P (≥4) baseband digital complex single-frequency signals sm,p (n). The number of sampling points of a single signal is N (≥ 1), and calculate the discrete time Fourier transform of each baseband digital complex single frequency signal sm,p (n) in each channel, and obtain the spectrum value Sm,p (ωp ) of each channel single frequency signal itself and the corresponding The spectrum value Sm,p (-ωp ) of the image signal is expressed as
其中，m＝1,2,3,…,M，p＝1,2,…,P，n＝1,2,…,N。Among them, m=1,2,3,...,M, p=1,2,...,P, n=1,2,...,N.
步骤4：根据各通道单频信号本身频谱值Sm,p(ωp)和对应镜像信号的频谱值Sm,p(-ωp)，计算各通道IQ不平衡频率响应第一补偿值W1,m(ωp)和第二补偿值W2,m(ωp)，其中，m＝1,2,3,…,M，p＝1,2,…,P；各通道的IQ不平衡频率响应补偿值的计算方法相同，对于第m个通道，具体过程为：Step 4: Calculate the first compensation value W of the IQ imbalance frequency response of each channel based on the spectrum value Sm,p (ωp ) of each channel's single-frequency signal and the spectrum value Sm,p (-ωp ) of the corresponding image signal.1,m (ωp ) and the second compensation value W2,m (ωp ), where m=1,2,3,…,M, p=1,2,…,P; the IQ of each channel does not The calculation method of the balanced frequency response compensation value is the same. For the m-th channel, the specific process is:
步骤4-1：根据通道单频信号本身频谱值Sm,p(ωp)和对应镜像信号的频谱值Sm,p(-ωp)，估计频点ωp处IQ不平衡的幅度误差ρp和相位误差ηp，即Step 4-1: Based on the spectrum value Sm,p (ωp ) of the channel single-frequency signal itself and the spectrum value Sm,p (-ωp ) of the corresponding image signal, estimate the amplitude error of the IQ imbalance at the frequency point ωp ρp and phase error ηp , that is
步骤4-2：根据频点ωp处IQ不平衡的幅度误差ρp和相位误差ηp，其中，p＝1,2,…,P，构造IQ不平衡误差频率响应第一估计值和第二估计值/>即Step 4-2: According to the amplitude error ρp and phase error ηp of the IQ imbalance at the frequency point ωp , where p = 1, 2,...,P, construct the first estimate of the frequency response of the IQ imbalance error and second estimate/> Right now
步骤4-3：由IQ不平衡误差频率响应第一估计值和第二估计值/>构造IQ不平衡频率响应第一补偿值W1,m(ωp)和第二补偿值W2,m(ωp)，即Step 4-3: First estimate of frequency response from IQ imbalance error and second estimate/> Construct the IQ unbalanced frequency response first compensation value W1,m (ωp ) and second compensation value W2,m (ωp ), that is
步骤5：根据IQ不平衡频率响应第一补偿值W1,m(ωp)和第二补偿值W2,m(ωp)，p＝1,2,…,P，分别估计IQ不平衡的第一FIR均衡器系数w1,m(n)和第二FIR均衡器系数w2,m(n)；具体过程为：Step 5: Estimate the IQ imbalance respectively according to the first compensation value W1,m (ωp ) and the second compensation value W2,m (ωp ), p=1,2,…,P according to the IQ imbalance frequency response The first FIR equalizer coefficient w1,m (n) and the second FIR equalizer coefficient w2,m (n); the specific process is:
步骤5-1：给定FIR均衡器的阶数D1，通过求解如下最优化问题，获得IQ不平衡的第一FIR均衡器系数w1,m(n)，即Step 5-1: Given the order D1 of the FIR equalizer, obtain the first FIR equalizer coefficient w1,m (n) of IQ imbalance by solving the following optimization problem, that is
其中，P>D1；Among them, P>D1 ;
步骤5-2：给定FIR均衡器的阶数D2，通过求解如下最优化问题，获得IQ不平衡的第二FIR均衡器系数w2,m(n)，即Step 5-2: Given the order D2 of the FIR equalizer, obtain the second FIR equalizer coefficient w2,m (n) of IQ imbalance by solving the following optimization problem, that is
其中，P>D2。Among them, P>D2 .
步骤6：利用IQ不平衡的第一FIR均衡器系数w1,m(n)和第二FIR均衡器系数w2,m(n)，对各通道内所有频点的基带数字复单频信号sm,p(n)进行补偿，得到IQ不平衡补偿后复信号表示为/>Step 6: Use the IQ unbalanced first FIR equalizer coefficient w1,m (n) and the second FIR equalizer coefficient w2,m (n) to complex the baseband digital single frequency signal of all frequency points in each channel sm,p (n) is compensated, and the complex signal after IQ imbalance compensation is obtained. Expressed as/>
步骤7：根据IQ不平衡补偿后复信号以第一个接收通道为参考，计算其他通道相对于参考通道的通道间频率响应补偿值Gm(ωp)，表示为Step 7: Compensate the signal based on IQ imbalance Taking the first receiving channel as a reference, calculate the inter-channel frequency response compensation value Gm (ωp ) of other channels relative to the reference channel, expressed as
步骤8：给定通道间FIR均衡器的阶数D3，根据通道间频率响应补偿值Gm(ωp)，p＝1,2,…,P，通过求解最优化问题，估计通道间的第三FIR均衡器系数cm(n)，通道间的第三FIR均衡器系数cm(n)满足的最优化问题为Step 8: Given the order D3 of the inter-channel FIR equalizer, according to the inter-channel frequency response compensation value Gm (ωp ), p=1,2,...,P, estimate the inter-channel by solving the optimization problem The optimization problem satisfied by the third FIR equalizer coefficient cm (n) and the third FIR equalizer coefficient cm (n) between channels is:
其中，P>D3。Among them, P>D3 .
步骤9：当阵列接收如步骤1所述的目标信号时，利用IQ不平衡的第一FIR均衡器系数w1,m(n)、第二FIR均衡器系数w2,m(n)及通道间的第三FIR均衡器系数cm(n)，按图2所示对对阵列接收的基带数字复信号依次补偿，得到IQ不平衡误差补偿及通道频率响应误差补偿后的复信号/>具体过程为：Step 9: When the array receives the target signal as described in step 1, use the IQ unbalanced first FIR equalizer coefficient w1,m (n), the second FIR equalizer coefficient w2,m (n) and the channel The third FIR equalizer coefficient cm (n) between, as shown in Figure 2 for the baseband digital complex signal received by the array Compensate in sequence to obtain the complex signal after IQ imbalance error compensation and channel frequency response error compensation/> The specific process is:
步骤9-1：利用IQ不平衡的第一FIR均衡器系数w1,m(n)和第二FIR均衡器系数w2,m(n)，对阵列接收的基带数字复信号进行补偿，得到IQ不平衡误差补偿后的复信号/>即Step 9-1: Use the IQ-unbalanced first FIR equalizer coefficient w1,m (n) and the second FIR equalizer coefficient w2,m (n) to analyze the baseband digital complex signal received by the array. Compensate and obtain the complex signal after IQ imbalance error compensation/> Right now
步骤9-2：利用通道间的第三FIR均衡器系数cm(n)，对IQ不平衡误差补偿后的复信号进行修正，得到通道频率响应误差补偿后的复信号/>即Step 9-2: Using the third FIR equalizer coefficient cm (n) between channels, the complex signal after IQ imbalance error compensation Make corrections to obtain the complex signal after channel frequency response error compensation/> Right now
以下对本发明的方案进行仿真验证，在仿真实验中，系统工作的射频频率为8GHz，系统采样率240MHz，工作带宽B＝200MHz，数字阵列为均匀线阵，阵元间距为11mm，阵元个数为64。The solution of the present invention is simulated and verified below. In the simulation experiment, the radio frequency frequency of the system operation is 8GHz, the system sampling rate is 240MHz, the working bandwidth B=200MHz, the digital array is a uniform linear array, the array element spacing is 11mm, and the number of array elements is 64.
场景1：单通道内，IQ不平衡误差估计与均衡Scenario 1: IQ imbalance error estimation and equalization in a single channel
对于零中频接收阵列的某一通道，目标信号um(n)对应的通道频率响应λ1,m(ω)如图3所示，镜像信号um(n)对应的通道频率响应λ2,m(ω)如图4所示。由图3和4可知，在整个频带内，镜像抑制比大概在20dB～25dB之间。For a certain channel of the zero-IF receiving array, the channel frequency response λ1,m (ω) corresponding to the target signal um (n) is shown in Figure 3, and the channel frequency response λ 2, corresponding to the image signal um (n), m (ω) is shown in Figure 4. It can be seen from Figures 3 and 4 that within the entire frequency band, the image rejection ratio is approximately between 20dB and 25dB.
阵列接收单频信号，信噪比为20dB，采集样本个数2400，在整个工作带宽内，共接收200个单频信号，这些信号的频点均匀分布在-100MHz到100MHz内。利用本发明所提方法对IQ不平衡的频率响应补偿值W1,m(ωp)和W2,m(ωp)进行估计，并用16阶FIR均衡器w1,m(n)和w2,m(n)对频率响应补偿值W1,m(ωp)和W2,m(ωp)进行拟合。IQ不平衡的频率响应补偿值W1,m(ωp)的估计与FIR均衡器w1,m(n)拟合幅频响应和相频响应分别如图5和图6所示。IQ不平衡的频率响应补偿值W2,m(ωp)的估计与FIR均衡器w2,m(n)拟合幅频响应和相频响应分别如图7和图8所示。The array receives single-frequency signals with a signal-to-noise ratio of 20dB, and the number of collected samples is 2400. Within the entire working bandwidth, a total of 200 single-frequency signals are received. The frequency points of these signals are evenly distributed between -100MHz and 100MHz. The method proposed by the present invention is used to estimate the frequency response compensation values W1,m (ωp ) and W2,m (ωp ) of the IQ imbalance, and the 16th-order FIR equalizer w1,m (n) and w2,m (n) fits the frequency response compensation values W1,m (ωp ) and W2,m (ωp ). The estimation of the frequency response compensation value W1,m (ωp ) of the IQ imbalance and the fitted amplitude-frequency response and phase-frequency response of the FIR equalizer w1,m (n) are shown in Figure 5 and Figure 6 respectively. The estimation of the frequency response compensation value W2,m (ωp ) of the IQ imbalance and the fitted amplitude-frequency response and phase-frequency response of the FIR equalizer w2,m (n) are shown in Figures 7 and 8 respectively.
200个单频标校复信号在IQ不平衡频率响应补偿前的频谱图如图9所示。经前述FIR均衡器补偿后，200个单频标校信号的频谱图如图10所示。由图可知，经标校和补偿后，通道内的IQ不平衡带来的镜像分量得到抑制。The spectrum diagram of 200 single-frequency calibration signals before IQ unbalanced frequency response compensation is shown in Figure 9. After compensation by the aforementioned FIR equalizer, the spectrum diagram of 200 single-frequency calibration signals is shown in Figure 10. It can be seen from the figure that after calibration and compensation, the image component caused by the IQ imbalance in the channel is suppressed.
场景2：双通道情况，IQ不平衡补偿后，通道间幅相误差估计Scenario 2: Dual-channel situation, after IQ imbalance compensation, inter-channel amplitude and phase error estimation
对于宽带零中频接收阵列的某两个通道，各通道原信号和镜像信号对应的IQ不平衡频率响应平均值如场景1中图3和图4所示。两个通道在各频点处除存在IQ平衡随机误差外，还存在通道间频率响应随机误差。在工作带宽内，幅度不平衡随机误差最大值为1dB，相位不平衡随机误差最大值为5度。通道间幅度随机误差最大值为1dB，相位随机误差最大值为10度。此场景下，单次仿真的通道间相对幅频响应和相频响应如图11所示。For certain two channels of the wideband zero-IF receiving array, the average IQ imbalance frequency response corresponding to the original signal and image signal of each channel is shown in Figure 3 and Figure 4 in Scenario 1. In addition to the IQ balance random errors of the two channels at each frequency point, there are also inter-channel frequency response random errors. Within the working bandwidth, the maximum random error of amplitude imbalance is 1dB, and the maximum random error of phase imbalance is 5 degrees. The maximum amplitude random error between channels is 1dB, and the maximum phase random error is 10 degrees. In this scenario, the relative amplitude-frequency response and phase-frequency response between channels in a single simulation are shown in Figure 11.
利用200个单频标校信号对整个工作频段进行扫描，单频标校复信号的信噪比为20dB，通道间的相对频率响应估计值如图12和图13所示。由图可知，利用本发明所提方法对通道间频率响应的估计值与图11给出的仿真值非常接近，验证了所提方法的正确性。200 single-frequency calibration signals are used to scan the entire working frequency band. The signal-to-noise ratio of the single-frequency calibration signal is 20dB. The relative frequency response estimates between channels are shown in Figures 12 and 13. It can be seen from the figure that the estimated value of the inter-channel frequency response using the method proposed in the present invention is very close to the simulation value shown in Figure 11, which verifies the correctness of the proposed method.
场景3：阵列通道标校与均衡前后，阵列波束图Scenario 3: Array beam pattern before and after array channel calibration and equalization
对于宽带零中频接收阵列，各通道原信号和镜像信号对应的IQ不平衡频率响应平均值如场景1中图3和图4所示。各通道在各频点处除存在IQ平衡随机误差外，还存在通道间频率响应随机误差。在工作带宽内，幅度不平衡随机误差最大值为1dB，相位不平衡随机误差最大值为5度。通道间幅度随机误差最大值为1dB，相位随机误差最大值为10度。For the wideband zero-IF receiving array, the average IQ imbalance frequency response corresponding to the original signal and image signal of each channel is shown in Figure 3 and Figure 4 in Scenario 1. In addition to IQ balance random errors in each channel at each frequency point, there are also inter-channel frequency response random errors. Within the working bandwidth, the maximum random error of amplitude imbalance is 1dB, and the maximum random error of phase imbalance is 5 degrees. The maximum amplitude random error between channels is 1dB, and the maximum phase random error is 10 degrees.
在工作带宽内，利用200个单频标校信号对整个工作带宽进行扫描，单频标校复信号的信噪比为20dB，IQ不平衡FIR均衡器阶数为16，通道间FIR均衡器阶数为16。经各通道的IQ不平衡频率响应误差和通道间频率响应误差补偿后，对各通道的复信号分别进行波束加权修正和分数时延补偿，完成宽带波束合成。Within the working bandwidth, 200 single-frequency calibration signals are used to scan the entire working bandwidth. The signal-to-noise ratio of the single-frequency calibration signal is 20dB. The IQ unbalanced FIR equalizer order is 16. The inter-channel FIR equalizer order The number is 16. After compensation for the IQ unbalanced frequency response error of each channel and the frequency response error between channels, the complex signal of each channel Beam weighting correction and fractional delay compensation are performed respectively to complete wideband beam synthesis.
对于目标射频频率8GHz，当不对通道误差补偿时，目标与镜像信号对应的阵列法向波束图如图14所示。由图可知，相比于单通道的IQ不平衡镜像抑制水平，波束合成对IQ不平衡带来的镜像分量没有抑制能力，波束合成后的镜像抑制比近似等于单通道在中心频率处的平均镜像抑制比。经通道误差补偿后，目标与镜像信号对应的阵列法向波束图如图15所示。由图可知，通道误差补偿将镜像抑制比从补偿前23dB左右提高到67dB以上。本发明所提通道标校方法可进一步提高波束合成输出的镜像抑制比。For the target RF frequency of 8GHz, when the channel error is not compensated, the array normal beam diagram corresponding to the target and image signals is shown in Figure 14. It can be seen from the figure that compared with the IQ imbalance image suppression level of a single channel, beamforming has no ability to suppress the image component caused by IQ imbalance. The image suppression ratio after beamforming is approximately equal to the average image of a single channel at the center frequency. Suppression ratio. After channel error compensation, the array normal beam pattern corresponding to the target and image signals is shown in Figure 15. It can be seen from the figure that channel error compensation increases the image rejection ratio from about 23dB before compensation to more than 67dB. The channel calibration method proposed by the present invention can further improve the image suppression ratio of the beam synthesis output.
对于目标射频频率7.9GHz和8.1GHz，波束指向设为45度，经通道误差补偿后，目标与镜像信号对应的阵列波束图分别如图16和17所示。由图可知，在系统最小和最大射频工作频率处，均可进行波束合成，通道误差补偿将频域镜像抑制比从补偿前23dB左右提高到87dB以上，将空域镜像抑制比从补偿前23dB左右提高到67dB以上。本发明所提通道标校与均衡方法可提高波束合成输出的空域和频域镜像抑制比，可用于宽带波束合成。For the target RF frequencies of 7.9GHz and 8.1GHz, the beam direction is set to 45 degrees. After channel error compensation, the array beam diagrams corresponding to the target and image signals are shown in Figures 16 and 17 respectively. It can be seen from the figure that beam synthesis can be performed at the minimum and maximum radio frequency operating frequencies of the system. Channel error compensation increases the frequency domain image suppression ratio from about 23dB before compensation to more than 87dB, and increases the air domain image suppression ratio from about 23dB before compensation. to more than 67dB. The channel calibration and equalization method proposed by the present invention can improve the spatial domain and frequency domain image suppression ratio of the beam synthesis output, and can be used for broadband beam synthesis.
综上所述，本发明的宽带零中频接收阵列通道标校与均衡方法原理简单，易于工程实现，其性能已通过仿真实验得到验证。本发明的方法不需要发送复杂的宽带标校信号，不但可以仅用点频信号作为校正源，简化校正源设计的复杂度，而且可以将通道间频率响应不一致性和通道内IQ不平衡误差分别标定与校正，降低校正装置实现复杂度。In summary, the wideband zero-IF receiving array channel calibration and equalization method of the present invention is simple in principle and easy to implement in engineering, and its performance has been verified through simulation experiments. The method of the present invention does not need to send complex broadband calibration signals. It can not only use point frequency signals as correction sources, simplifying the complexity of correction source design, but also can separate the frequency response inconsistency between channels and the IQ imbalance error within the channels. Calibration and correction reduce the complexity of the correction device.
以上实施例仅用以说明本发明的技术方案，而非对其限制；尽管参照前述实施例对本发明进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本发明各实施例技术方案的精神和范围。The above embodiments are only used to illustrate the technical solutions of the present invention, but not to limit them; although the present invention has been described in detail with reference to the foregoing embodiments, those of ordinary skill in the art should understand that they can still modify the technical solutions of the foregoing embodiments. The recorded technical solutions may be modified, or some of the technical features thereof may be equivalently replaced; however, these modifications or substitutions shall not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of each embodiment of the present invention.
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