技术领域Technical field
本发明涉及一种变流器控制方法，涉及DCDC变换器控制技术领域，具体涉及一种双向DCDC变流器控制方法。The present invention relates to a converter control method, relates to the technical field of DCDC converter control, and specifically relates to a bidirectional DCDC converter control method.
背景技术Background technique
为改善蓄电池不能提供脉冲型功率负载的缺点，蓄电池-超级电容混合储能系统在电动汽车中得到广泛应用。在电动汽车混合储能系统中，蓄电池与超级电容通过双向DCDC(Direct Current)变流器相连接。超级电容主要承担负载需求功率中的峰值功率，蓄电池则承担平均功率，从而达到超级电容对蓄电池“削峰填谷”的作用，优化蓄电池的充放电过程，提高了燃料的经济性。双向同步整流Buck-Boost变换器是常用的混合供电系统拓扑。In order to improve the shortcoming of batteries that cannot provide pulse-type power loads, battery-supercapacitor hybrid energy storage systems are widely used in electric vehicles. In the hybrid energy storage system of electric vehicles, the battery and supercapacitor are connected through a bidirectional DCDC (Direct Current) converter. The supercapacitor mainly bears the peak power of the load demand power, while the battery bears the average power, thereby achieving the role of the supercapacitor in "peak-shaving and valley-filling" of the battery, optimizing the battery's charging and discharging process, and improving fuel economy. Bidirectional synchronous rectification Buck-Boost converter is a commonly used hybrid power supply system topology.
在Buck工作模式，主开关管Q1和同步开关管Q2互补导通。变换器两端均为有源型负载，如果变换器工作在非连续导通模式，在电感电流恢复至零后，由于同步开关管仍处于导通状态，电路将会进入强制连续模式，电感电流反向持续增大直到下个开关周期到来。由于电感存在磁损耗和直流电阻损耗，强制连续模式将会降低系统效率。In the Buck operating mode, the main switchQ1 and the synchronous switchQ2 are complementary to each other. Both ends of the converter are active loads. If the converter operates in discontinuous conduction mode, after the inductor current returns to zero, since the synchronous switch is still on, the circuit will enter the forced continuous mode, and the inductor current will The reverse direction continues to increase until the next switching cycle. Due to the magnetic losses and DC resistance losses in the inductor, forcing continuous mode will reduce system efficiency.
目前常用方法有大电感储能和零电流关断两种。大电感储能是采用较大值电感，使变换器工作在连续导通模式，但是这一方法会增大变换器的体积、不利于功率密度的提升，且成本较高；零电流关断是在电流为零时关断同步开关管，由于过零点的检测易受测量噪声、传导干扰等因素的影响，容易影响控制效果。Currently, two commonly used methods include large inductance energy storage and zero current shutdown. Large inductor energy storage uses a larger value inductor to make the converter work in continuous conduction mode. However, this method will increase the size of the converter, is not conducive to the improvement of power density, and is costly; zero current shutdown is When the current is zero, the synchronous switch tube is turned off. Since the detection of the zero-crossing point is easily affected by factors such as measurement noise and conduction interference, it is easy to affect the control effect.
发明内容Contents of the invention
为了解决背景技术中存在的问题，本发明所提供一种高效能双向DCDC变流器控制方法，解决双向DCDC变流电路工作在强制连续模式的问题，从而降低磁损耗和直流电阻损耗。In order to solve the problems existing in the background technology, the present invention provides a high-efficiency bidirectional DCDC converter control method to solve the problem of the bidirectional DCDC converter circuit operating in the forced continuous mode, thereby reducing magnetic losses and DC resistance losses.
本发明采用的技术方案是：The technical solution adopted by the present invention is:
本发明的高效能双向DCDC变流器控制方法，包括：The high-efficiency bidirectional DCDC converter control method of the present invention includes:
步骤一，根据双向DCDC变流器的电感电流判断瞬时电流流向。Step 1: Determine the instantaneous current flow direction based on the inductor current of the bidirectional DCDC converter.
步骤二，根据双向DCDC变流器的瞬时电流流向，在考虑双向DCDC变流器的工作模式的情况下，获得双向DCDC变流器在时间宽度上的驱动信号调制因子。Step 2: According to the instantaneous current flow direction of the bidirectional DCDC converter, and considering the operating mode of the bidirectional DCDC converter, obtain the drive signal modulation factor of the bidirectional DCDC converter in the time width.
步骤三，根据双向DCDC变流器的驱动信号调制因子对双向DCDC变流器的驱动信号进行调制，实现双向DCDC变流器的控制。Step 3: Modulate the drive signal of the bidirectional DCDC converter according to the drive signal modulation factor of the bidirectional DCDC converter to realize control of the bidirectional DCDC converter.
所述的步骤一中，使用基于迟滞比较器的电流阈值比较进行瞬时电流流向的判断，有利于消除纹波、抖动等干扰的影响；当电感电流iL小于迟滞比较器的第一预设阈值ith_l时，判断瞬时电流流向为负，此时工作模式为双向DCDC变流器的同步开关管Q2关断，由同步开关管Q2的反并联二极管D2续流；当电感电流iL大于迟滞比较器的第二预设阈值ith_h时，判断瞬时电流流向为正，此时工作模式为同步开关管Q2开通，由同步开关管Q2续流；当电感电流iL在第一预设阈值ith_l和第二预设阈值ith_h之间，即ith_l≤iL≤ith_h时，ith_l<0<ith_h，则判断瞬时电流流向及工作模式与上一瞬时的相同，避免了功率器件的频繁开通、关断。In the described step one, the current threshold comparison based on the hysteresis comparator is used to determine the instantaneous current flow direction, which is beneficial to eliminating the effects of interference such as ripple and jitter; when the inductor current iL is less than the first preset threshold of the hysteresis comparator When ith_l , it is judged that the instantaneous current flow direction is negative. At this time, the working mode is that the synchronous switchQ2 of the bidirectional DCDC converter is turned off, and the anti-parallel diodeD2 of the synchronous switchQ2 continues to flow; when the inductor current iL When it is greater than the second preset threshold ith_h of the hysteresis comparator, it is judged that the instantaneous current flow direction is positive. At this time, the working mode is that the synchronous switchQ2 is turned on, and the synchronous switchQ2 continues to flow; when the inductor current iL is at the first Between the preset threshold ith_l and the second preset threshold ith_h , that is, when ith_l ≤ iL ≤ ith_h , ith_l <0 < ith_h , then it is judged that the instantaneous current flow direction and operating mode are the same as the previous instant, Frequent turning on and off of power devices is avoided.
迟滞比较器对应两个门限电流ith_l和ith_h，输入单方向变化时，输出只跳变一次，输入由大变小时，对应小的门限电流；输入由小变大时，对应大的门限电流；在两个门限电流之间，输出保持原来的输出。The hysteresis comparator corresponds to two threshold currents ith_l and ith_h . When the input changes in one direction, the output only jumps once. When the input changes from large to small, it corresponds to a small threshold current; when the input changes from small to large, it corresponds to a large threshold current. ;Between the two threshold currents, the output maintains its original output.
双向DCDC变流器包括主功率管Q1、同步功率管Q2、储能电感L和控制电路，控制电路主要用于信号处理、产生PWM信号，控制主功率管Q1和同步开关管导通。控制电路包括：主控芯片、电感电流检测电路和信号处理电路。信号处理电路包括电流阈值比较电路、逻辑变量计数电路、逻辑输出电路和信号调制电路。The bidirectional DCDC converter includes a main power tube Q1 , a synchronous power tube Q2 , an energy storage inductor L and a control circuit. The control circuit is mainly used for signal processing, generating PWM signals, and controlling the conduction of the main power tube Q1 and the synchronous switch tube. . The control circuit includes: main control chip, inductor current detection circuit and signal processing circuit. The signal processing circuit includes a current threshold comparison circuit, a logic variable counting circuit, a logic output circuit and a signal modulation circuit.
所述的步骤二中，首先根据双向DCDC变流器的电感电流判断瞬时电流流向，在时间宽度上获得逻辑变量加和，根据逻辑变量加和进行比较获得中间调制因子，进而在考虑双向DCDC变流器的工作模式的情况下，根据中间调制因子获得驱动信号调制因子。In the second step, first determine the instantaneous current flow direction based on the inductor current of the bidirectional DCDC converter, obtain the sum of logical variables in the time width, and compare the sum of logical variables to obtain the intermediate modulation factor, and then consider the bidirectional DCDC converter. In the case of the operating mode of the current converter, the driving signal modulation factor is obtained according to the intermediate modulation factor.
所述的逻辑变量加和εm具体如下：The details of the logical variable sum εm are as follows:
εm＝∑εεm =∑ε
其中，εm为逻辑变量加和，ε为逻辑变量；iL为双向DCDC变流器的电感电流；ith_l和ith_h分别为迟滞比较器的第一预设阈值和第二预设阈值。Among them, εm is the sum of logical variables, ε is a logical variable; iL is the inductor current of the bidirectional DCDC converter; ith_l and ith_h are the first preset threshold and the second preset threshold of the hysteresis comparator respectively.
电流阈值比较电路，用于比较电感电流与阈值电流，输出逻辑变量ε。逻辑变量计数电路，用于完成k次阈值比较结果的统计，并求得加和εm。The current threshold comparison circuit is used to compare the inductor current with the threshold current and output the logic variable ε. The logical variable counting circuit is used to complete the statistics of k threshold comparison results and obtain the sum εm .
所述的根据逻辑变量加和εm进行比较获得中间调制因子F0，具体如下：The intermediate modulation factor F0 is obtained by comparison based on the logical variable sum εm , as follows:
其中，F0为中间调制因子；εm为逻辑变量加和；k为时间因子，即计数次数。Among them, F0 is the intermediate modulation factor; εm is the sum of logical variables; k is the time factor, that is, the number of counts.
时间宽度需要连续计数k次，将k次计数内阈值比较结果的加和εm与k进行比较，判定电流流向，并得到中间调制因子F0。如果连续k次ith_l<iL，则判定电流流向为正，F0＝1；如果连续k次iL<ith_h则判定电流流向为负，F0＝0；否则，认为电流流向未改变，F0＝F0。The time width requires k consecutive counts, and the sum εm of the threshold comparison results within k counts is compared with k to determine the current flow direction and obtain the intermediate modulation factor F0 . If ith_l <iL for k consecutive times, the current flow direction is determined to be positive, F0 =1; if iL <ith_h for k consecutive times, the current flow direction is determined to be negative, F0 =0; otherwise, the current flow direction is considered to have not changed. , F0 =F0 .
当完成k次计数之后，将εm清零。k的大小取决于谐波噪声干扰情况以及采样周期Ts。After k times of counting are completed, εm is cleared to zero. The size of k depends on the harmonic noise interference situation and the sampling period Ts .
所述的在考虑双向DCDC变流器的工作模式的情况下，根据中间调制因子获得驱动信号调制因子，具体为将中间调制因子F0与工作模式因子f0做“或”运算，得到驱动信号调制因子F：When considering the working mode of the bidirectional DCDC converter, the driving signal modulation factor is obtained according to the intermediate modulation factor. Specifically, the driving signal is obtained by performing an "OR" operation on the intermediate modulation factor F0 and the working mode factor f0 Modulation factor F:
其中，F为驱动信号调制因子；F0为中间调制因子。Among them, F is the driving signal modulation factor; F0 is the intermediate modulation factor.
双向DCDC变流器的工作模式包括升压Boost工作模式和降压Buck工作模式。The working modes of the bidirectional DCDC converter include boost working mode and buck working mode.
逻辑输出电路，用于完成k次逻辑变量加和εm与±k的比较，并输出调制因子F。Logic output circuit is used to complete the comparison of k times of logical variable addition εm and ±k, and output the modulation factor F.
所述的工作模式因子f0根据双向DCDC变流器的工作模式确定，具体如下：The working mode factor f0 is determined according to the working mode of the bidirectional DCDC converter, as follows:
其中，f0为工作模式因子。Among them, f0 is the working mode factor.
当电路处于升压Boost工作模式时，Q2在电感电流为负向时需要导通，因此调制信号应该考虑电路工作模式。当电路处于Boost模式时，定义模式因子f0＝1；当电路处于Buck模式时，定义模式因子f0＝0。When the circuit is in the boost operating mode, Q2 needs to be turned on when the inductor current is negative, so the modulation signal should consider the circuit operating mode. When the circuit is in Boost mode, the mode factor f0 =1 is defined; when the circuit is in Buck mode, the mode factor f0 =0 is defined.
当电感电流纹波较大时，为避免误触发，只能扩大阈值范围，从而导致电感电流通过D2续流时间较长，造成损耗增大。为解决这一问题，提出采用电流幅值和时间宽度双重标准调制同步整流脉冲，控制同步开关管的开通关断，可以降低阈值宽度，进而减小系统损耗。When the inductor current ripple is large, in order to avoid false triggering, the threshold range can only be expanded, resulting in a longer freewheeling time of the inductor current throughD2 , resulting in increased losses. In order to solve this problem, it is proposed to use dual standards of current amplitude and time width to modulate synchronous rectification pulses and control the turning on and off of synchronous switching tubes, which can reduce the threshold width and thereby reduce system losses.
所述的步骤三中根据双向DCDC变流器的驱动信号调制因子对双向DCDC变流器的驱动信号进行调制，具体如下：In the third step, the driving signal of the bidirectional DCDC converter is modulated according to the driving signal modulation factor of the bidirectional DCDC converter, as follows:
其中，Vgs2为双向DCDC变流器的驱动信号，即双向DCDC变流器的同步功率管的驱动脉冲；PWM2为双向DCDC变流器的主控芯片输出同步整流脉冲；F为驱动信号调制因子。Among them, Vgs2 is the drive signal of the bidirectional DCDC converter, that is, the drive pulse of the synchronous power tube of the bidirectional DCDC converter; PWM2 is the synchronous rectification pulse output by the main control chip of the bidirectional DCDC converter; F is the drive signal modulation factor.
信号调制电路，用于获得同步功率管的驱动脉冲Vgs2，具体为驱动信号调制因子F与主控芯片输出同步整流脉冲PWM2做逻辑“与”运算，得到双向DCDC变流器的驱动信号，并进行调制。The signal modulation circuit is used to obtain the driving pulse Vgs2 of the synchronous power tube. Specifically, the driving signal modulation factor F and the main control chip output synchronous rectification pulse PWM2 are logically ANDed to obtain the driving signal of the bidirectional DCDC converter, and Make adjustments.
在Buck模式下，只有当电感电流大于0且上管关闭时，同步开关管才导通，即Vgs2＝1，即只有当滤波电感需要通过同步开关续流时，同步开关管导通。Boost模式不受控制电路影响，即最大限度保证了正常工作不受干扰，又能够降低系统损耗。In Buck mode, the synchronous switch is turned on only when the inductor current is greater than 0 and the upper transistor is turned off, that is, Vgs2 =1. That is, the synchronous switch is turned on only when the filter inductor needs to freewheel through the synchronous switch. Boost mode is not affected by the control circuit, which ensures normal operation without interference to the greatest extent and reduces system losses.
本发明的有益效果是：The beneficial effects of the present invention are:
1)本发明解决了双向DCDC变换器工作于强制连续模式的问题。1) The present invention solves the problem of the bidirectional DCDC converter operating in the forced continuous mode.
2)本发明采用迟滞比较器做电流过零判别，消除了噪声干扰的影响。2) The present invention uses a hysteresis comparator for current zero-crossing determination, eliminating the influence of noise interference.
3)本发明采用电流幅值和时间宽度双重标准调制同步开关信号，缩小了迟滞比较阈值范围，降低了系统损耗。3) The present invention uses dual standards of current amplitude and time width to modulate the synchronous switching signal, narrowing the hysteresis comparison threshold range and reducing system losses.
4)本发明全部逻辑判断电路均可采用硬件电路搭建，电路简单、成本低廉、提升快速性和可靠性；由于控制逻辑清晰简洁，针对响应速度要求不高的系统，又可以由软件数字化实现控制，计算量较低。4) All logical judgment circuits of the present invention can be built using hardware circuits, which are simple, low-cost, and improve speed and reliability; because the control logic is clear and concise, for systems that do not require high response speed, they can also be controlled digitally by software , the computational complexity is low.
总之，本发明方法将电感电流作为控制信号，调制同步开关管驱动脉冲，解决双向DCDC变流电路工作在强制连续模式的问题，从而降低磁损耗和直流电阻损耗。针对电路中高噪声信号干扰，采用迟滞比较的方式进行净化；同时，采用电流幅值和时间宽度双重标准调制同步整流脉冲，降低阈值宽度，进一步降低系统损耗。控制逻辑可以由硬件电路实现，保障了控制的快速性和可靠性。In short, the method of the present invention uses the inductor current as a control signal to modulate the synchronous switch tube drive pulse, thereby solving the problem of the bidirectional DCDC converter circuit working in the forced continuous mode, thereby reducing the magnetic loss and DC resistance loss. Aiming at interference from high-noise signals in the circuit, hysteresis comparison is used for purification; at the same time, dual standards of current amplitude and time width are used to modulate synchronous rectification pulses to reduce the threshold width and further reduce system losses. The control logic can be implemented by hardware circuits, ensuring the speed and reliability of control.
附图说明Description of the drawings
图1为本发明方法所涉及的电动汽车充电机拓扑结构；Figure 1 shows the topological structure of the electric vehicle charger involved in the method of the present invention;
图2为本发明方法所涉及的双向DCDC变换器及其高效能控制电路；Figure 2 shows the bidirectional DCDC converter and its high-efficiency control circuit involved in the method of the present invention;
图3为本发明方法工作电流与传统控制方法工作时的工作波形，其中，图3的(a)为图2中本发明方法和传统控制方法工作时Q1的驱动信号，图3的(b)为图2中传统控制方法工作时Q2的驱动信号，图3的(c)为图2中本发明方法工作时Q2的驱动信号，图3的(d)为本发明方法工作时的电感电流，图3的(e)为传统控制方法工作时的电感电流；Figure 3 is the operating waveform of the operating current of the method of the present invention and the traditional control method. Figure 3(a) is the driving signal ofQ1 when the method of the present invention and the traditional control method are working in Figure 2. Figure 3(b) ) is the driving signal of Q2 when the traditional control method is working in Figure 2, (c) of Figure 3 is the driving signal of Q 2 when the method of the present invention is working in Figure 2, and (d) of Figure 3 is the driving signal of Q2 when the method of the present invention is working. Inductor current, (e) in Figure 3 shows the inductor current when the traditional control method is working;
图4为本发明方法控制电路框图；Figure 4 is a block diagram of the control circuit of the method of the present invention;
图5为本发明方法控制电路工作流程图。Figure 5 is a working flow chart of the control circuit of the method of the present invention.
具体实施方式Detailed ways
下面结合附图及具体实施例对本发明作进一步详细说明。The present invention will be further described in detail below in conjunction with the accompanying drawings and specific embodiments.
如图4和图5所示，本发明的高效能双向DCDC变流器控制方法，包括：As shown in Figures 4 and 5, the high-efficiency bidirectional DCDC converter control method of the present invention includes:
步骤一，根据双向DCDC变流器的电感电流判断瞬时电流流向。Step 1: Determine the instantaneous current flow direction based on the inductor current of the bidirectional DCDC converter.
步骤一中，使用基于迟滞比较器的电流阈值比较进行瞬时电流流向的判断，有利于消除纹波、抖动等干扰的影响；当电感电流iL小于迟滞比较器的第一预设阈值ith_l时，判断瞬时电流流向为负，此时工作模式为双向DCDC变流器的同步开关管Q2关断，由同步开关管Q2的反并联二极管D2续流；当电感电流iL大于迟滞比较器的第二预设阈值ith_h时，判断瞬时电流流向为正，此时工作模式为同步开关管Q2开通，由同步开关管Q2续流；当电感电流iL在第一预设阈值ith_l和第二预设阈值ith_h之间，即ith_l≤iL≤ith_h时，ith_l<0<ith_h，则判断瞬时电流流向及工作模式与上一瞬时的相同，避免了功率器件的频繁开通、关断。In step one, the current threshold comparison based on the hysteresis comparator is used to determine the instantaneous current flow direction, which is beneficial to eliminating the effects of interference such as ripple and jitter; when the inductor current iL is less than the first preset threshold ith_l of the hysteresis comparator , it is judged that the instantaneous current flow direction is negative. At this time, the working mode isthat the synchronous switch Q2 of the bidirectional DCDC converter is turned off, and the anti-parallel diode D2 of the synchronous switch Q2 freewheels; when the inductor current i Lisgreaterthan the hysteresis comparison When the second preset threshold ith_h of the inductor is determined, the instantaneous current flow direction is positive. At this time, the working mode is that the synchronous switchQ2 is turned on, and the synchronous switchQ2 continues to flow; when the inductor current iL is at the first preset threshold Between ith_l and the second preset threshold ith_h , that is, when ith_l ≤ iL ≤ ith_h , ith_l <0 < ith_h , then it is judged that the instantaneous current flow direction and operating mode are the same as the previous instant, avoiding the power Frequent turning on and off of devices.
迟滞比较器对应两个门限电流ith_l和ith_h，输入单方向变化时，输出只跳变一次，输入由大变小时，对应小的门限电流；输入由小变大时，对应大的门限电流；在两个门限电流之间，输出保持原来的输出。The hysteresis comparator corresponds to two threshold currents ith_l and ith_h . When the input changes in one direction, the output only jumps once. When the input changes from large to small, it corresponds to a small threshold current; when the input changes from small to large, it corresponds to a large threshold current. ;Between the two threshold currents, the output maintains its original output.
双向DCDC变流器包括主功率管Q1、同步功率管Q2、储能电感L和控制电路，控制电路主要用于信号处理、产生PWM信号，控制主功率管Q1和同步开关管导通。控制电路包括：主控芯片、电感电流检测电路和信号处理电路。信号处理电路包括电流阈值比较电路、逻辑变量计数电路、逻辑输出电路和信号调制电路。The bidirectional DCDC converter includes a main power tube Q1 , a synchronous power tube Q2 , an energy storage inductor L and a control circuit. The control circuit is mainly used for signal processing, generating PWM signals, and controlling the conduction of the main power tube Q1 and the synchronous switch tube. . The control circuit includes: main control chip, inductor current detection circuit and signal processing circuit. The signal processing circuit includes a current threshold comparison circuit, a logic variable counting circuit, a logic output circuit and a signal modulation circuit.
步骤二，根据双向DCDC变流器的瞬时电流流向，在考虑双向DCDC变流器的工作模式的情况下，获得双向DCDC变流器在时间宽度上的驱动信号调制因子。Step 2: According to the instantaneous current flow direction of the bidirectional DCDC converter, and considering the operating mode of the bidirectional DCDC converter, obtain the drive signal modulation factor of the bidirectional DCDC converter in the time width.
步骤二中，首先根据双向DCDC变流器的电感电流判断瞬时电流流向，在时间宽度上获得逻辑变量加和，根据逻辑变量加和进行比较获得中间调制因子，进而在考虑双向DCDC变流器的工作模式的情况下，根据中间调制因子获得驱动信号调制因子。In step two, first determine the instantaneous current flow direction based on the inductor current of the bidirectional DCDC converter, obtain the sum of logical variables in the time width, and compare based on the sum of logical variables to obtain the intermediate modulation factor, and then consider the bidirectional DCDC converter. In the case of working mode, the driving signal modulation factor is obtained according to the intermediate modulation factor.
逻辑变量加和εm具体如下：The details of the logical variable sum εm are as follows:
εm＝∑εεm =∑ε
其中，εm为逻辑变量加和，ε为逻辑变量；iL为双向DCDC变流器的电感电流；ith_l和ith_h分别为迟滞比较器的第一预设阈值和第二预设阈值。Among them, εm is the sum of logical variables, ε is a logical variable; iL is the inductor current of the bidirectional DCDC converter; ith_l and ith_h are the first preset threshold and the second preset threshold of the hysteresis comparator respectively.
电流阈值比较电路，用于比较电感电流与阈值电流，输出逻辑变量ε。逻辑变量计数电路，用于完成k次阈值比较结果的统计，并求得加和εm。The current threshold comparison circuit is used to compare the inductor current with the threshold current and output the logic variable ε. The logical variable counting circuit is used to complete the statistics of k threshold comparison results and obtain the sum εm .
根据逻辑变量加和εm进行比较获得中间调制因子F0，具体如下：The intermediate modulation factor F0 is obtained by comparison based on the logical variable sum εm , as follows:
其中，F0为中间调制因子；εm为逻辑变量加和；k为时间因子，即计数次数。Among them, F0 is the intermediate modulation factor; εm is the sum of logical variables; k is the time factor, that is, the number of counts.
时间宽度需要连续计数k次，将k次计数内阈值比较结果的加和εm与k进行比较，判定电流流向，并得到中间调制因子F0。如果连续k次ith_l<iL，则判定电流流向为正，F0＝1；如果连续k次iL<ith_h则判定电流流向为负，F0＝0；否则，认为电流流向未改变，F0＝F0。The time width requires k consecutive counts, and the sum εm of the threshold comparison results within k counts is compared with k to determine the current flow direction and obtain the intermediate modulation factor F0 . If ith_l <iL for k consecutive times, the current flow direction is determined to be positive, F0 =1; if iL <ith_h for k consecutive times, the current flow direction is determined to be negative, F0 =0; otherwise, the current flow direction is considered to have not changed. , F0 =F0 .
当完成k次计数之后，将εm清零。k的大小取决于谐波噪声干扰情况以及采样周期Ts。After k times of counting are completed, εm is cleared to zero. The size of k depends on the harmonic noise interference situation and the sampling period Ts .
在考虑双向DCDC变流器的工作模式的情况下，根据中间调制因子获得驱动信号调制因子，具体为将中间调制因子F0与工作模式因子f0做“或”运算，得到驱动信号调制因子F：Considering the working mode of the bidirectional DCDC converter, the driving signal modulation factor is obtained according to the intermediate modulation factor. Specifically, the driving signal modulation factor F is obtained by performing an "OR" operation on the intermediate modulation factor F0 and the working mode factor f0 :
其中，F为驱动信号调制因子；F0为中间调制因子；Among them, F is the driving signal modulation factor; F0 is the intermediate modulation factor;
双向DCDC变流器的工作模式包括升压Boost工作模式和降压Buck工作模式。The working modes of the bidirectional DCDC converter include boost working mode and buck working mode.
逻辑输出电路，用于完成k次逻辑变量加和εm与±k的比较，并输出调制因子F。Logic output circuit is used to complete the comparison of k times of logical variable addition εm and ±k, and output the modulation factor F.
工作模式因子f0根据双向DCDC变流器的工作模式确定，具体如下：The working mode factor f0 is determined according to the working mode of the bidirectional DCDC converter, as follows:
其中，f0为工作模式因子。Among them, f0 is the working mode factor.
当电路处于升压Boost工作模式时，Q2在电感电流为负向时需要导通，因此调制信号应该考虑电路工作模式。当电路处于Boost模式时，定义模式因子f0＝1；当电路处于Buck模式时，定义模式因子f0＝0。When the circuit is in the boost operating mode, Q2 needs to be turned on when the inductor current is negative, so the modulation signal should consider the circuit operating mode. When the circuit is in Boost mode, the mode factor f0 =1 is defined; when the circuit is in Buck mode, the mode factor f0 =0 is defined.
当电感电流纹波较大时，为避免误触发，只能扩大阈值范围，从而导致电感电流通过D2续流时间较长，造成损耗增大。为解决这一问题，提出采用电流幅值和时间宽度双重标准调制同步整流脉冲，控制同步开关管的开通关断，可以降低阈值宽度，进而减小系统损耗。When the inductor current ripple is large, in order to avoid false triggering, the threshold range can only be expanded, resulting in a longer freewheeling time of the inductor current throughD2 , resulting in increased losses. In order to solve this problem, it is proposed to use dual standards of current amplitude and time width to modulate synchronous rectification pulses and control the turning on and off of synchronous switching tubes, which can reduce the threshold width and thereby reduce system losses.
步骤三，根据双向DCDC变流器的驱动信号调制因子对双向DCDC变流器的驱动信号进行调制，实现双向DCDC变流器的控制。Step 3: Modulate the drive signal of the bidirectional DCDC converter according to the drive signal modulation factor of the bidirectional DCDC converter to realize control of the bidirectional DCDC converter.
步骤三中根据双向DCDC变流器的驱动信号调制因子对双向DCDC变流器的驱动信号进行调制，具体如下：In step three, the driving signal of the bidirectional DCDC converter is modulated according to the driving signal modulation factor of the bidirectional DCDC converter, as follows:
其中，Vgs2为双向DCDC变流器的驱动信号，即双向DCDC变流器的同步功率管的驱动脉冲；PWM2为双向DCDC变流器的主控芯片输出同步整流脉冲；F为驱动信号调制因子。Among them, Vgs2 is the drive signal of the bidirectional DCDC converter, that is, the drive pulse of the synchronous power tube of the bidirectional DCDC converter; PWM2 is the synchronous rectification pulse output by the main control chip of the bidirectional DCDC converter; F is the drive signal modulation factor.
信号调制电路，用于获得同步功率管的驱动脉冲Vgs2，具体为驱动信号调制因子F与主控芯片输出同步整流脉冲PWM2做逻辑“与”运算，得到双向DCDC变流器的驱动信号，并进行调制。The signal modulation circuit is used to obtain the driving pulse Vgs2 of the synchronous power tube. Specifically, the driving signal modulation factor F and the main control chip output synchronous rectification pulse PWM2 are logically ANDed to obtain the driving signal of the bidirectional DCDC converter, and Make adjustments.
在Buck模式下，只有当电感电流大于0且上管关闭时，同步开关管才导通，即Vgs2＝1，即只有当滤波电感需要通过同步开关续流时，同步开关管导通。Boost模式不受控制电路影响，即最大限度保证了正常工作不受干扰，又能够降低系统损耗。In Buck mode, the synchronous switch is turned on only when the inductor current is greater than 0 and the upper transistor is turned off, that is, Vgs2 =1. That is, the synchronous switch is turned on only when the filter inductor needs to freewheel through the synchronous switch. Boost mode is not affected by the control circuit, which ensures normal operation without interference to the greatest extent and reduces system losses.
本发明的具体实施方式如下：The specific implementation of the present invention is as follows:
如图1所示，为本发明所涉及的电动汽车充电机拓扑结构图，包括电网、AC/DC变换器，电池、逆变器、电机、双向DCDC变流器和超级电容，双向DCDC变流器的两端分别为电池和超级电容，超级电容用于向电池供电。As shown in Figure 1, it is a topological structure diagram of the electric vehicle charger involved in the present invention, including a power grid, an AC/DC converter, a battery, an inverter, a motor, a bidirectional DCDC converter and a supercapacitor. The bidirectional DCDC converter The two ends of the device are the battery and the supercapacitor, and the supercapacitor is used to supply power to the battery.
如图2所示，当超级电容向电池供电时，如果工作在轻载工况下，双向DCDC变流器将会工作在强制励磁模式：每个周期电流降为零之后反向流通，产生额外损耗、影响变流器寿命。As shown in Figure 2, when the supercapacitor supplies power to the battery, if it works under light load conditions, the bidirectional DCDC converter will work in the forced excitation mode: the current will flow in the reverse direction after falling to zero in each cycle, resulting in additional loss, affecting the life of the converter.
为了证明本发明方法的有效性，搭建了可控制充电系统测试平台。测试平台拓扑如图2所示，由锂电池、超级电容和同步Buck变流器组成，采用电流环的方式，对PWM控制脉冲进行调制，控制器采用结构简单、易于实现的PI控制器。在该平台上测试了本发明方法的控制性能。被测试的充电系统参数如表1所示。In order to prove the effectiveness of the method of the present invention, a controllable charging system test platform was built. The test platform topology is shown in Figure 2. It consists of lithium batteries, supercapacitors and synchronous Buck converters. It uses a current loop to modulate PWM control pulses. The controller uses a PI controller with a simple structure and easy to implement. The control performance of the method of the invention was tested on this platform. The tested charging system parameters are shown in Table 1.
表1Table 1
如图3(a)、图3的(b)、图3的(c)、图3的(d)和图3的(e)所示，比较了本发明方法和传统控制方案及控制结果的不同，其中，VGS1和VGS2分别为Q1和Q2的驱动信号，Ton和Toff分别为Q1和Q2的导通和关断时间，△t为死区时间；为了证明本发明的优势，比较了本发明方法和传统控制方案的工作效率，结果如表2所示。As shown in Figure 3(a), Figure 3(b), Figure 3(c), Figure 3(d) and Figure 3(e), the method of the present invention is compared with the traditional control scheme and control results. Different, among them, VGS1 and VGS2 are the driving signals of Q1 and Q2 respectively, Ton and Toff are the turn-on and turn-off times of Q1 and Q2 respectively, △t is the dead time; in order to prove this The advantages of the invention compare the working efficiency of the method of the invention and the traditional control scheme. The results are shown in Table 2.
表2控制方案比较Table 2 Comparison of control schemes
由表2可知，相对比传统控制方法，本发明方法可以在保证输出电压稳定的前提下，提升工作效率，验证了发明方法的有效性。It can be seen from Table 2 that compared with the traditional control method, the inventive method can improve the work efficiency while ensuring the stability of the output voltage, which verifies the effectiveness of the inventive method.
本发明实施例对各器件的型号除做特殊说明的以外，其他器件的型号不做限制，只要能完成上述功能的器件均可。本领域技术人员可以理解附图只是一个优选实施例的示意图，上述本发明实施例序号仅仅为了描述，不代表实施例的优劣。本发明并不限于上文描述的实施方式。以上对具体实施方式的描述旨在描述和说明本发明的技术方案，上述的具体实施方式仅仅是示意性的，并不是限制性的。在不脱离本发明宗旨和权利要求所保护的范围情况下，本领域的普通技术人员在本发明的启示下还可做出很多形式的具体变换，这些均属于本发明的保护范围之内。The embodiments of the present invention do not limit the models of each device unless otherwise specified, as long as the devices can complete the above functions. Those skilled in the art can understand that the accompanying drawing is only a schematic diagram of a preferred embodiment, and the above-mentioned serial numbers of the embodiments of the present invention are only for description and do not represent the advantages and disadvantages of the embodiments. The invention is not limited to the embodiments described above. The above description of the specific embodiments is intended to describe and illustrate the technical solution of the present invention. The above specific embodiments are only illustrative and not restrictive. Without departing from the spirit of the present invention and the scope protected by the claims, those of ordinary skill in the art can make many specific changes based on the inspiration of the present invention, and these all fall within the protection scope of the present invention.
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| CN202310938200.0ACN116979809B (en) | 2023-07-27 | 2023-07-27 | A high-efficiency bidirectional DCDC converter control method |
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| US20160276945A1 (en)* | 2015-03-16 | 2016-09-22 | Delta Electronics, Inc. | Power converter and controlling method thereof |
| CN214380664U (en)* | 2021-02-25 | 2021-10-08 | 中国电子科技集团公司第五十八研究所 | An Adaptive Power Consumption Optimization Circuit with Interlocking Function |
| CN113809920A (en)* | 2021-08-19 | 2021-12-17 | 广州金升阳科技有限公司 | BUCK converter control method |
| CN113872443A (en)* | 2021-11-05 | 2021-12-31 | 上海杰瑞兆新信息科技有限公司 | Negative current control method and circuit for synchronous rectification converter |
| Publication number | Priority date | Publication date | Assignee | Title |
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| US20160276945A1 (en)* | 2015-03-16 | 2016-09-22 | Delta Electronics, Inc. | Power converter and controlling method thereof |
| CN214380664U (en)* | 2021-02-25 | 2021-10-08 | 中国电子科技集团公司第五十八研究所 | An Adaptive Power Consumption Optimization Circuit with Interlocking Function |
| CN113809920A (en)* | 2021-08-19 | 2021-12-17 | 广州金升阳科技有限公司 | BUCK converter control method |
| WO2023020407A1 (en)* | 2021-08-19 | 2023-02-23 | 广州金升阳科技有限公司 | Control method and control circuit for buck converter |
| CN113872443A (en)* | 2021-11-05 | 2021-12-31 | 上海杰瑞兆新信息科技有限公司 | Negative current control method and circuit for synchronous rectification converter |
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