CN111030479B - Active-clamp flyback power converter and associated control method - Google Patents
Active-clamp flyback power converter and associated control methodDownload PDFInfo
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- CN111030479B CN111030479BCN201811180757.8ACN201811180757ACN111030479BCN 111030479 BCN111030479 BCN 111030479BCN 201811180757 ACN201811180757 ACN 201811180757ACN 111030479 BCN111030479 BCN 111030479B
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- 238000000034methodMethods0.000titleclaimsabstractdescription25
- 238000004804windingMethods0.000claimsabstractdescription54
- 239000003990capacitorSubstances0.000claimsabstractdescription18
- 238000001514detection methodMethods0.000claimsdescription27
- 230000005347demagnetizationEffects0.000claimsdescription10
- 230000000903blocking effectEffects0.000claimsdescription5
- 239000013256coordination polymerSubstances0.000description15
- 238000006243chemical reactionMethods0.000description9
- 230000010355oscillationEffects0.000description9
- 230000008859changeEffects0.000description7
- 230000000873masking effectEffects0.000description7
- 230000009467reductionEffects0.000description7
- 230000000295complement effectEffects0.000description6
- 230000001965increasing effectEffects0.000description5
- 230000004044responseEffects0.000description4
- 238000012360testing methodMethods0.000description3
- 230000001052transient effectEffects0.000description3
- 238000010521absorption reactionMethods0.000description2
- 230000008901benefitEffects0.000description2
- 230000007423decreaseEffects0.000description2
- 238000013459approachMethods0.000description1
- 238000013461designMethods0.000description1
- 230000006698inductionEffects0.000description1
- 230000001939inductive effectEffects0.000description1
- 238000002955isolationMethods0.000description1
- 230000005415magnetizationEffects0.000description1
- 238000004519manufacturing processMethods0.000description1
- 238000012986modificationMethods0.000description1
- 230000004048modificationEffects0.000description1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
- H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
- H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
- H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M3/00—Conversion of DC power input into DC power output
- H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
- H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
- H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
- H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
- H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
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- Power Engineering (AREA)
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Abstract
Translated fromChinese
Description
Translated fromChinese技术领域technical field
本发明大致关于有源钳位反激式电源转换器以及相关的控制方法，尤其是关于可以具有无能量损耗的有源钳位电路的有源钳位反激式电源转换器的相关技术。The present invention generally relates to active clamp flyback power converters and related control methods, and more particularly to the related art of active clamp flyback power converters that can have active clamp circuits with no energy loss.
背景技术Background technique
反激式电源转换器(flyback power converter)已经是广泛的使用于许多电子产品的电源供应器中，举例来说，像是家电、计算机、电池充电器等等。为了提升电能转换效能，有源钳位(active-clamp)电路用来改善了一般反激式电源转换器的缓冲器(snubber)的能量损耗的问题。一般具有有源钳位电路的反激式电源转换器，称为有源钳位反激式(active clamp flyback，ACF)电源转换器。在重载情形下，ACF电源转换器往往在电能转换效能上表现优异。但是在轻载情形下，ACF电源转换器却往往因为主绕组中的循环电流(circulated current)，具有高的电能损失。Flyback power converters have been widely used in power supplies of many electronic products, such as home appliances, computers, battery chargers, and so on. In order to improve the power conversion performance, an active-clamp circuit is used to improve the energy loss of a snubber of a general flyback power converter. Generally, a flyback power converter with an active clamp circuit is called an active clamp flyback (active clamp flyback, ACF) power converter. Under heavy load conditions, ACF power converters often excel in power conversion efficiency. However, under light load conditions, ACF power converters tend to have high power losses due to the circulating current in the main winding.
德州仪器提供了以UCC28780的反激控制器所控制的反激式电源转换器。UCC28780可以随着负载大小，切换于四种操作模式之中，只是，从UCC28780的规格书(datasheet)中可以发现，UCC28780的系统应用电路中，有源钳位电路中还是需要一个泄流电阻(bleederresistor)，来将有源钳位电路中电容所存放的电能，缓慢的释放掉。显然，UCC28780并没有完全发挥有源钳位电路的优点。因为泄流电阻的存在，德州仪器所提供的有源钳位电路还是会有能量的损耗。Texas Instruments offers a flyback power converter controlled by the flyback controller of the UCC28780. UCC28780 can switch between four operating modes according to the load size. However, it can be found from the datasheet of UCC28780 that in the system application circuit of UCC28780, a bleeder resistor ( bleederresistor), to slowly release the energy stored in the capacitor in the active clamp circuit. Obviously, the UCC28780 does not take full advantage of the active clamp circuit. Because of the existence of the bleeder resistor, the active clamp circuit provided by Texas Instruments will still have energy loss.
而且，习知的ACF电源转换器，在系统设计上，也往往困扰于电磁干扰问题(electromagnetic interference，EMI)以及/或是杂音(audible noise)的问题。Moreover, the conventional ACF power converters are often plagued by electromagnetic interference (EMI) and/or audible noise issues in system design.
发明内容SUMMARY OF THE INVENTION
依据本发明所实施的一种控制方法，适用于一种有源钳位反激式电源转换器。该有源钳位反激式电源转换器包含有一有源钳位电路，其与一变压器的一主绕组并联。该有源钳位电路包含有串联的一上臂开关与一电容。该有源钳位反激式电源转换器并包含有一下臂开关，其连接该主绕组至一第一电源线。该控制方法包含有：开关一下臂开关，产生N个连续开关周期，其中，N为大于1的整数，至少该第N个开关周期为改良式反激周期，其他为正常反激周期；使每一开关周期不小于一遮蔽时间，其中，该遮蔽时间系依据该有源钳位反激式电源转换器的一负载而产生；于每一正常反激周期中，固定维持该上臂开关关闭；以及，于每一改良式反激周期中，在该遮蔽时间之后，开启该上臂开关，产生一上臂开启时间，以使该下臂开关进行零电压切换。A control method implemented according to the present invention is suitable for an active clamp flyback power converter. The active clamp flyback power converter includes an active clamp circuit connected in parallel with a main winding of a transformer. The active clamp circuit includes an upper arm switch and a capacitor connected in series. The active clamp flyback power converter includes a lower arm switch which connects the main winding to a first power line. The control method includes: switching the lower arm switch to generate N continuous switching cycles, wherein N is an integer greater than 1, at least the Nth switching cycle is an improved flyback cycle, and the others are normal flyback cycles; A switching period is not less than a shadowing time, wherein the shadowing time is generated according to a load of the active clamp flyback power converter; in each normal flyback cycle, the upper arm switch is kept off; and , in each improved flyback cycle, after the shielding time, the upper arm switch is turned on to generate an upper arm turn-on time, so that the lower arm switch performs zero-voltage switching.
本发明的实施例提供一种有源钳位反激式电源转换器，包含有一下臂开关、一上臂开关、以及一控制电路。该下臂开关连接一变压器的一主绕组至一第一电源线。该上臂开关与一电容串联以构成一有源钳位电路。该有源钳位电路并联于该主绕组。该控制电路架构来依据一补偿信号以及一电流检测信号，控制该上臂开关以及该下臂开关，用以调整该有源钳位反激式电源转换器的一输出电压。该控制电路可选择性地操作于数个操作模式中的一个。该等操作模式包含有一反激模式。当该控制电路操作于该反激模式时，该控制电路开关该下臂开关，产生数个开关周期，包含有一改良式反激周期以及一正常反激周期。于每一正常反激周期中，该控制电路使该上臂开关维持关闭。于每一改良式反激周期中，该控制电路使该上臂开关于一遮蔽时间后开启，产生一上臂开启时间，以使该下臂开关进行零电压切换。该控制电路系依据该有源钳位反激式电源转换器的一负载，产生该遮蔽时间。Embodiments of the present invention provide an active clamp flyback power converter, which includes a lower arm switch, an upper arm switch, and a control circuit. The lower arm switch connects a main winding of a transformer to a first power line. The upper arm switch is connected in series with a capacitor to form an active clamp circuit. The active clamp circuit is connected in parallel with the main winding. The control circuit structure controls the upper arm switch and the lower arm switch according to a compensation signal and a current detection signal to adjust an output voltage of the active clamp flyback power converter. The control circuit is selectively operable in one of several modes of operation. The operating modes include a flyback mode. When the control circuit operates in the flyback mode, the control circuit switches the lower arm switch to generate several switching cycles, including an improved flyback cycle and a normal flyback cycle. During each normal flyback cycle, the control circuit keeps the upper arm switch off. In each improved flyback cycle, the control circuit enables the upper arm switch to be turned on after a shielding time to generate an upper arm turn-on time so that the lower arm switch performs zero-voltage switching. The control circuit generates the blocking time according to a load of the active clamp flyback power converter.
附图说明Description of drawings
图1为依据本发明所实施的ACF电源转换器10。FIG. 1 is an
图2举例显示ACF模式以及反激模式。Figure 2 shows an example of ACF mode and flyback mode.
图3A为ACF电源转换器10操作于ACF模式时的一些信号的波形。FIG. 3A shows the waveforms of some signals when the ACF
图3B为ACF电源转换器10操作于反激模式时的一些信号的波形。FIG. 3B shows the waveforms of some signals when the ACF
图4放大说明图3A中，下臂开启时间TON-L内的电流检测电压VCS的波形。FIG. 4 is an enlarged illustration of the waveform of the current detection voltage VCS in the lower arm turn-on time TON-L in FIG. 3A .
图5A显示图1的实施例中，开关频率fCYC与补偿电压VCOMP的关系。FIG. 5A shows the relationship between the switching frequency fCYC and the compensation voltage VCOMP in the embodiment of FIG. 1 .
图5B显示图1的实施例中，峰值VCS-PEAK与补偿电压VCOMP的关系。FIG. 5B shows the relationship between the peak value VCS-PEAK and the compensation voltage VCOMP in the embodiment of FIG. 1 .
图5C显示图1的实施例中，处于稳态时，输出电流IO与补偿电压VCOMP的关系，以及ACF模式与反激模式之间的切换。FIG. 5C shows the relationship between the output currentIO and the compensation voltage VCOMP and the switching between the ACF mode and the flyback mode in the steady state in the embodiment of FIG. 1 .
图6显示图1的ACF电源转换器10操作于反激模式时的数个连续的开关周期TCYC。FIG. 6 shows several consecutive switching cycles TCYC when the
图7为电源控制器14中所使用的控制方法60。FIG. 7 is a
具体实施方式Detailed ways
在本说明书中，有一些相同的符号，其表示具有相同或是类似的结构、功能、原理的组件，且为业界具有一般知识能力者可以依据本说明书的教导而推知。为说明书的简洁度考虑，相同的符号的组件将不再重述。In this specification, there are some identical symbols, which represent components having the same or similar structures, functions, and principles, and those with ordinary knowledge in the industry can infer them according to the teachings of this specification. For the sake of brevity of the description, the components of the same symbols will not be repeated.
图1为依据本发明所实施的ACF电源转换器10。桥式整流器BD将交流市电VAC整流，提供输入电源线IN以及输入地电源线GNDI。输入电压VIN于输入电源线IN上。变压器TF包含有主绕组LP、二次侧绕组LS以及辅助绕组LA，彼此电感性地耦接在一起。主绕组LP、下臂开关LSS、以及电流检测电阻RCS相串联于输入电源线IN与输入地电源线GNDI之间。下臂开关LSS以及电流检测电阻RCS连接主绕组LP至输入地电源线GNDI。通过电流检测接脚CS，电流检测电阻RCS提供电流检测电压VCS给电源控制器14。上臂开关HSS与电容CAC相串联，构成有源钳位电路ACC。有源钳位电路ACC跟主绕组LP相并联。当下臂开关LSS导通时，电流检测电压VCS可以代表流经主绕组LP的绕组电流IM。FIG. 1 is an
电源控制器14，其可以是一集成电路，通过接脚HD与LD，来控制驱动器DVR。驱动器DVR可以是另一个集成电路，提供上臂信号DRVHS与下臂信号DRVLS，分别控制上臂开关HSS与下臂开关LSS。上臂开关HSS与下臂开关LSS可以是耐高压的GaN晶体管或是MOS晶体管。在另一个实施例中，驱动器DVR、上臂开关HSS与下臂开关LSS全部整合在一个封装好的集成电路中。电源控制器14与驱动器DVR一起可以视为一控制电路，提供上臂信号DRVHS与下臂信号DRVLS，分别控制上臂开关HSS与下臂开关LSS。The
电源控制器14通过上臂开关HSS与下臂开关LSS的开关，使得绕组电流IM产生变化，也使得位于二次侧的二次侧绕组LS通过电感性感应，产生了交流电压/电流。整流二次侧绕组LS上的交流电压/电流可以提供输出电源线OUT以及输出地电源线GNDO。输出电源线OUT上的输出电压VOUT可以用来对负载13供电，而流经负载13的电流为输出电流IO。举例来说，负载13是一可充电式电池。The
为了调整供应给负载13的输出电压VOUT，误差放大器EA、光耦合器OPT、以及补偿电容CCOMP一起，提供负回馈控制给予电源控制器14。位于二次侧的误差放大器EA比较输出电压VOUT与目标电压VREF-TAR，通过提供直流隔绝的光耦合器OPT，来控制产生补偿电容CCOMP上的补偿电压VCOMP。举例来说，当输出电压VOUT高过目标电压VREF-TAR时，补偿电压VCOMP下降，ACF电源转换器10转换给负载13的电能将变少，目标是使输出电压VOUT大约维持在目标电压VREF-TAR附近。In order to adjust the output voltage VOUT supplied to the
位于一次侧的辅助绕组LA上的交流电压/电流，经过整流后，产生操作电源VCC，其连接在电源控制器14的电源接脚VCC上，大致提供电源控制器14所需要的操作电能。电阻RA与RB相串联，构成一个分压电路，其与辅助绕组LA并联。电阻RA与RB之间的连接点则连接到电源控制器14的回馈接脚FB，其上具有回馈电压VFB。The AC voltage/current on the auxiliary winding LA on the primary side is rectified to generate an operating power VCC , which is connected to the power pin VCC of the
电源控制器14与驱动器DVR，以电流检测电压VCS、补偿电压VCOMP、以及回馈电压VFB作为输入，来产生上臂信号DRVHS与下臂信号DRVLS。The
在一实施例中，电源控制器14选择性的切换操作于两个操作模式，只是本发明并不限于此。在另一实施例中，电源控制器14可以选择性的切换于三个或以上的操作模式。图2举例显示两个操作模式，以下称为ACF模式以及反激(flyback)模式。大致上来说，ACF模式大约适用于负载13为重载或是中载的状态，而反激模式大致适用于负载13为中载或是轻载的状态。In one embodiment, the
如同图2所示，当操作于ACF模式时，电源控制器14：1)使得上臂信号DRVHS与下臂信号DRVLS大致为互补，并进行零电压切换(zero-voltage switching，ZVS)；2)大约固定开关频率fCYC，但有抖频(jittering)；以及，3)依据补偿电压VCOMP调变峰值VCS-PEAK。峰值VCS-PEAK代表电流检测电压VCS的局部最大值，稍后将详细解释。当操作于ACF模式时，电源控制器14检查依据电流检测电压VCS所定义的正电流时间TON-P与负电流时间TON-N是否符合一预定关系，来判断是否脱离ACF模式，进入反激模式。正电流时间TON-P与负电流时间TON-N表示当下臂开关LSS导通时，电流检测电压VCS分别为正与为负的时间。As shown in FIG. 2, when operating in the ACF mode, the power controller 14: 1) makes the upper arm signal DRVHS and the lower arm signal DRVLS substantially complementary, and performs zero-voltage switching (ZVS); 2 ) about a fixed switching frequency fCYC , but with jittering; and, 3) modulate the peak value VCS-PEAK according to the compensation voltage VCOMP . The peak value VCS-PEAK represents the local maximum value of the current sense voltage VCS , which will be explained in detail later. When operating in the ACF mode, the
当操作于ACF模式时，相较于补偿电压VCOMP，由正电流时间TON-P与负电流时间TON-N-所构成的预定关系，更能代表负载13的状态。When operating in the ACF mode, the predetermined relationship formed by the positive current time TON-P and the negative current time TON-N- can better represent the state of the
当操作于反激模式时，电源控制器14：1)大致使上臂开关HSS维持关闭；2)固定峰值VCS-PEAK；以及，3)依据补偿电压VCOMP调变开关频率fCYC，并加入抖频。同时，电源控制器14检查是否补偿电压VCOMP是否大于参考电压VCOMP-REF，来判断是否脱离反激模式，进入ACF模式。When operating in the flyback mode, the power supply controller 14: 1) keep the upper arm switch HSS substantially closed; 2) fix the peak value VCS-PEAK ; and, 3) modulate the switching frequency fCYC according to the compensation voltage VCOMP , and add jitter. At the same time, the
请同时参考图2与图3A。图3A为ACF电源转换器10操作于ACF模式时的一些信号的波形。由上到下，图3A中的波形分别是电源控制器14内部自己产生的频率信号CLK、上臂信号DRVHS、下臂信号DRVLS、电流检测电压VCS、位于上臂开关HSS与下臂开关LSS之间连接点上的开关电压VSW、位于辅助绕组LA上的绕组电压VAUX。Please refer to FIG. 2 and FIG. 3A at the same time. FIG. 3A shows the waveforms of some signals when the
电源控制器14具有一频率产生器(未显示)，可提供频率信号CLK，可以定义出开关周期TCYC。频率信号CLK的频率，也就是开关周期TCYC的倒数，大约会等于下臂信号DRVLS的开关频率fCYC。The
当操作于ACF模式时，开关频率fCYC大约为一固定频率，也可以加上抖频。此固定频率独立于补偿电压VCOMP。举例来说，当操作于ACF模式时，开关频率fCYC以200kHz为一中心频率，周期性地抖动变化于190kHz与210kHz之间，变化频率为400Hz，如此可以降低ACF模式时的EMI的问题。When operating in ACF mode, the switching frequency fCYC is about a fixed frequency, and frequency jittering can also be added. This fixed frequency is independent of the compensation voltage VCOMP . For example, when operating in ACF mode, the switching frequency fCYC takes 200 kHz as a center frequency, and periodically jitters between 190 kHz and 210 kHz, and the changing frequency is 400 Hz, which can reduce the EMI problem in ACF mode.
当ACF电源转换器10操作于ACF模式时，电源控制器14使得上臂信号DRVHS与下臂信号DRVLS大致为互补(complementary)，如同图3A中的上臂信号DRVHS与下臂信号DRVLS所表示的。所以ACF模式也可以说是一互补模式。当上臂信号DRVHS由逻辑上的“1”转变为“0”后，经过了空白时间(dead time)TDF后，下臂信号DRVLS就互补地由逻辑上的“0”转变为“1”。而当下臂信号DRVLS由逻辑上的“1”转变为“0”后，经过了空白时间(dead time)TDR后，上臂信号DRVHS就互补地由逻辑上的“0”转变为“1”。When the
空白时间TDR与TDF是很短的，他们的存在，除了避免上臂开关HSS与下臂开关LSS同时开启导通所造成的短路穿透(short through)现象，也可以使上臂开关HSS与下臂开关LSS进行零电压切换(zero-voltage switching，ZVS)。整体来说，尽管有空白时间TDR与TDF，上臂信号DRVHS与下臂信号DRVLS还是可以视为互补。举例来说，当下臂信号DRVLS由逻辑上的“1”转变为“0”后，绕组电压VAUX会开始从负电压VN开始快速上升，往正电压VP逼近，而开关电压VSW从0V开始快速上升，往电压VCP接近，如同图3A所示。电压VCP为上臂开关HSS与电容CAC的连接点上的电压。电源控制器14通过回馈电压VFB来检测绕组电压VAUX。一旦发现绕组电压VAUX快要抵达正电压VP了，那意味了开关电压VSW也差不多要等于电压VCP了，所以电源控制器14使上臂信号DRVHS逻辑上的“0”转变为“1”，使上臂开关HSS进行ZVS。类似的，当上臂信号DRVHS由逻辑上的“1”转变为“0”后，电源控制器14可以检测绕组电压VAUX，来辨识开关电压VSW是否大约掉到0V了，并在开关电压VSW大约为0V时，使下臂信号DRVLS由逻辑上的“0”转变为“1”，使下臂开关LSS进行ZVS。The blank times TDR and TDF are very short. Their existence not only avoids the short-through phenomenon caused by the simultaneous turn-on of the upper-arm switch HSS and the lower-arm switch LSS, but also enables the upper-arm switch HSS to be connected to the lower-arm switch LSS. The arm switch LSS performs zero-voltage switching (ZVS). Overall, the upper arm signal DRVHS and the lower arm signal DRVLS can be regarded as complementary despite the blank timeTDR andTDF . For example, after the lower arm signal DRVLS changes from logical "1" to "0", the winding voltage VAUX will start to rise rapidly from the negative voltage VN , approaching the positive voltage VP , and the switching voltage VSW It rises rapidly from 0V and approaches the voltage VCP , as shown in Figure 3A. The voltage VCP is the voltage at the connection point of the upper arm switch HSS and the capacitor CAC. The
下臂开启时间TON-L为下臂信号DRVLS为逻辑上的“1”时的时段，也就是下臂开关LSS为导通的时段；相对的，上臂开启时间TON-H为上臂信号DRVHS为逻辑上的“1”时的时段，也就是上臂开关HSS为导通的时段。The lower arm turn-on time TON-L is the time period when the lower arm signal DRVLS is logically "1", that is, the time period when the lower arm switch LSS is turned on; relatively, the upper arm turn-on time TON-H is the upper arm signal The period when DRVHS is logically "1", that is, the period when the upper arm switch HSS is turned on.
图3A也显示了电源控制器14如何调变峰值VCS-PEAK。在图3A中，缩减补偿电压VCOMP-SC大约线性的关联于补偿电压VCOMP。举例来说，VCOMP-SC＝K*VCOMP，其中K为介于0与1之间的常数。电路上可以以一分压电阻电路切割补偿电压VCOMP，而产生缩减补偿电压VCOMP-SC。缩减补偿电压VCOMP-SC可以用来控制峰值VCS-PEAK。举例来说，在下臂开启时间TON-L时，电流检测电压VCS随着时间而上升。当电源控制器14发现电流检测电压VCS超过缩减补偿电压VCOMP-SC时，电源控制器14就结束下臂开启时间TON-L，并经过空白时间TDR后，开始上臂开启时间TON-H。因此，在空白时间TDR内，电流检测电压VCS变成0V，产生了峰值VCS-PEAK，其大约等于缩减补偿电压VCOMP-SC，如同图3A所示。因此，电源控制器14依据补偿电压VCOMP，来调变峰值VCS-PEAK。相较于图3A的左边的开关周期，图3A的右边的开关周期中，缩减补偿电压VCOMP-SC增加了，所以峰值VCS-PEAK也增加了。换言之，电源控制器14使峰值VCS-PEAK大约线性地关联于补偿电压VCOMP。FIG. 3A also shows how the
在图3A中，一个开关周期TCYC，依序由空白时间TDF、下臂开启时间TON-L、空白时间TDR、与上臂开启时间TON-H所构成。频率信号CLK的一脉冲结束了上臂开启时间TON-H，开始空白时间TDF。开关电压VSW大约为0V时，空白时间TDF结束，下臂开启时间TON-L开始。当电流检测电压VCS超过缩减补偿电压VCOMP-SC时，下臂开启时间TON-L结束，空白时间TDR开始。当开关电压VSW大约为电压VCP时，空白时间TDR结束，上臂开启时间TON-H开始。频率信号CLK的下一脉冲结束了上臂开启时间TON-H，也结束了一个开关周期TCYC。In FIG. 3A , one switching period TCYC is sequentially composed of blank time TDF , lower arm turn-on time TON-L , blank time TDR , and upper arm turn-on time TON-H . One pulse of the frequency signal CLK ends the upper arm turn-on time TON-H and starts the blank time TF . When the switching voltage VSW is about 0V, the blank time TDF ends, and the lower arm turn-on time TON-L starts. When the current detection voltage VCS exceeds the reduction compensation voltage VCOMP-SC , the lower arm turn-on time TON-L ends, and the blank time TR starts. When the switching voltage VSW is about the voltage VCP , the blank time TR ends and the upper arm turn-on time TON-H begins. The next pulse of the frequency signal CLK ends the upper arm turn-on time TON-H and also ends one switching period TCYC .
ACF模式也是一种连续导通模式(continuous conduction mode，CCM)，因为流经主绕组LP的绕组电流IM一直在变化，不会停止在0A。The ACF mode is also a continuous conduction mode (CCM), because the winding currentIM flowing through the main winding LP is constantly changing and will not stop at 0A.
请同时参考图2与图3B。图3B为ACF电源转换器10操作于反激模式时的一些信号的波形。由上到下，图3B中的波形分别是频率信号CLK、上臂信号DRVHS、下臂信号DRVLS、电流检测电压VCS、开关电压VSW、以及绕组电压VAUX。Please refer to FIG. 2 and FIG. 3B at the same time. FIG. 3B shows the waveforms of some signals when the
如同图3B所示，当操作于反激模式时，如同字面上所代表的，上臂信号DRVHS大致维持于逻辑上的“0”，使上臂开关HSS为关闭状态，只有以下臂信号DRVLS来切换下臂开关LSS。反激模式是一非互补模式，因为上臂信号DRVHS与下臂信号DRVLS并不相互补。As shown in FIG. 3B, when operating in the flyback mode, as literally represented, the upper arm signal DRVHS is approximately maintained at a logical "0", so that the upper arm switch HSS is turned off, and only the lower arm signal DRVLS comes from Toggle the lower arm switch LSS. The flyback mode is a non-complementary mode because the upper arm signal DRVHS and the lower arm signal DRVLS are not complementary to each other.
在图3B中，频率信号CLK的一脉冲开始一开关周期TCYC，也开始了下臂开启时间TON-L。当电流检测电压VCS超过固定的参考电压VCS-REF时，下臂开启时间TON-L结束，解磁时间TDMG开始。参考电压VCS-REF独立于补偿电压VCOMP。在解磁时间TDMG中，二次侧绕组LS释放能量，用以建立输出电压VOUT。当二次侧绕组LS释放能量完毕，解磁时间TDMG结束，振荡时间TOSC开始，开关电压VSW开始振荡，如同图3B所示。之后，频率信号CLK的下一个脉冲结束了振荡时间TOSC，也结束了一开关周期TCYC。如同图3B所示，当操作于反激模式时，一开关周期TCYC是由下臂开启时间TON-L、解磁时间TDMG、与振荡时间TOSC所构成。In FIG. 3B, a pulse of the frequency signal CLK starts a switching period TCYC and also starts the lower arm turn-on time TON-L . When the current detection voltage VCS exceeds the fixed reference voltage VCS-REF , the lower arm turn-on time TON-L ends and the demagnetization time TDMG starts. The reference voltage VCS-REF is independent of the compensation voltage VCOMP . During the demagnetization time TDMG , the secondary winding LS releases energy to establish the output voltage VOUT . When the secondary side winding LS releases energy, the demagnetization time TDMG ends, the oscillation time TOSC begins, and the switching voltage VSW begins to oscillate, as shown in FIG. 3B . After that, the next pulse of the frequency signal CLK ends the oscillation time TOSC and also ends a switching period TCYC . As shown in FIG. 3B , when operating in the flyback mode, one switching period TCYC is composed of the lower arm turn-on time TON-L , the demagnetization time TDMG , and the oscillation time TOSC .
如同图3B所示，当操作于反激模式时，峰值VCS-PEAK并不随着缩减补偿电压VCOMP-SC或是补偿电压VCOMP变化而改变，大约维持等于固定的参考电压VCS-REF。因此，峰值VCS-PEAK独立于补偿电压VCOMP。As shown in FIG. 3B, when operating in flyback mode, the peak value VCS-PEAK does not change with the reduction of the compensation voltage VCOMP-SC or the change of the compensation voltage VCOMP , and is approximately maintained at the fixed reference voltage VCS-REF . Therefore, the peak value VCS-PEAK is independent of the compensation voltage VCOMP .
当操作于反激模式时，产生频率信号CLK的频率产生器受到补偿电压VCOMP所控制。相较于图3B的左边的开关周期，在图3B的右边之开关周期中，缩减补偿电压VCOMP-SC减少了，造成了开关周期TCYC的长度增加。When operating in the flyback mode, the frequency generator that generates the clock signal CLK is controlled by the compensation voltage VCOMP . Compared with the switching period on the left side of FIG. 3B , in the switching period on the right side of FIG. 3B , the reduction compensation voltage VCOMP-SC is reduced, resulting in an increase in the length of the switching period TCYC .
当操作于反激模式时，也可以加上抖频，用以降低EMI的问题。举例来说，当操作于反激模式时，开关频率fCYC以一平均频率为中心，周期性地变化于上限频率与下限频率之间，而平均频率是补偿电压VCOMP的函数。When operating in flyback mode, frequency jittering can also be added to reduce EMI problems. For example, when operating in the flyback mode, the switching frequency fCYC periodically varies between the upper and lower frequencies centered on an average frequency, and the average frequency is a function of the compensation voltage VCOMP .
尽管图3B显示上臂开关HSS固定为关闭状态，但本发明并不限于此。在另一个实施例中，当操作于反激模式时，上臂开关HSS并没有在下臂开启时间TON-L与解磁时间TDMG为导通状态，但是可以在振荡时间TOSC内短暂的开启，用以释放主绕组LP的漏感存放在电容CAC上的电能。Although FIG. 3B shows that the upper arm switch HSS is fixed in an off state, the present invention is not limited thereto. In another embodiment, when operating in the flyback mode, the upper arm switch HSS is not turned on during the lower arm turn-on time TON-L and the demagnetization time TDMG , but can be turned on briefly during the oscillation time TOSC , to release the electrical energy stored in the capacitor CAC by the leakage inductance of the main winding LP.
反激模式也是一种非连续导通模式(discontinuous conduction mode，DCM)，因为流经主绕组LP的绕组电流IM有一段时间会停止在0A。The flyback mode is also a discontinuous conduction mode (DCM) because the winding currentIM flowing through the main winding LP stops at 0A for a period of time.
当操作于反激模式时，如果电源控制器14发现补偿电压VCOMP大于参考电压VCOMP-REF，电源控制器14就可以脱离反激模式，进入ACF模式。When operating in the flyback mode, if the
图4放大说明图3A中，下臂开启时间TON-L内的电流检测电压VCS的波形。当操作于ACF模式时，在下臂开启时间TON-L一开始时，主绕组LP的绕组电流IM可能是负的，因此导致电流检测电压VCS一开始为负。在下臂开启时间TON-L内，因为输入电压VIN对主绕组的增磁，电流检测电压VCS-随着时间线性的增加，直到电流检测电压VCS-超过缩减补偿电压VCOMP-SC时。如同图4所示，当电流检测电压VCS为负的时段，称为负电流时间TON-N；当电流检测电压VCS为正的时段，称为正电流时间TON-P。唯有正电流时间TON-P大于负电流时间TON-N时，ACF电源转换器10才能对输出电压VOUT提供电能。换言之，当正电流时间TON-P非常接近负电流时间TON-N时，代表负载13可能不是处于重载状态，可能是处于中载状态或是轻载状态。FIG. 4 is an enlarged illustration of the waveform of the current detection voltage VCS in the lower arm turn-on time TON-L in FIG. 3A . When operating in ACF mode, the winding currentIM of the main winding LP may be negative at the beginning of the lower arm turn-on time TON-L , thus causing the current sense voltage VCS to be initially negative. During the turn-on time TON-L of the lower arm, the current detection voltage VCS- increases linearly with time due to the magnetization of the main winding by the input voltage VIN , until the current detection voltage VCS- exceeds the reduction compensation voltage VCOMP-SC Time. As shown in FIG. 4 , the period when the current detection voltage VCS is negative is called the negative current time TON-N ; the period when the current detection voltage VCS is positive is called the positive current time TON-P . The
图4也可以发现，操作于ACF模式时，补偿电压VCOMP或是缩减补偿电压VCOMP-SC并不能代表负载13的状态，因为有负电流时间TON-N的存在。所以，相较于依据补偿电压VCOMP来脱离ACF模式，依据正电流时间TON-P与负电流时间TON-N来决定是否脱离ACF模式，将是比较好的选择。It can also be found in FIG. 4 that when operating in the ACF mode, the compensation voltage VCOMP or the reduced compensation voltage VCOMP-SC cannot represent the state of the
如同图2所举例的，在一实施例中，电源控制器14检查正电流时间TON-P与负电流时间TON-N是否符合一预定关系，来判断是否脱离ACF模式，进入反激模式。举例来说，当TON-P<TON-N+KT时，电源控制器14可以脱离ACF模式，进入反激模式，其中，KT为一固定值。这预定关系并非限定于比较正电流时间TON-P与负电流时间TON-N的大小，在另一个实施例中，电源控制器14检查吸能工作周期DON-P是否小于一定值，其中吸能工作周期DON-P定义为TON-P/(TON-P+TON-N)。当吸能工作周期DON-P小于该定值时，电源控制器14可以脱离ACF模式，进入反激模式。As exemplified in FIG. 2 , in one embodiment, the
在一实施例中，当正电流时间TON-P与负电流时间TON-N符合该预定关系时，电源控制器14就马上脱离ACF模式，进入反激模式，但本发明并不限于此。在另一个实施例中，当该预定关系持续符合一段预定时间，例如1ms，电源控制器14才脱离ACF模式，进入反激模式。这样延迟一段预定时间才脱离ACF模式的方法，可以在负载瞬时反应(load transientresponse)测试时得到好处。假定这一段预定时间是1ms，且在负载瞬时反应测试下，负载13的轻重载切换周期小于1ms，意味着负载13脱离重载状态不超过1ms，就会回到重载状态。那在如此负载瞬时反应测试下，电源控制器14就会一直操作于ACF模式，不会进入反激模式。如此ACF电源转换器10享有比较快的反应速度以及比较稳定的输出电压。In one embodiment, when the positive current time TON-P and the negative current time TON-N conform to the predetermined relationship, the
图5A显示图1的实施例中，开关频率fCYC与补偿电压VCOMP的关系。当操作于ACF模式时，开关频率fCYC与补偿电压VCOMP的关系，以线条CfCYC-ACF表示；当操作于反激模式时，则以线条CfCYC-FLY表示。线条CfCYC-ACF显示，操作于ACF模式时，开关频率fCYC为固定值fH，独立于补偿电压VCOM。线条CfCYC-FLY，在补偿电压VCOMP介于4.3V与0.7V之间，则显示了操作于反激模式时，开关频率fCYC与补偿电压VCOMP有一正向的线性关系；开关频率fCYC随着补偿电压VCOMP增加而线性地增加。当图1的实施例具有抖频的功能时，线条CfCYC-ACF与CfCYC-FLY，就分别表示操作于ACF模式与反激模式时，开关频率fCYC抖频时的平均频率。FIG. 5A shows the relationship between the switching frequency fCYC and the compensation voltage VCOMP in the embodiment of FIG. 1 . When operating in the ACF mode, the relationship between the switching frequency fCYC and the compensation voltage VCOMP is represented by the line CfCYC-ACF ; when operating in the flyback mode, it is represented by the line CfCYC-FLY . The line CfCYC-ACF shows that when operating in ACF mode, the switching frequency fCYC is a fixed value fH , independent of the compensation voltage VCOM . The line CfCYC-FLY , when the compensation voltage VCOMP is between 4.3V and 0.7V, shows that when operating in the flyback mode, the switching frequency fCYC and the compensation voltage VCOMP have a positive linear relationship; the switching frequency fCYC increases linearly as the compensation voltageVCOMP increases. When the embodiment of FIG. 1 has the function of frequency jittering, the lines CfCYC-ACF and CfCYC-FLY respectively represent the average frequency of the switching frequency fCYC when operating in the ACF mode and the flyback mode.
图5A也显示了，当补偿电压VCOMP低于0.5V时，电源控制器14可以操作于丛发模式(burst mode)，不论先前是操作于ACF模式或是反激模式。丛发模式可以节省开关损失，提升轻载或是无载状态时的电能转换效率。当输出电流IO很低但大于0A，使得补偿电压VCOMP低于0.5V时，电源控制器14就关闭上臂开关HSS以及下臂开关LSS，使开关频率fCYC为0，停止电能转换。但是，既然输出电流IO大于0A，没有电能转换的结果，最终将会导致补偿电压VCOMP随着时间而上升。一旦电源控制器14发现补偿电压VCOMP超过0.7V，电源控制器14就恢复为操作于ACF模式或是反激模式，开始电能转换。如果输出电流IO依然很低，ACF电源转换器10供应给负载13的电能大于负载13所消耗的电能，一段时间后，补偿电压VCOMP将会再度低于0.5V，导致电能转换停止。如此，开关频率fCYC会循环地一段时间不为0Hz，另一段时间为0Hz，这称为丛发模式。FIG. 5A also shows that when the compensation voltage VCOMP is lower than 0.5V, the
图5B显示图1的实施例中，峰值VCS-PEAK与补偿电压VCOMP的关系。当操作于ACF模式时，峰值VCS-PEAK与补偿电压VCOMP的关系，以线条CVCS-P-ACF表示；当操作于反激模式时，则以线条CVCS-P-FLY表示。线条CVCS-P-ACF显示，操作于ACF模式时，峰值VCS-PEAK与补偿电压VCOMP有一正向的线性关系；峰值VCS-PEAK随着补偿电压VCOMP增加而增加。线条CVCS-P-FLY则显示了，操作于反激模式时，峰值VCS-PEAK为固定的参考电压VCS-REF，独立于补偿电压VCOMP。FIG. 5B shows the relationship between the peak value VCS-PEAK and the compensation voltage VCOMP in the embodiment of FIG. 1 . When operating in the ACF mode, the relationship between the peak value VCS-PEAK and the compensation voltage VCOMP is represented by the line CVCS-P-ACF ; when operating in the flyback mode, it is represented by the line CVCS-P-FLY . The line CVCS-P-ACF shows that when operating in ACF mode, the peak VCS-PEAK has a positive linear relationship with the compensation voltage VCOMP ; the peak VCS-PEAK increases as the compensation voltage VCOMP increases. The line CVCS-P-FLY shows that when operating in flyback mode, the peak VCS-PEAK is a fixed reference voltage VCS-REF , independent of the compensation voltage VCOMP .
图5C显示图1的实施例中，处于稳态时，输出电流IO与补偿电压VCOMP的关系，以及ACF模式与反激模式之间的切换。当操作于ACF模式时，输出电流IO与补偿电压VCOMP的关系以线条CIO-ACF表示；当操作于反激模式时，则以线条CIO-FLY表示。假定ACF电源转换器10的起始状态是输出电流IO小于参考电流IO-2，依据图5C，电源控制器14一开始是操作于反激模式。输出电流IO的改变，将会使补偿电压VCOMP依据线条CIO-FLY而跟着变化。之后，假定输出电流IO逐渐地增加。当输出电流IO超过参考电流IO-1时，电源控制器14发现了补偿电压VCOMP大于参考电压VCOMP-REF。因此，电源控制器14脱离反激模式，进入ACF模式，补偿电压VCOMP会跳跃式的增高，如同图5C所示。之后，输出电流IO的改变，将会使补偿电压VCOMP依据线条CIO-ACF而跟着变化。之后，当输出电流IO缩小到参考电流IO-2时，电源控制器14发现了正电流时间TON-P与负电流时间TON-N已经符合脱离ACF模式的预定关系，因此脱离ACF模式，进入反激模式，补偿电压VCOMP会跳跃式的减少。FIG. 5C shows the relationship between the output currentIO and the compensation voltage VCOMP and the switching between the ACF mode and the flyback mode in the steady state in the embodiment of FIG. 1 . When operating in the ACF mode, the relationship between the output currentIO and the compensation voltage VCOMP is represented by the line CIO-ACF ; when operating in the flyback mode, it is represented by the line CIO-FLY . Assuming that the initial state of the
图6显示图1的ACF电源转换器10操作于反激模式时的数个连续的开关周期TCYC。在图6中，ACF电源转换器10以下臂信号DRVLS开关下臂开关LSS，持续地、周期性地产生连续N个开关周期TCYC，其中N为大于1的整数，举例来说，N可以是8。FIG. 6 shows several consecutive switching cycles TCYC when the
由上而下，图6中的波形分别是频率信号CLK、上臂信号DRVHS、下臂信号DRVLS、电流检测电压VCS、开关电压VSW、遮蔽信号SBLAN、以及计数CNT。From top to bottom, the waveforms in FIG. 6 are the frequency signal CLK, the upper arm signal DRVHS , the lower arm signal DRVLS , the current detection voltage VCS , the switching voltage VSW , the mask signal SBLAN , and the count CNT, respectively.
遮蔽信号SBLAN由电源控制器14内部所产生，用以提供一遮蔽时间TBLAN(blankingtime)。在每一个开关周期TCYC中，至少遮蔽时间TBLAN过去了之后，下一个开关周期TCYC才可以开始。因此，每个开关周期TCYC不小于遮蔽时间TBLAN。遮蔽时间TBLAN可以依据负载13而产生。在实施例中，遮蔽时间TBLAN依据补偿电压VCOMP而产生，而遮蔽频率fBLAN(＝1/TBLAN)与补偿电压VCOMP的关系，大致可以以图5A的线条CfCYC-FLY表示。The blanking signal SBLAN is internally generated by the
电源控制器14中提供有一计数器(未显示)，存有计数CNT，用来数算这些开关周期TCYC。图6显示，当计数CNT显示有N个开关周期TCYC出现后，计数CNT重置为1，重新计数。A counter (not shown) is provided in the
图7为电源控制器14中所使用的控制方法60。当计数CNT为1到N-1时，表示当下为第1到第N-1个开关周期TCYC，步骤62中检查计数CNT是否为N的检查结果将为否定，控制方法60将从步骤62前进到正常反激周期中的步骤。因此，第1到第N-1个开关周期TCYC视为正常反激周期。当计数CNT为N时，表示当下为第N个开关周期TCYC，步骤62中的检查结果将为肯定，控制方法60从步骤62前进到改良式反激周期中的步骤。因此，第N个开关周期TCYC视为改良式反激周期。FIG. 7 is a
换言之，图6中的计数CNT为1到N-1时的开关周期TCYC，都是正常反激周期；计数CNT为N的开关周期TCYC，则是改良式反激周期。在改良式反激周期结束时，计数CNT重置成1，重新计数。图6显示了每N个连续开关周期TCYC中，有一改良式反激周期，其他都是正常反激周期。但本发明并不限于此。在另一个实施例中，每N个连续开关周期TCYC中，有数个连续的改良式反激周期，其他都是正常反激周期。In other words, the switching period TCYC when the count CNT is 1 to N-1 in FIG. 6 is a normal flyback period; the switching period TCYC when the count CNT is N is an improved flyback period. At the end of the modified flyback cycle, the count CNT is reset to 1 and counted again. Figure 6 shows that in every N continuous switching cycles TCYC , there is an improved flyback cycle, and the others are normal flyback cycles. However, the present invention is not limited to this. In another embodiment, in every N consecutive switching cycles TCYC , there are several consecutive improved flyback cycles, and the others are normal flyback cycles.
以图6为例，正常反激周期跟改良式反激周期之间的主要差异其中之一，在于上臂信号DRVHS的波形。在一正常反激周期内，上臂信号DRVHS固定是逻辑上的“0”，保持上臂开关HSS为关闭状态。不一样的，改良式反激周期中，上臂信号DRVHS大部分时间是在逻辑上的“0”，但只有在改良式反激周期快要结束时，才短暂的为逻辑上的“1”，短暂地使上臂开关HSS为开启的导通状态。因此，在改良式反激周期中，开关周期TCYC包含有上臂开启时间TON-H，如同图6的第N个开关周期TCYC所显示。Taking Fig. 6 as an example, one of the main differences between the normal flyback period and the improved flyback period is the waveform of the upper arm signal DRVHS . In a normal flyback period, the upper arm signal DRVHS is fixed at logical "0", keeping the upper arm switch HSS in an off state. Differently, in the improved flyback cycle, the upper arm signal DRVHS is a logical "0" most of the time, but only when the improved flyback period is about to end, it is a logical "1" for a short time. The upper arm switch HSS is briefly turned on to conduct. Therefore, in the modified flyback period, the switching period TCYC includes the upper arm turn-on time TON-H , as shown in the Nth switching period TCYC of FIG. 6 .
每个正常反激周期中，因为上臂开关HSS一直保持在关闭状态，因此，主绕组的漏感在每个下臂开启时间TON-L所产生的磁能，将转换成电能，累积于电容CAC上，使得电压VCP增加。每个改良式反激周期中，因为上臂开关HSS短暂的开启，所以电容CAC上累积的电能，可以释放转换给输出电压VOUT，提高转换效率。而且，也可以降低电压VCP以及下臂开关LSS于关闭时必须承受的电压，避免下臂开关LSS因为过高的电压应力而损伤。In each normal flyback cycle, because the upper arm switch HSS is kept in the off state, the magnetic energy generated by the leakage inductance of the main winding at each lower arm turn-on time TON-L will be converted into electric energy and accumulated in the capacitor CAC , so that the voltage VCP increases. In each improved flyback cycle, because the upper arm switch HSS is briefly turned on, the electric energy accumulated on the capacitor CAC can be released and converted to the output voltage VOUT , thereby improving the conversion efficiency. In addition, the voltage VCP and the voltage that the lower arm switch LSS must withstand when turned off can also be reduced, so as to avoid damage to the lower arm switch LSS due to excessive voltage stress.
因此，本发明的实施例中，有源钳位电路并不需要一个泄流电阻，不但可以增加转换效率，也可以降低生产成本。如同图1的ACF电源转换器10所举例的，有源钳位电路ACC在上臂开关HSS关闭时，不会有能量的损耗，为一能量无损耗有源钳位电路。Therefore, in the embodiment of the present invention, the active clamp circuit does not need a bleeder resistor, which can not only increase the conversion efficiency, but also reduce the production cost. As exemplified by the
请参阅图7，在每一正常反激周期中，步骤64a以下臂信号DRVLS开启下臂开关LSS，产生下臂开启时间TON-L，并大约固定峰值VCS-PEAK。以图6中的第1个开关周期TCYC为例。第1个开关周期TCYC中大部分的信号波形可以参考图3B以及相关解释而得知，不再重述。如同图6中的第1个开关周期TCYC所举例，遮蔽时间TBLAN与下臂开启时间TON-L一起开始。遮蔽时间TBLAN的长短可以随着负载13而调整。在第1个开关周期TCYC中，遮蔽时间TBLAN涵盖了下臂开启时间TON-L、解磁时间TDMG、以及部份的振荡时间TOSC。第1个开关周期TCYC的振荡时间TOSC内，开关电压VSW振荡，产生波峰PK1、PK2以及波谷VY1、VY2、VY3。Referring to FIG. 7 , in each normal flyback cycle,
图7的步骤66a等待遮蔽时间TBLAN过去。图6中的第1个开关周期TCYC中，遮蔽时间TBLAN大约结束于波峰PK2出现后。
图7的步骤68接续步骤66a，检测并等待一波谷出现。当步骤68发现到波谷出现时，此正常反激周期结束，并在步骤70中使计数CNT增加1。图6的第1个开关周期TCYC中，在时间点tDET检测到波谷VY3的出现，因此，频率信号CLK结束第1个开关周期TCYC。在时间点tDET，计数CNT增加1，第2个开关周期TCYC开始。
请参阅图7，在每一改良式反激周期中的步骤64b与66b，相同于正常反激周期中的步骤64a与66a，可以参阅先前的说明得知，不再重述。图6中的第N个开关周期TCYC为一改良式反激周期，其中，遮蔽时间TBLAN涵盖了下臂开启时间TON-L、解磁时间TDMG、以及部份的振荡时间TOSC。第N个开关周期TCYC的振荡时间TOSC内，开关电压VSW振荡，产生波峰PK1、PK2、PK3以及波谷VY1、VY2、VY3。Referring to FIG. 7 , steps 64b and 66b in each improved flyback cycle are the same as
图7的步骤72接续步骤66b，检测并等待一波峰出现。当步骤72发现到波峰出现时，步骤74接着开启上臂开关HSS，开始上臂开启时间TON-H。如同图6的第N个开关周期TCYC中所示，波峰PK3为遮蔽时间TBLAN之后第一个出现的波峰。所以上臂开启时间TON-H开始于波峰PK3出现时。在上臂开启时间TON-H内，上臂开关HSS与电容CAC的连接点上的电压VCP将会因放电而下降。
在一些实施例中，每一改良式反激周期内的上臂开启时间TON-H都只有出现一次，而且是在遮蔽时间TBLAN结束之后，如同图6所举例。In some embodiments, the upper arm turn-on time TON-H occurs only once in each modified flyback period, and it is after the shadowing time TBLAN ends, as exemplified in FIG. 6 .
在一些实施例中，每一改良式反激周期内的上臂开启时间TON-H都一样，为一预设的时间长度，但本发明不限于此。在一些实施例中，上臂开启时间TON-H的长度系由上臂开关HSS与电容CAC的连接点上的电压VCP所决定，而电压VCP可由绕组电压VAUX所感应，而绕组电压VAUX可通过回馈接脚FB被电源控制器14所检测。举例来说，在上臂开启时间TON-H内，电源控制器14通过回馈接脚FB，检测电压VCP。当电源控制器14发现电压VCP已经低于一参考值时，电源控制器14才结束一改良式反激周期内的上臂开启时间TON-H。In some embodiments, the upper arm turn-on time TON-H in each improved flyback period is the same, which is a predetermined time length, but the invention is not limited thereto. In some embodiments, the length of the upper arm turn-on time TON-H is determined by the voltage VCP at the connection point of the upper arm switch HSS and the capacitor CAC, and the voltage VCP can be induced by the winding voltage VAUX , and the winding voltage VAUX can be detected by the
接续步骤74的上臂开启时间TON-H结束后，图7的步骤76等待空白时间TDF，并在开关电压VSW大约为0V时，使下臂信号DRVLS由逻辑上的“0”转变为“1”，也就是使下臂开关LSS进行ZVS。步骤78结束第N个开关周期TCYC，重置计数CNT，使其为1，让下一个开关周期TCYC开始。After the upper arm turn-on time TON-H in
从图6以及图7中的例子可知，在ACF电源转换器10操作于反激模式时，ACF电源转换器10可以视为一准谐振电源转准换器(quasi-resonant power converter)，因为正常反激周期与改良式反激周期都大约结束于一波谷出现时，显示出波谷切换。波谷切换可以降低切换损失，但本发明并不限于此。在ACF电源转换器10操作于反激模式时，ACF电源转换器10不一定要进行波谷切换。举例来说，在一些实施例中，图7中的步骤68可以省略。也就是在一些实施例中的正常反激周期中，大约当遮蔽时间TBLAN一结束，就开始下一个开关周期。It can be known from the examples in FIG. 6 and FIG. 7 that when the
尽管图6以及图7举例了一改良式反激周期中，上臂开启时间TON-H开始于波峰PK3出现时，但本发明并不限于此。在其他实施例中，图7中的步骤72可以修改或是省略。在一些实施例中，图7中的步骤72修改为检测等待一波谷的出现，也就是上臂开启时间TON-H开始于遮蔽时间TBLAN结束后的第一个波谷。在一些实施例中，图7中的步骤72省略，也就是上臂开启时间TON-H紧接于遮蔽时间TBLAN结束后。Although FIGS. 6 and 7 illustrate an improved flyback cycle, the upper arm turn-on time TON-H starts when the peak PK3 appears, but the present invention is not limited thereto. In other embodiments,
尽管先前的解说中，N为一固定整数，但本发明并不限于此。在一实施例中，N可以适应性地被改变。举例来说，在图6中的第N个开关周期的上臂开启时间TON-H快要结束，电源控制器14可以通过回馈接脚FB来检测辅助绕组LA的绕组电压VAUX，等同检测上臂开关HSS与电容CAC的连接点上的电压VCP。当电压VCP大于一默认合理范围时，表示N可能太多个了，因此在第N个开关周期结束时，将N减少1，相对地增加改良式反激周期出现的频率；相对的，当电压VCP小于那默认合理范围时，N可以增加1，等同增加正常开关周期出现的频率。Although in the previous explanation, N is a fixed integer, the present invention is not limited to this. In an embodiment, N can be adaptively changed. For example, when the upper arm turn-on time TON-H of the Nth switching cycle in FIG. 6 is about to end, the
以上所述仅为本发明之较佳实施例，凡依本发明申请专利范围所做之均等变化与修饰，皆应属本发明之涵盖范围。The above descriptions are only preferred embodiments of the present invention, and all equivalent changes and modifications made according to the scope of the patent application of the present invention shall fall within the scope of the present invention.
附图标记列表List of reference signs
10 ACF电源转换器10 ACF Power Converters
13 负载13 load
14 电源控制器14 Power Controller
60 控制方法60 Control methods
62、62、64a、64b、66a、66b、70、72、74、76、78 步骤62, 62, 64a, 64b, 66a, 66b, 70, 72, 74, 76, 78 steps
ACC 有源钳位电路ACC Active Clamp
BD 桥式整流器BD Bridge Rectifier
CAC 电容CAC capacitor
CCOMP 补偿电容CCOMP compensation capacitor
CIO-ACF、CIO-FLY、CVCS-P-ACF、CVCS-P-FLY、CfCYC-ACF、CfCYC-FLY 线条CIO-ACF , CIO-FLY , CVCS-P-ACF , CVCS-P-FLY , CfCYC-ACF , CfCYC-FLY lines
CLK 频率信号CLK frequency signal
CNT 计数CNT count
CS 电流检测接脚CS current detection pin
DVR 驱动器DVR drive
DRVHS 上臂信号DRVHS upper arm signal
DRVLS 下臂信号DRVLS lower arm signal
EA 误差放大器EA Error Amplifier
fCYC 开关频率fCYC switching frequency
fH 固定值fH fixed value
FB 回馈接脚FB feedback pin
GNDI 输入地电源线GNDI input ground power line
GNDO 输出地电源线GNDO output ground power line
HD、LD 接脚HD, LD pins
HSS 上臂开关HSS upper arm switch
IM 绕组电流IM winding current
IN 输入电源线IN input power cord
IO 输出电流IO output current
IO-1、IO-2 参考电流IO-1 ,IO-2 reference current
LA 辅助绕组LA auxiliary winding
LP 主绕组LP main winding
LS 二次侧绕组LS secondary winding
LSS 下臂开关LSS lower arm switch
OPT 光耦合器OPT optocoupler
OUT 输出电源线OUT output power cord
PK1、PK2、PK3 波峰PK1 , PK2 , PK3 peaks
RA、RB 电阻RA, RB resistance
RCS 电流检测电阻RCS current sense resistor
SBLAN 遮蔽信号SBLAN masking signal
tDET 时间点tDET time point
TBLAN 遮蔽时间TBLAN shade time
TCYC 开关周期TCYC switching cycle
TDF、TDR 空白时间TDF ,TDR blank time
TF 变压器TF transformer
TDMG 解磁时间TDMG demagnetization time
TON-H 上臂开启时间TON-H upper arm opening time
TON-L 下臂开启时间TON-L lower arm opening time
TON-N 负电流时间TON-N Negative Current Time
TON-P 正电流时间TON-P positive current time
TOSC 振荡时间TOSC oscillation time
VAC 交流市电VAC AC mains
VAUX 绕组电压VAUX winding voltage
VCC 操作电源VCC operating power supply
VCOMP 补偿电压VCOMP compensation voltage
VCOMP-REF 参考电压VCOMP-REF reference voltage
VCOMP-SC 缩减补偿电压VCOMP-SC Reduced Compensation Voltage
VCP 电压VCP voltage
VCS 电流检测电压VCS current sense voltage
VCS-PEAK 峰值VCS-PEAK peak
VCS-REF 参考电压VCS-REF reference voltage
VCC 电源接脚VCC power pin
VFB 回馈电压VFB feedback voltage
VIN 输入电压VIN input voltage
VOUT 输出电压VOUT output voltage
VP 正电压VP positive voltage
VREF-TAR 目标电压VREF-TAR target voltage
VSW 开关电压VSW switch voltage
VY1、VY2、VY3 波谷VY1 , VY2 , VY3 trough
Claims (10)
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| CN201811180757.8ACN111030479B (en) | 2018-10-09 | 2018-10-09 | Active-clamp flyback power converter and associated control method |
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| CN112003476B (en)* | 2020-08-06 | 2022-02-15 | 东南大学 | A control method for reducing the conduction time of the body diode of an ACF power tube |
| CN112701927A (en)* | 2021-01-25 | 2021-04-23 | 东莞市石龙富华电子有限公司 | Programmable multi-mode flyback automatic frequency increasing method for switching power supply |
| CN113708640B (en)* | 2021-08-25 | 2023-01-20 | 深圳中科乐普医疗技术有限公司 | Active clamping flyback converter, control method thereof and switching power supply system |
Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102122890A (en)* | 2010-10-11 | 2011-07-13 | 南京航空航天大学 | Control method for auxiliary switching tube of active clamp flyback converter |
| CN103795260A (en)* | 2014-01-21 | 2014-05-14 | 广州金升阳科技有限公司 | Non-complementary flyback active clamp converter |
| TWI514742B (en)* | 2014-07-16 | 2015-12-21 | Grenergy Opto Inc | Power controllers and related control methods |
| CN107181410A (en)* | 2016-03-12 | 2017-09-19 | 快捷韩国半导体有限公司 | Active clamp flyback converter |
| CN107453610A (en)* | 2017-07-31 | 2017-12-08 | 西安矽力杰半导体技术有限公司 | Inverse excitation type converter and its active-clamp control circuit and active-clamp control method |
Family Cites Families (3)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US10038387B2 (en)* | 2013-04-15 | 2018-07-31 | Semiconductor Components Industries, Llc | Control circuit for active clamp flyback power converter with predicted timing control |
| US9991800B2 (en)* | 2015-05-20 | 2018-06-05 | Semiconductor Components Industries, Llc | Switched mode power supply with efficient operation at light loads and method therefor |
| US10148181B2 (en)* | 2016-10-07 | 2018-12-04 | Semiconductor Components Industries, Llc | Switched mode power supply with dynamic frequency foldback |
- 2018
- 2018-10-09CNCN201811180757.8Apatent/CN111030479B/enactiveActive
Patent Citations (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN102122890A (en)* | 2010-10-11 | 2011-07-13 | 南京航空航天大学 | Control method for auxiliary switching tube of active clamp flyback converter |
| CN103795260A (en)* | 2014-01-21 | 2014-05-14 | 广州金升阳科技有限公司 | Non-complementary flyback active clamp converter |
| TWI514742B (en)* | 2014-07-16 | 2015-12-21 | Grenergy Opto Inc | Power controllers and related control methods |
| CN107181410A (en)* | 2016-03-12 | 2017-09-19 | 快捷韩国半导体有限公司 | Active clamp flyback converter |
| CN107453610A (en)* | 2017-07-31 | 2017-12-08 | 西安矽力杰半导体技术有限公司 | Inverse excitation type converter and its active-clamp control circuit and active-clamp control method |
Non-Patent Citations (1)
| Title |
|---|
| "选路控制的光伏并网微型逆变器研究";李晓义，等;《燕山大学学报》;20170131;第41卷(第1期);全文* |
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