技术领域technical field
本发明涉及DC/DC电源变换器，尤其涉及一种具有双谐振频率的LLC谐振电源变换器。The invention relates to a DC/DC power converter, in particular to an LLC resonant power converter with double resonance frequencies.
背景技术Background technique
目前，对于应用在商用直流电源和工业电源单元的现代开关电源而言，都有高效率、高功率密度、小体积以及高可靠性的要求。电源通常由两级结构组成，第一级是升压功率因数校正级，第二级是高频链DC/DC转换级。全桥结构由于其低电压应力和较小的变压器尺寸，得以广泛使用在第二级电路中。At present, for modern switching power supplies used in commercial DC power supplies and industrial power supply units, there are requirements for high efficiency, high power density, small size, and high reliability. The power supply usually consists of a two-stage structure, the first stage is a step-up power factor correction stage, and the second stage is a high-frequency link DC/DC conversion stage. The full bridge structure is widely used in the second stage due to its low voltage stress and smaller transformer size.
由于传统全桥结构的开关器件在开关过程中损耗和振荡都比较大，现在逐渐被零电压开关(ZVS)全桥结构所取代，其中移相全桥(PSFB)结构就实现了开关管的零电压导通。然而，诸如续流阶段的高环路电流损耗、副边占空比丢失、变压器磁通密度偏差以及负载电流减小时滞后臂难于实现ZVS等等这些缺点，极大地限制了PSFB拓扑在高频领域的应用。Due to the relatively large loss and oscillation of the switching device in the traditional full-bridge structure, it is gradually replaced by the zero-voltage switching (ZVS) full-bridge structure, and the phase-shifted full-bridge (PSFB) structure realizes the zero switching of the switching tube. The voltage is turned on. However, disadvantages such as high loop current loss in the freewheeling phase, loss of secondary duty cycle, transformer flux density deviation, and difficulty in achieving ZVS in the lagging arm when the load current decreases greatly limit the PSFB topology in the high-frequency field. Applications.
LLC串联谐振DC/DC变换器实现了在宽负载范围内原边的ZVS以及副边的零电流开关(ZCS)，LLC串联谐振变换器的软开关过程使其应用在许多场合，特别是在高输出电压和低输出电流的场合。通常，LLC串联谐振变换器通过调整开关频率来调节输出电压，这是因为若其工作在感性区，降低其开关频率就会增大电压转换率。对于LLC串联谐振变换器而言，很难通过频率调制(FM)设计得到宽输入/输出范围的变换器。The LLC series resonant DC/DC converter realizes ZVS on the primary side and zero current switching (ZCS) on the secondary side in a wide load range. The soft switching process of the LLC series resonant converter makes it suitable for many occasions, especially in high output voltage and low output current applications. Usually, the LLC series resonant converter regulates the output voltage by adjusting the switching frequency, because if it works in the inductive region, reducing its switching frequency will increase the voltage conversion rate. For the LLC series resonant converter, it is difficult to design a converter with a wide input/output range through frequency modulation (FM).
目前，针对LLC串联谐振变换器，考虑降低开关频率来得到高轻载效率的方案比较多。例如通过增加辅助电路或采用不同的控制方案。然而，这些方案使用了复杂的辅助绕组或非对称的脉宽调制(APWM)技术，具有功率损耗大或磁通密度偏差等缺点。At present, for the LLC series resonant converter, there are many schemes considering reducing the switching frequency to obtain high light-load efficiency. For example by adding auxiliary circuits or adopting different control schemes. However, these schemes use complex auxiliary windings or asymmetric pulse width modulation (APWM) techniques, which have disadvantages such as large power loss or deviation of magnetic flux density.
发明内容Contents of the invention
本发明目的是针对现有技术存在的缺陷提供一种具有双谐振频率的LLC谐振电源变换器，The object of the present invention is to provide a kind of LLC resonant power converter with double resonant frequency in view of the defects existing in the prior art,
本发明为实现上述目的，采用如下技术方案：一种具有双谐振频率的LLC谐振电源变换器，LLC谐振电源变换器的谐振网络设有四个NMOS管M3、M4、M5及M6，电感Lr和Lm以及电容Cr，NMOS管M3和M4构成一个桥臂，NMOS管M5和M6构成另一个桥臂，NMOS管M3的源极连接NMOS管M4的漏极和电感Lr的一端，NMOS管M5的源极连接NMOS管M6的漏极和电感Lm的一端，电感Lm的另一端串接电容Cr后连接电感Lr的另一端，NMOS管M3和M5的漏极均连接输入电压VIN正端，NMOS管M4和M6的源极均连接输入电压VIN负端并连接输入地，NMOS管M3、M4、M5及M6的源、漏极之间均分别并联有体二极管和寄生电容；In order to achieve the above object, the present invention adopts the following technical scheme: an LLC resonant power converter with dual resonant frequencies, the resonant network of the LLC resonant power converter is provided with four NMOS transistors M3 , M4 , M5 and M6 , inductance Lr and Lm and capacitor Cr, NMOS transistorsM3 andM4 constitute a bridge arm, NMOS transistorsM5 andM6 constitute another bridge arm, the source of NMOS transistorM3 is connected to the drain of NMOS transistorM4 and One end of the inductor Lr, the source of the NMOS transistorM5 is connected to the drain of the NMOS transistorM6 and one end of the inductor Lm, the other end of the inductor Lm is connected in series with the capacitor Cr and then connected to the other end of the inductor Lr, the NMOS transistorsM3 andM5 The drains of the NMOS transistors M4 and M 6 are connected to the positive terminal of the input voltage VIN , the sources of the NMOS transistors M 4 and M6 are connected to the negative terminal of the input voltage VIN and the input ground, and the sources of the NMOS transistors M3 , M4 , M5 and M6 Body diodes and parasitic capacitances are respectively connected in parallel between the drains and the drains;
其特征在于：增设包括NMOS管M1、M2和电感La构成的有源网络，该有源网络与包括四个NMOS管M3、M4、M5及M6的LLC谐振电源变换器的谐振网络共同构成附加有源谐振的谐振网络，其中增设的NMOS管M1和M2构成第三个桥臂，NMOS管M1的源极连接NMOS管M2的漏极和电感La的一端，电感La的另一端连接NMOS管M3源极和NMOS管M4漏极与电感Lr的连接端，NMOS管M1的漏极连接输入电压VIN正端，NMOS管M2的源极连接输入电压VIN负端并连接输入地，NMOS管M1和M2的源、漏极之间均分别并联有体二极管和寄生电容；It is characterized in that an active network composed of NMOS transistors M1 , M2 and inductor La is added, and the active network is connected with an LLC resonant power converter including four NMOS transistors M3 , M4 , M5 and M6 The resonant network together constitutes a resonant network with additional active resonance, wherein the added NMOS transistorsM1 andM2 form the third bridge arm, and the source of the NMOS transistorM1 is connected to the drain of the NMOS transistorM2 and one end of the inductor La. The other end of the inductor La is connected to the source of the NMOS transistorM3 , the drain of the NMOS transistorM4 and the connection end of the inductor Lr, the drain of the NMOS transistorM1 is connected to the positive terminal of the input voltage VIN , and the source of the NMOS transistorM2 is connected to the input The negative terminal of the voltage VIN is connected to the input ground, and the sources and drains of the NMOS transistors M1 and M2 are respectively connected in parallel with body diodes and parasitic capacitances;
附加有源谐振的谐振网络的输出依次串接隔离变压器和Class D全桥整流，ClassD全桥整流通过负载依次连接输出采样电路、误差放大电路、STM32F407微控制器和高频栅驱动电路，高频栅驱动电路的输出驱动附加有源网络的谐振网络中三个桥臂共六个MOS管M1～M6的正常工作；The output of the resonant network with additional active resonance is connected in series with an isolation transformer and a Class D full-bridge rectifier. The Class D full-bridge rectifier is sequentially connected to the output sampling circuit, error amplifier circuit, STM32F407 microcontroller and high-frequency gate drive circuit through the load. The output of the gate drive circuit drives the normal operation of six MOS transistors M1 to M6 in total in three bridge arms in the resonant network of the additional active network;
隔离变压器原边的同名端和异名端与电感Lm并联连接；The same-named end and the different-named end of the primary side of the isolation transformer are connected in parallel with the inductance Lm;
Class D全桥整流包括二极管D7、D8、D9、D10和滤波电容Cf，二极管D7的阳极连接二极管D8的阴极和隔离变压器副边的同名端，二极管D9的阳极连接二极管D10的阴极和隔离变压器副边的异名端，二极管D7的阴极连接二极管D9的阴极和滤波电容Cf的正端并作为ClassD全桥整流的输出端连接负载电阻R的一端，二极管D8的阳极连接二极管D10的阳极和滤波电容Cf的负端并连接输出地；Class D full-bridge rectification includes diodes D7 , D8 , D9 , D10 and filter capacitor Cf , the anode of diode D7 is connected to the cathode of diode D8 and the terminal of the same name on the secondary side of the isolation transformer, and the anode of diode D9 is connected to The cathode of diode D10 is connected to the opposite end of the secondary side of the isolation transformer, and the cathode of diode D7 is connected to the cathode of diode D9 and the positive end of filter capacitor Cf and connected to one end of the load resistor R as the output end of Class D full-bridge rectification. The anode of the diode D8 is connected to the anode of the diode D10 and the negative terminal of the filter capacitor Cf and connected to the output ground;
输出采样电路包括电阻R1、R2和Rs，电阻R1、R2构成输出电压采样电路，电阻R1的一端连接Class D全桥整流的输出端，电阻R1的另一端连接电阻R2的一端并作为输出电压采样电路的输出端，电阻R2的另一端接输出地；电阻Rs构成输出电流采样电路，电阻Rs的一端连接负载电阻R的另一端并作为输出电流采样电路的输出端，电阻Rs的另一端连接输出地；The output sampling circuit includes resistors R1 , R2 and Rs. The resistors R1 and R2 form an output voltage sampling circuit. One end of the resistor R1 is connected to the output terminal of the Class D full-bridge rectifier, and the other end of the resistor R1 is connected to the resistor R2 One end of the resistor R2 is used as the output terminal of the output voltage sampling circuit, and the other end of the resistorR2 is connected to the output ground; the resistor Rs constitutes the output current sampling circuit, and one end of the resistor Rs is connected to the other end of the load resistor R as the output terminal of the output current sampling circuit , the other end of the resistor Rs is connected to the output ground;
误差放大电路包括两个运算放大器，其中一个运算放大器的负端连接输出电压采样电路的输出端，另一个运算放大器的负端连接输出电流采样电路的输出端，两个运算放大器的正端均连接输出地；The error amplification circuit includes two operational amplifiers, wherein the negative terminal of one operational amplifier is connected to the output terminal of the output voltage sampling circuit, the negative terminal of the other operational amplifier is connected to the output terminal of the output current sampling circuit, and the positive terminals of the two operational amplifiers are connected to output;
STM32F407微控制器包括A/D变换器、迟滞比较器、数字PI控制器、频率调制器和模式选择电路，误差放大电路中两个运算放大器的输出分别为输出电压的放大信号和输出电流的放大信号，均连接STM32F407微控制器的A/D转换接口，A/D转换后得到的数字电压反馈信号VFB与参考电压VREF进行比较后输出给数字PI控制器，数字PI控制器将电压反馈信号VFB与参考电压VREF之差VE经过比例、积分运算，得到的电压信号输出给频率调制器，频率调制器由STM32F407微控制器中的定时器实现，根据数字PI控制器输出电压的大小得到一对频率可调的互补脉冲输出信号并将其输出给模式选择电路；A/D转换后得到的数字电流反馈信号IFB与参考电流IREF经迟滞比较器后亦输出给模式选择电路，模式选择电路输出G1和G2，G3和G4以及G5和G6三对信号，迟滞比较器的输出确定使能信号的电平，进而决定G1和G2，G3和G4两对输出信号的工作状态；The STM32F407 microcontroller includes an A/D converter, a hysteresis comparator, a digital PI controller, a frequency modulator, and a mode selection circuit. The outputs of the two operational amplifiers in the error amplifier circuit are the amplified signal of the output voltage and the amplified output current. The signals are all connected to the A/D conversion interface of the STM32F407 microcontroller. The digital voltage feedback signal VFB obtained after A/D conversion is compared with the reference voltage VREF and then output to the digital PI controller. The digital PI controller feeds back the voltage The difference VE between the signal VFB and the reference voltage VREF is proportional and integrated, and the obtained voltage signal is output to the frequency modulator. The frequency modulator is realized by the timer in the STM32F407 microcontroller. According to the output voltage of the digital PI controller Get a pair of complementary pulse output signals with adjustable frequency And output it to the mode selection circuit; the digital current feedback signal IFB and the reference current IREF obtained after A/D conversion are also output to the mode selection circuit after passing through the hysteresis comparator, and the mode selection circuit outputs G1 and G2 , G3 and G4 and G5 and G6 three pairs of signals, the output of the hysteresis comparator determines the enable signal Level, and then determine the working status of the two pairs of output signals G1 and G2 , G3 and G4 ;
高频栅驱动电路包括三个相同的驱动电路，每个驱动电路对应连接模式选择电路输出的G1和G2，G3和G4以及G5和G6三对信号中的一对信号，每个驱动电路均设有隔离栅驱动芯片以及并联在隔离栅驱动芯片输出端的两组相同的外部负关断电压产生电路，每组外部负关断电压产生电路均包括电容Cb、二极管Dn、电阻Rg和Rgd，其中一组外部负关断电压产生电路中的电容Cb的一端连接隔离栅驱动芯片的一个输出端，电容Cb的另一端连接二极管Dn的阴极、电阻Rg的一端和电阻Rgd的一端，电阻Rg的另一端输出的信号控制附加有源谐振的谐振网络三个桥臂中其中一个桥臂中的上开关管栅极，电阻Rgd的另一端连接二极管Dn的阳极和隔离栅驱动芯片的另一个输出端，该端输出的信号控制上述桥臂中的上开关管源极；另一组外部负关断电压产生电路中的电容Cb的一端连接隔离栅驱动芯片的第三个输出端，电容Cb的另一端连接二极管Dn的阴极、电阻Rg的一端和电阻Rgd的一端，电阻Rg的另一端输出的信号控制附加有源谐振的谐振网络中上述桥臂的下开关管栅极，电阻Rgd的另一端连接二极管Dn的阳极和隔离栅驱动芯片的第四个输出端，该端输出的信号作为控制附加有源谐振的谐振网络中上述桥臂的下开关管源极；The high-frequency gate drive circuit includes three identical drive circuits, and each drive circuit corresponds to one of the three pairs of signals output by the connection mode selection circuit, G1 and G2 , G3 and G4 , and G5 and G6 , Each drive circuit is equipped with an isolation gate driver chip and two sets of identical external negative turn-off voltage generation circuits connected in parallel at the output end of the isolation gate driver chip. Each set of external negative turn-off voltage generation circuits includes a capacitor Cb , a diode Dn, Resistors Rg and Rgd, one end of capacitorCb in a group of external negative turn-off voltage generation circuits is connected to an output end of the isolation barrier driver chip, and the other end of capacitor Cb is connected to the cathode of diode Dn, one end of resistor Rg and resistor Rgd One end of the resistor Rg, the signal output from the other end of the resistor Rg controls the grid of the upper switching tube in one of the three bridge arms of the additional active resonant resonant network, and the other end of the resistor Rgd is connected to the anode of the diode Dn and the isolation barrier drive The other output terminal of the chip, the signal output from this terminal controls the source of the upper switch tube in the above-mentioned bridge arm; one terminal of the capacitor Cb in the other set of external negative turn-off voltage generation circuit is connected to the third output of the isolation barrier drive chip The other end of the capacitorCb is connected to the cathode of the diode Dn, one end of the resistor Rg, and one end of the resistor Rgd, and the signal output from the other end of the resistor Rg controls the grid of the lower switching tube of the above-mentioned bridge arm in the resonant network with additional active resonance , the other end of the resistor Rgd is connected to the anode of the diode Dn and the fourth output end of the isolation barrier drive chip, and the signal output by this end is used as the source of the lower switching tube of the above-mentioned bridge arm in the resonant network for controlling the additional active resonance;
高频栅驱动电路根据模式选择电路输出的G1和G2，G3和G4以及G5和G6三对信号的工作状态决定附加有源网络的谐振网络的工作模式究竟是A还是B，模式A和B分别对应了两种不同的谐振频率，不同的谐振频率下，附加有源网络的谐振网络中三个桥臂共六个MOS管M1～M6的工作状态不同：The high-frequency gate drive circuit determines whether the working mode of the resonant network of the additional active network is A or B according to the working status of the three pairs of signalsG1 andG2 ,G3 andG4 , andG5 andG6 output by the mode selection circuit , modes A and B respectively correspond to two different resonant frequencies. Under different resonant frequencies, the working states of six MOS tubes M1 to M6 in three bridge arms in the resonant network of the additional active network are different:
如果EN＝1，G3和G4以及G5和G6为脉冲信号，G1和G2为0，附加有源网络的谐振网络工作在模式A；反之，如果EN＝0，G1和G2以及G5和G6为脉冲信号，G3和G4为0，附加有源网络的谐振网络工作在模式B；在轻载情况下，附加有源网络的谐振网络的工作模式由模式A切换到模式B；If EN=1, G3 and G4 and G5 and G6 are pulse signals, G1 and G2 are 0, and the resonant network of the additional active network works in mode A; otherwise, if EN=0, G1 and G2 and G5 and G6 are pulse signals, G3 and G4 are 0, and the resonant network with the additional active network works in mode B; under light load conditions, the working mode of the resonant network with the additional active network is switched from mode A to mode B;
模式A：附加有源网络的谐振网络包括MOS管M3、M4、M5及M6，电感Lr和Lm以及电容Cr，在此模式下，谐振电感为Lr；Mode A: The resonant network of the additional active network includes MOS transistors M3 , M4 , M5 and M6 , inductors Lr and Lm, and capacitor Cr. In this mode, the resonant inductance is Lr;
模式B：附加有源网络的谐振网络包括MOS管M1、M2、M5及M6，电感La、Lr和Lm以及电容Cr，在此模式下，谐振电感为(La+Lr)。Mode B: The resonant network of the additional active network includes MOS transistors M1 , M2 , M5 and M6 , inductors La, Lr and Lm, and capacitor Cr. In this mode, the resonant inductance is (La+Lr).
上述附加有源谐振的谐振网络中的六个MOS管M1～M6均采用碳化硅功率MOS管作为开关管，双谐振频率的LLC谐振电源变换器工作在感性区。The six MOS transistors M1 -M6 in the above-mentioned resonant network with additional active resonance all use silicon carbide power MOS transistors as switching transistors, and the LLC resonant power converter with dual resonant frequencies works in the inductive region.
本发明的优点及显著效果：Advantage of the present invention and remarkable effect:
1)双谐振频率LLC谐振电源变换器具有两个谐振频率，与之对应的，双谐振频率LLC谐振电源变换器具有两种工作模式，根据输出功率的范围来确定变换器工作的模式。1) The dual-resonant-frequency LLC resonant power converter has two resonant frequencies. Correspondingly, the dual-resonant-frequency LLC resonant power converter has two operating modes, and the operating mode of the converter is determined according to the output power range.
2)通过调节工作模式，在不影响重载效率的情况下，大大提高LLC谐振变换器的轻载效率。2) By adjusting the working mode, the light-load efficiency of the LLC resonant converter is greatly improved without affecting the heavy-load efficiency.
3)双谐振频率LLC谐振电源变换器采用碳化硅功率MOS管，并在感性区实现了ZVS，两种工作模式使得开关频率范围较窄，整个输出范围内对称工作。3) The dual resonant frequency LLC resonant power converter adopts silicon carbide power MOS transistors and realizes ZVS in the inductive region. The two working modes make the switching frequency range narrow and work symmetrically in the entire output range.
4)电路简单，无需专用集成电路的复杂控制，成本低，可靠性好。4) The circuit is simple, no complex control of an ASIC is required, the cost is low, and the reliability is good.
附图说明Description of drawings
图1是本发明整体原理图；Fig. 1 is the overall schematic diagram of the present invention;
图2是本发明谐振网络原理图；Fig. 2 is a schematic diagram of the resonant network of the present invention;
图3是本发明谐振网络工作波形图；Fig. 3 is the operating waveform diagram of the resonant network of the present invention;
图4是本发明谐振网络工作模态图；Fig. 4 is a working mode diagram of the resonant network of the present invention;
图5是重载下开关管的关键波形图；Figure 5 is a key waveform diagram of the switching tube under heavy load;
图6是轻载下开关管的关键波形图；Figure 6 is a key waveform diagram of the switching tube under light load;
图7是实际效率与模式选择图；Figure 7 is a diagram of actual efficiency and mode selection;
图8是轻载时的效率比较图。Figure 8 is a comparison chart of efficiency at light load.
具体实施方式Detailed ways
下面结合附图对发明的技术方案进行详细说明。The technical solution of the invention will be described in detail below in conjunction with the accompanying drawings.
如图1所示，传统的LLC谐振电源变换器的谐振网络设有四个NMOS管M3、M4、M5及M6，电感Lr和Lm以及电容Cr，NMOS管M3和M4构成一个桥臂，NMOS管M5和M6构成另一个桥臂，NMOS管M3的源极连接NMOS管M4的漏极和电感Lr的一端，NMOS管M5的源极连接NMOS管M6的漏极和电感Lm的一端，电感Lm的另一端串接电容Cr后连接电感Lr的另一端，NMOS管M3和M5的漏极均连接输入电压VIN正端，NMOS管M4和M6的源极均连接输入电压VIN负端并连接输入地，NMOS管M3、M4、M5及M6的源、漏极之间均分别并联有体二极管和寄生电容。As shown in Figure 1, the resonant network of the traditional LLC resonant power converter is composed of four NMOS transistors M3 , M4 , M5 and M6 , inductors Lr and Lm and capacitor Cr, and NMOS transistors M3 and M4 One bridge arm, NMOS transistorsM5 andM6 form another bridge arm, the source of NMOS transistorM3 is connected to the drain of NMOS transistorM4 and one end of the inductor Lr, and the source of NMOS transistorM5 is connected to NMOS transistorM6 The drain of the inductor Lm and one end of the inductor Lm, the other end of the inductor Lm is connected to the other end of the inductor Lr after the capacitor Cr is connected in series, the drains of the NMOS transistorsM3 andM5 are connected to the positive end of the input voltage VIN , and the NMOS transistorsM4 and The sources of M6 are all connected to the negative terminal of the input voltage VIN and connected to the input ground, and the sources and drains of the NMOS transistors M3 , M4 , M5 and M6 are respectively connected in parallel with body diodes and parasitic capacitances.
本发明在上述电路的基础上，增设包括NMOS管M1、M2和电感La构成的有源网络，该有源网络与包括传统的四个NMOS管M3、M4、M5及M6的LLC谐振电源变换器的谐振网络共同构成附加有源谐振的谐振网络1，其中增设的NMOS管M1和M2构成第三个桥臂，NMOS管M1的源极连接NMOS管M2的漏极和电感La的一端，电感La的另一端连接NMOS管M3源极和NMOS管M4漏极与电感Lr的连接端，NMOS管M1的漏极连接输入电压VIN正端，NMOS管M2的源极连接输入电压VIN负端并连接输入地，NMOS管M1和M2的源、漏极之间均分别并联有体二极管和寄生电容。On the basis of the above circuit, the present invention adds an active network composed of NMOS transistors M1 , M2 and inductor La, and the active network includes traditional four NMOS transistors M3 , M4 , M5 and M6 The resonant network of the LLC resonant power converter together constitutes an additional active resonant resonant network 1, in which the added NMOS transistorsM1 andM2 form the third bridge arm, and the source of the NMOS transistorM1 is connected to the NMOS transistorM2. The drain and one end of the inductor La, the other end of the inductor La is connected to the source of the NMOS transistorM3 and the drain of the NMOS transistorM4 and the connection end of the inductor Lr, the drain of the NMOS transistorM1 is connected to the positive terminal of the input voltage VIN , and the NMOS The source of the transistorM2 is connected to the negative terminal of the input voltage VIN and connected to the input ground, and the sources and drains of the NMOS transistorsM1 andM2 are respectively connected in parallel with a body diode and a parasitic capacitance.
附加有源谐振的谐振网络1的输出依次串接隔离变压器2和Class D全桥整流3，Class D全桥整流3通过负载R依次连接输出采样电路4、误差放大电路5、STM32F407微控制器6和高频栅驱动电路7，高频栅驱动电路7的输出驱动附加有源网络的谐振网络1中三个桥臂共六个MOS管M1～M6的正常工作。The output of the resonant network 1 with additional active resonance is sequentially connected to the isolation transformer 2 and the Class D full-bridge rectifier 3, and the Class D full-bridge rectifier 3 is sequentially connected to the output sampling circuit 4, the error amplifier circuit 5, and the STM32F407 microcontroller 6 through the load R and the high-frequency gate drive circuit 7, the output of the high-frequency gate drive circuit 7 drives the normal operation of six MOS transistors M1 -M6 in three bridge arms in the resonant network 1 of the additional active network.
隔离变压器2原边的同名端和异名端与电感Lm并联连接；The same-named end and the different-named end of the primary side of the isolation transformer 2 are connected in parallel with the inductance Lm;
Class D全桥整流3包括二极管D7、D8、D9、D10和滤波电容Cf，二极管D7的阳极连接二极管D8的阴极和隔离变压器副边的同名端，二极管D9的阳极连接二极管D10的阴极和隔离变压器副边的异名端，二极管D7的阴极连接二极管D9的阴极和滤波电容Cf的正端并作为ClassD全桥整流的输出端连接负载电阻R的一端，二极管D8的阳极连接二极管D10的阳极和滤波电容Cf的负端并连接输出地。Class D full-bridge rectifier 3 includes diodes D7 , D8 , D9 , D10 and filter capacitor Cf , the anode of diode D7 is connected to the cathode of diode D8 and the terminal of the same name on the secondary side of the isolation transformer, and the anode of diode D9 Connect the cathode of diode D10 to the opposite end of the secondary side of the isolation transformer, connect the cathode of diode D7 to the cathode of diode D9 and the positive end of filter capacitor Cf , and connect one end of the load resistor R as the output end of Class D full-bridge rectification , the anode of the diode D8 is connected to the anode of the diode D10 and the negative terminal of the filter capacitor Cf and connected to the output ground.
输出采样电路4包括电阻R1、R2和Rs，电阻R1、R2构成输出电压采样电路，电阻R1的一端连接Class D全桥整流的输出端，电阻R1的另一端连接电阻R2的一端并作为输出电压采样电路的输出端，电阻R2的另一端接输出地；电阻Rs构成输出电流采样电路，电阻Rs的一端连接负载电阻R的另一端并作为输出电流采样电路的输出端，电阻Rs的另一端连接输出地。The output sampling circuit 4 includes resistors R1 , R2 and Rs. The resistors R1 and R2 form an output voltage sampling circuit. One end of the resistor R1 is connected to the output terminal of the Class D full-bridge rectifier, and the other end of the resistor R1 is connected to the resistor R One end of2 is used as the output end of the output voltage sampling circuit, and the other end of the resistor R2 is connected to the output ground; the resistor Rs constitutes the output current sampling circuit, and one end of the resistor Rs is connected to the other end of the load resistor R as the output of the output current sampling circuit end, and the other end of the resistor Rs is connected to the output ground.
误差放大电路5包括两个运算放大器，运算放大器I的负端连接输出电压采样电路的输出端，运算放大器II的负端连接输出电流采样电路的输出端，两个运算放大器的正端均连接输出地。The error amplification circuit 5 includes two operational amplifiers, the negative terminal of the operational amplifier I is connected to the output terminal of the output voltage sampling circuit, the negative terminal of the operational amplifier II is connected to the output terminal of the output current sampling circuit, and the positive terminals of the two operational amplifiers are connected to the output land.
STM32F407微控制器6包括A/D变换器、迟滞比较器、数字PI控制器、频率调制器和模式选择电路，误差放大电路中两个运算放大器的输出分别为输出电压的放大信号和输出电流的放大信号，均连接STM32F407微控制器的A/D转换接口，A/D转换后得到的数字电压反馈信号VFB与参考电压VREF进行比较后输出给数字PI控制器，数字PI控制器将电压反馈信号VFB与参考电压VREF之差VE经过比例、积分运算，得到的电压信号输出给频率调制器，频率调制器由STM32F407微控制器中的定时器实现，根据数字PI控制器输出电压的大小得到一对频率可调的互补脉冲输出信号并将其输出给模式选择电路；A/D转换后得到的数字电流反馈信号IFB与参考电流IREF经迟滞比较器后亦输出给模式选择电路，模式选择电路输出G1和G2，G3和G4以及G5和G6三对信号，迟滞比较器的输出确定使能信号的电平，进而决定G1和G2，G3和G4两对输出信号的工作状态。STM32F407 microcontroller 6 includes A/D converter, hysteresis comparator, digital PI controller, frequency modulator and mode selection circuit. The outputs of the two operational amplifiers in the error amplifier circuit are the amplified signal of the output voltage and the output current respectively. The amplified signal is connected to the A/D conversion interface of the STM32F407 microcontroller. The digital voltage feedback signal VFB obtained after A/D conversion is compared with the reference voltage VREF and then output to the digital PI controller. The digital PI controller converts the voltage The difference VE between the feedback signal VFB and the reference voltage VREF undergoes proportional and integral operations, and the obtained voltage signal is output to the frequency modulator. The frequency modulator is realized by the timer in the STM32F407 microcontroller, and the output voltage of the digital PI controller is The size of a pair of frequency-adjustable complementary pulse output signals And output it to the mode selection circuit; the digital current feedback signal IFB and the reference current IREF obtained after A/D conversion are also output to the mode selection circuit after passing through the hysteresis comparator, and the mode selection circuit outputs G1 and G2 , G3 and G4 and G5 and G6 three pairs of signals, the output of the hysteresis comparator determines the enable signal Level, and then determine the working state of two pairs of output signals G1 and G2 , G3 and G4 .
高频栅驱动电路7包括三个相同的驱动电路，每个驱动电路对应连接模式选择电路输出的G1和G2，G3和G4以及G5和G6三对信号中的一对信号，每个驱动电路均设有隔离栅驱动芯片以及并联在隔离栅驱动芯片输出端的两组相同的外部负关断电压产生电路。两组外部负关断电压产生电路均包括电容Cb、二极管Dn、电阻Rg和Rgd。以图中画出的驱动MOS管M1、M2的桥臂为例，一组外部负关断电压产生电路中的电容Cb1的一端连接隔离栅驱动芯片的一个输出端，电容Cb1的另一端连接二极管Dn1的阴极、电阻Rg1的一端和电阻Rgd1的一端，电阻Rg的另一端输出的信号gM1控制上开关管M1栅极，电阻Rgd1的另一端连接二极管Dn1的阳极和隔离栅驱动芯片的另一个输出端，该端输出的信号SM1控制上开关管M1源极；另一组外部负关断电压产生电路中的电容Cb2的一端连接隔离栅驱动芯片的第三个输出端，电容Cb2的另一端连接二极管Dn2的阴极、电阻Rg2的一端和电阻Rgd2的一端，电阻Rg2的另一端输出的信号gM2控制下开关管M2栅极，电阻Rg2的另一端连接二极管Dn2的阳极和隔离栅驱动芯片的第四个输出端，该端输出的信号SM2控制下开关管M2源极。驱动MOS管M3、M4桥臂和驱动MOS管M5、M6桥臂的电路与驱动MOS管M1、M2桥臂的电路相同(未示出)。The high-frequency gate drive circuit 7 includes three identical drive circuits, and each drive circuit corresponds to one of the three pairs of signalsG1 andG2 ,G3 andG4 , andG5 andG6 output by the connection mode selection circuit Each drive circuit is provided with an isolation gate drive chip and two sets of identical external negative turn-off voltage generating circuits connected in parallel at the output terminals of the isolation gate drive chip. Both sets of external negative turn-off voltage generating circuits include capacitor Cb , diode Dn , resistors Rg and Rgd. Take the bridge arms of driving MOS transistors M1 and M2 shown in the figure as an example, one end of the capacitor Cb1 in a group of external negative turn-off voltage generation circuits is connected to an output end of the isolation gate driver chip, and the capacitor Cb1 The other end is connected to the cathode of the diodeDn1 , one end of the resistorRg1 and one end of the resistorRgd1 , the signal g M1 output from the other end of the resistorRg controls the gate of the upper switching tubeM1 , and the other end of the resistorRgd1 is connected to the diodeDn1 The anode of the anode and the other output end of the isolation gate drive chip, the signal SM1 output from this end controls the source of the upper switch tubeM1 ; one end of the capacitor Cb2 in the other set of external negative turn-off voltage generation circuits is connected to the isolation gate drive The third output terminal of the chip, the other end of the capacitor Cb2 is connected to the cathode of the diode Dn2 , one end of the resistor Rg2 and one end of the resistor Rgd2 , and the signal gM2 output from the other end of the resistor Rg2 controls the lower switch tube M2 The gate, the other end of the resistor Rg2 is connected to the anode of the diode Dn2 and the fourth output end of the isolation gate driver chip, and the signal SM2 output from this end controls the source of the lower switching tube M2 . The circuit for driving the bridge arms of MOS transistors M3 and M4 and the bridge arms of MOS transistors M5 and M6 is the same as the circuit for driving the bridge arms of MOS transistors M1 and M2 (not shown).
高频栅驱动电路7根据模式选择电路输出的G1和G2，G3和G4以及G5和G6三对信号的工作状态决定附加有源网络的谐振网络1的工作模式究竟是A还是B，模式A和B分别对应了两种不同的谐振频率，不同的谐振频率下，附加有源网络的谐振网络中三个桥臂共六个MOS管M1～M6的工作状态不同：The high-frequency gate drivecircuit 7 determines whether the working mode of the resonant network1 of the additional active networkis A Still B, modes A and B respectively correspond to two different resonant frequencies. Under different resonant frequencies, the working states of the six MOS tubes M1 to M6 in the three bridge arms in the resonant network of the additional active network are different:
如果EN＝1，G3和G4以及G5和G6为脉冲信号，G1和G2为0，附加有源网络的谐振网络工作在模式A；反之，如果EN＝0，G1和G2以及G5和G6为脉冲信号，附加有源网络的谐振网络工作在模式B；在轻载情况下，附加有源网络的谐振网络的工作模式由模式A切换到模式B。If EN=1, G3 and G4 and G5 and G6 are pulse signals, G1 and G2 are 0, and the resonant network of the additional active network works in mode A; otherwise, if EN=0, G1 and G G2 , G5 and G6 are pulse signals, and the resonant network of the additional active network works in mode B; under light load conditions, the working mode of the resonant network of the additional active network is switched from mode A to mode B.
模式A：附加有源网络的谐振网络包括MOS管M3、M4、M5及M6，电感Lr和Lm以及电容Cr，在此模式下，谐振电感为Lr。Mode A: The resonant network of the additional active network includes MOS transistors M3 , M4 , M5 and M6 , inductors Lr and Lm, and capacitor Cr. In this mode, the resonant inductance is Lr.
模式B：附加有源网络的谐振网络包括MOS管M1、M2、M5及M6，电感La、Lr和Lm以及电容Cr，在此模式下，谐振电感为(La+Lr)。Mode B: The resonant network of the additional active network includes MOS transistors M1 , M2 , M5 and M6 , inductors La, Lr and Lm, and capacitor Cr. In this mode, the resonant inductance is (La+Lr).
上述附加有源谐振的谐振网络中的六个MOS管M1～M6均采用碳化硅功率MOS管作为开关管，双谐振频率的LLC谐振电源变换器工作在感性区。The six MOS transistors M1 -M6 in the above-mentioned resonant network with additional active resonance all use silicon carbide power MOS transistors as switching transistors, and the LLC resonant power converter with dual resonant frequencies works in the inductive region.
如图2所示，所述带附件有源网络的谐振网络1与传统的LLC谐振网络相比，增加了由一对小功率开关管M1～M2和辅助电感La组成的有源网络。谐振网络1中的每个开关管受50％占空比的方波控制，每个桥臂上下两管之间插入一段死区时间防止发生直通现象。D1～D6和C1～C6分别为M1～M6的体二极管和寄生电容。As shown in FIG. 2 , compared with the traditional LLC resonant network, the resonant network 1 with an accessory active network adds an active network composed of a pair of low-power switch tubes M1 -M2 and an auxiliary inductor La. Each switching tube in the resonant network 1 is controlled by a square wave with a duty ratio of 50%, and a dead time is inserted between the upper and lower tubes of each bridge arm to prevent shoot-through. D1 -D6 and C1 -C6 are body diodes and parasitic capacitances of M1 -M6 respectively.
如图3所示，为开关频率小于谐振频率下，双谐振频率LLC谐振电源变换器的关键波形图。VGS为高电平表示NMOS导通，M3～M6工作在模式A，M1～M2和M5～M6工作在模式B。VPRI为隔离变压器2的原边电压，VDS(M6)为M6的漏源电压。iD(M6)和iD(M5)分别为M6和M5的导通电流，iM为电感Lm上的电流。iPRI为谐振电感上的电流。Vrect(D8，D9)、Vrect(D7，D10)为Class D全桥整流的二极管D8，D9和D7，D10上的电压。As shown in Fig. 3, it is a key waveform diagram of a dual-resonant frequency LLC resonant power converter when the switching frequency is lower than the resonant frequency. When VGS is at a high level, it means that the NMOS is turned on, M3 -M6 work in mode A, and M1 -M2 and M5 -M6 work in mode B. VPRI is the primary side voltage of the isolation transformer 2, and VDS(M6) is the drain-source voltage of M6. iD(M6) and iD(M5) are the conduction currents of M6 and M5 respectively, and iM is the current on the inductor Lm. iPRI is the current on the resonant inductor. Vrect(D8, D9) and Vrect(D7, D10) are voltages on diodes D8 , D9 and D7 , D10 of Class D full-bridge rectification.
如图4所示，图中器件标号上的删除线表示在此工作阶段该器件不工作。以模式A为例，双谐振频率LLC谐振变化器具体工作模态如下：As shown in Figure 4, the strikeout line on the device label in the figure indicates that the device does not work during this working phase. Taking mode A as an example, the specific working mode of the dual-resonance frequency LLC resonance changer is as follows:
1)[t0～t1]阶段：t0时刻，M3和M6因体二极管反向偏置而导通，输入电压Vin全部加在谐振网络，谐振电流ir(t)和励磁电感电流iM均正向增加。谐振电流ir(t)以频率为fs的正弦波的形式增加。由于D7和D10正向偏置，Lm上的电压为副边输出电压反射到原边的电压nVo。这一阶段持续到励磁电流增加到与谐振电路相等。1) [t0～t1] stage: at time t0, M3 and M6 are turned on due to the reverse bias of the body diode, the input voltage Vin is all applied to the resonant network, and the resonant current ir(t) and the exciting inductor current iM both increase positively . The resonant current ir(t) increases in the form of a sine wave with frequency fs . Due to the forward bias of D7 and D10, the voltage on Lm is the voltage nVo reflected from the output voltage of the secondary side to the primary side. This phase continues until the field current increases to equal the resonant circuit.
2)[t1～t2]阶段：t1时刻，励磁电流增加到与谐振电路相等，因此变压器原边绕组的电流下降为0，因此，D7和D10为ZCS关断。在此阶段，没有能量传递到副边。这一阶段持续到t2，此时对角开关管关断。2) [t1～t2] stage: At t1, the excitation current increases to be equal to the resonant circuit, so the current of the primary winding of the transformer drops to 0, therefore, D7 and D10 are turned off for ZCS. At this stage, no energy is transferred to the secondary side. This phase lasts until t2, when the diagonal switch is turned off.
3)[t2～t3]阶段：t2时刻，关断M3和M6。VDS(M3)和VDS(M6)由于寄生电容C3～C6的作用逐渐从0开始上升，与此同时VDS(M4)和VDS(M5)逐渐从Vin下降到0。由于谐振网络的电流变化滞后于电压的变化，所以谐振电流ir(t)开始对C3～C6充电。3) [t2-t3] stage: at t2, turn off M3 and M6. VDS(M3) and VDS(M6) gradually rise from 0 due to the effect of parasitic capacitances C3~C6, while VDS(M4) and VDS(M5) gradually drop from Vin to 0. Because the current change of the resonant network lags behind the voltage change, the resonant current ir(t) begins to charge C3-C6.
4)[t3～t4]阶段：VDS(M3)和VDS(M6)在t3时刻增加到Vin，并在此阶段保持Vin不变。VDS(M4)和VDS(M5)在t3时刻下降到0。因而，D4和D5正向偏置。谐振网络上的电压为(-Vin)，励磁电流和谐振电流开始下降，iD(M4)和iD(M5)为负。这一阶段维持到t4，此时VGS(M4)和VGS(M5)被触发，iD(M4)和iD(M5)达到最大值。4) [t3-t4] stage: VDS (M3) and VDS (M6) increase to Vin at t3, and keep Vin unchanged at this stage. VDS(M4) and VDS(M5) drop to 0 at t3. Thus,D4 andD5 are forward biased. The voltage on the resonant network is (-Vin), the excitation current and the resonant current start to drop, iD(M4) and iD(M5) are negative. This stage is maintained until t4, when VGS(M4) and VGS(M5) are triggered, and iD(M4) and iD(M5) reach the maximum value.
5)[t4～t5]阶段：t4时刻，开关管正向偏置，iD(M4)和iD(M5)开始下降。当反向电流达到0时，原边电流开始正向流过开关管。t5时刻与t0时刻类似。5) [t4-t5] stage: At t4, the switching tube is forward-biased, and iD(M4) and iD(M5) begin to decrease. When the reverse current reaches 0, the primary current begins to flow forward through the switch tube. Time t5 is similar to timet0 .
开关管的损耗分为三部分：栅驱动损耗Pdrive、导通损耗Pcond和开关损耗Psw。在特定的负载条件下，Pcond在整个开关频率范围内基本保持不变。因此Pcond几乎不影响任何开关模式下的效率。Pdrive和Psw受频率影响，而Pdrive远远低于Psw，因而Pdrive可以忽略不计。The loss of the switching tube is divided into three parts: gate drive loss Pdrive, conduction loss Pcond and switching loss Psw. Under certain load conditions, Pcond remains essentially constant over the entire switching frequency range. Therefore Pcond hardly affects any switching mode efficiency. Pdrive and Psw are affected by frequency, and Pdrive is much lower than Psw, so Pdrive can be ignored.
本发明工作过程如下：The working process of the present invention is as follows:
在LLC谐振全桥变换器中，由于开关管是ZVS导通，因而Psw仅仅是关断时刻的关断损耗，而且，Psw随着开关频率fs的增加而增加。根据频率调制原则，LLC谐振变换器在轻载时，其开关频率fs是增加的，所以轻载时Psw也是增加的，轻载效率降低。为了减小轻载时的Psw，设计了根据负载电流大小而切换LLC谐振网络的工作模式的方案：在轻载情况下，附加有源网络的谐振网络的工作模式由模式A切换到模式B。In the LLC resonant full-bridge converter, because the switch tube is ZVS conduction, Psw is only the turn-off loss at the turn-off moment, and Psw increases with the increase of the switching frequency fs. According to the principle of frequency modulation, when the LLC resonant converter is light-loaded, its switching frequency fs increases, so Psw also increases when it is light-loaded, and the light-load efficiency decreases. In order to reduce the Psw at light load, a scheme is designed to switch the working mode of the LLC resonant network according to the load current: in the case of light load, the working mode of the resonant network with the additional active network is switched from mode A to mode B.
本发明中，采用STM32F407微控制器实现该控制。输出电压和输出电流被采样，放大并转换为数字反馈信号VFB和IFB。内部的PI功能单元计算VFB与参考电压VREF的差VE，将此值作为频率调制器的输入信号，频率调制器产生两个互补的脉冲信号。除此之外，迟滞比较器功能单元也用来实现模式控制，内部的参考电流IREF与滞环IHYS都用来控制模式的开关。模式选择电路基于比较器的输出来决定如果EN＝1，G3和G4作为脉冲信号，G1和G2信号为0，因此变换器工作在模式A。相反的，如果EN＝0，变换器工作在模式B。In the present invention, the STM32F407 microcontroller is used to realize the control. The output voltage and output current are sampled, amplified and converted into digital feedback signals VFB and IFB. The internal PI function unit calculates the difference VE between VFB and the reference voltage VREF, and uses this value as the input signal of the frequency modulator, and the frequency modulator generates two complementary pulse signals. In addition, the hysteresis comparator functional unit is also used to implement mode control, and the internal reference current IREF and hysteresis loop IHYS are both used to control the mode switch. The mode selection circuit decides based on the output of the comparator If EN=1, G3 and G4 are used as pulse signals, G1 and G2 signals are 0, so the converter works in mode A. Conversely, if EN=0, the converter operates in mode B.
下面以样机为例，描述本发明：Take the prototype as an example below to describe the present invention:
参数及说明如下：The parameters and description are as follows:
Vin＝400V，Vo＝100V，Po(max)＝1200W，开关频率500～800kHz，满载500kHz。Vin＝400V, Vo＝100V, Po(max)＝1200W, switching frequency 500～800kHz, full load 500kHz.
如图5所示为本发明重载下M6的关键波形图，开关频率为600kHz，输出电压为120V，输出电流为10A。由图可见在重载条件下，M6实现了ZVS导通。由于开关管是对称工作的，因此所有的开关管均实现了ZVS导通。As shown in Fig. 5, the key waveform diagram ofM6 under heavy load of the present invention, the switching frequency is 600kHz, the output voltage is 120V, and the output current is 10A. It can be seen from the figure that under heavy load conditions,M6 realizes ZVS conduction. Since the switch tubes work symmetrically, all the switch tubes realize ZVS conduction.
如图6所示为本发明轻载条件下M6的关键波形图，开关频率为680kHz，输出电压为100V，输出电流为4A。由图可见在轻载条件下，M6实现了ZVS导通。由于开关管是对称工作的，因此所有的开关管均实现了ZVS导通。As shown in Figure 6, the key waveform diagram ofM6 under the light load condition of the present invention, the switching frequency is 680kHz, the output voltage is 100V, and the output current is 4A. It can be seen from the figure that under light load conditions,M6 realizes ZVS conduction. Since the switch tubes work symmetrically, all the switch tubes realize ZVS conduction.
如图7所示，在输入Vin为400V，输出Vo为100V的条件下实际测试了本发明LLC谐振变换器的效率，并且在整个输出功率范围内均实现了软开关过程。变换器的最大效率在输出为1kW(83.3％输出功率)时测得，在0.5～1.2kW(40％～100％输出功率)范围内效率均维持在93％以上。As shown in Fig. 7, the efficiency of the LLC resonant converter of the present invention is actually tested under the condition that the input Vin is 400V and the output Vo is 100V, and the soft switching process is realized in the whole output power range. The maximum efficiency of the converter is measured when the output is 1kW (83.3% output power), and the efficiency is maintained above 93% in the range of 0.5-1.2kW (40%-100% output power).
如图7所示，输出功率在500～700W的范围内，两种模式下的变换器效率基本一样，因此模式切换设置如下：As shown in Figure 7, when the output power is in the range of 500-700W, the efficiency of the converter in the two modes is basically the same, so the mode switching settings are as follows:
a)当输出保持满载，变换器工作在模式A；a) When the output remains at full load, the converter works in mode A;
b)当输出功率低于500W，变换器工作在模式B，一直维持到输出功率高于700W。b) When the output power is lower than 500W, the converter works in mode B until the output power is higher than 700W.
如图8所示，为轻载时的效率比较图。可见采用本发明LLC谐振变换器的轻载效率比不采用时得到较大的提高。As shown in Figure 8, it is a comparison chart of efficiency at light load. It can be seen that the light-load efficiency of the LLC resonant converter of the present invention is greatly improved compared with that of the non-use.
| Application Number | Priority Date | Filing Date | Title |
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| CN201610575347.8ACN106059314B (en) | 2016-07-21 | 2016-07-21 | A kind of LLC resonant power converters with dual resonance frequency |
| Application Number | Priority Date | Filing Date | Title |
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| CN201610575347.8ACN106059314B (en) | 2016-07-21 | 2016-07-21 | A kind of LLC resonant power converters with dual resonance frequency |
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| CN201610575347.8AActiveCN106059314B (en) | 2016-07-21 | 2016-07-21 | A kind of LLC resonant power converters with dual resonance frequency |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN106786667B (en)* | 2016-12-23 | 2019-04-16 | 芜湖国睿兆伏电子有限公司 | A kind of phase shift frequency modulation mixing control circuit for LLC resonant power |
| CN106655872B (en)* | 2017-02-14 | 2023-05-23 | 华南理工大学 | A series transformer type LLC positive and negative pulse dual battery charging power supply system |
| CN108720081A (en)* | 2017-04-13 | 2018-11-02 | 湖南中烟工业有限责任公司 | A kind of ultrasonic electronic cigarette circuit and implementation method |
| CN107395022B (en)* | 2017-07-25 | 2023-08-08 | 杭州士兰微电子股份有限公司 | Resonant switching converter and control method thereof |
| CN107359799A (en)* | 2017-07-28 | 2017-11-17 | 西南交通大学 | A kind of control method and its device of LCC resonance DC DC converters |
| CN109510501B (en)* | 2017-09-12 | 2021-07-09 | 华为技术有限公司 | A soft switching converter and wireless charging system |
| CN109980941B (en)* | 2019-03-20 | 2021-04-13 | 深圳市皓文电子有限公司 | Switch control unit of LCC resonant DCDC converter and converter |
| CN110212767B (en)* | 2019-04-30 | 2020-08-04 | 东南大学 | A Digital Control Method for Realizing Multi-step Frequency Modulation of LLC Resonant Converter |
| CN110601543B (en)* | 2019-09-11 | 2020-08-18 | 广州金升阳科技有限公司 | Wide gain control method of LLC resonant converter and resonant converter thereof |
| TWI768454B (en)* | 2020-09-02 | 2022-06-21 | 僑威科技股份有限公司 | High-efficiency llc resonant converter |
| CN112713782B (en)* | 2021-03-29 | 2021-07-13 | 深圳市正浩创新科技股份有限公司 | Resonant converter and its synchronous rectification control method |
| TWI752891B (en)* | 2021-06-25 | 2022-01-11 | 台達電子工業股份有限公司 | Llc resonance converter, control unit, and method of controlling the same |
| CN115001282A (en)* | 2022-06-07 | 2022-09-02 | 华为电动技术有限公司 | Resonant converter and control method and system thereof |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN104022675A (en)* | 2014-05-29 | 2014-09-03 | 燕山大学 | Single-stage bidirectional isolation AC-DC converter |
| CN104716843A (en)* | 2015-03-16 | 2015-06-17 | 哈尔滨工业大学深圳研究生院 | High-efficiency disconnecting switch power source suitable for wide-voltage-range inputting |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JP2004120863A (en)* | 2002-09-25 | 2004-04-15 | Sony Corp | Switching power circuit |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN104022675A (en)* | 2014-05-29 | 2014-09-03 | 燕山大学 | Single-stage bidirectional isolation AC-DC converter |
| CN104716843A (en)* | 2015-03-16 | 2015-06-17 | 哈尔滨工业大学深圳研究生院 | High-efficiency disconnecting switch power source suitable for wide-voltage-range inputting |
| Publication number | Publication date |
|---|---|
| CN106059314A (en) | 2016-10-26 |
| Publication | Publication Date | Title |
|---|---|---|
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