技术领域technical field
本发明涉及一种电流型单向DC-DC变换器及用于对变换器进行控制的对称双PWM加移相控制方法，尤其涉及一种隔离式电流型单向DC-DC变换器及用于对变换器进行控制的对称双PWM加移相控制方法，属于电力电子领域的高频开关电源方向。The present invention relates to a current-type unidirectional DC-DC converter and a symmetrical double PWM plus phase-shift control method for controlling the converter, in particular to an isolated current-type unidirectional DC-DC converter and a method for controlling the converter. The invention relates to a symmetrical dual PWM plus phase-shift control method for controlling a converter, which belongs to the direction of high-frequency switching power supply in the field of power electronics.
背景技术Background technique
对于燃料电池和光伏发电等电力电子应用场合，有着很宽的输入电压范围，并希望有尽可能小的输入电流纹波。如果输入电流纹波过大，将会缩短燃料电池的寿命，光伏组的最大跟踪效率也会受到影响。对DC-DC变换器。高的传递效率是特别重要的要求。功率器件的零电压开关(ZVS)可以提高变换器效率。对于实现零电压软开关，移相全桥(PSFB)DC-DC变换器是一种很流行的方法。但是有一些问题需要被克服。首先，轻载下，滞后桥臂不能实现软开关；其次，由于变压器漏感引起的占空比丢失在低压输入时特别严重。占空比丢失使得变换器在正常变比时很难输出额定电压。通过提高变压器变比能够对占空比丢失进行补偿，但是这会提高输出整流管的电压应力和原边的电流应力。因为耐压高的二极管有着更高的导通压降，所以这会提高二极管导通损耗，同时原边开关管的导通损耗也会增大。为了降低副边整流二极管的电压尖峰，需要额外的箝位电路。所以，在低压输入时移相全桥DC-DC变换器的效率会受到很大限制。因此，移相全桥DC-DC变换器在低压输入高压输出场合是不适用的。For power electronics applications such as fuel cells and photovoltaic power generation, there is a wide input voltage range and it is desirable to have as small an input current ripple as possible. If the input current ripple is too large, the life of the fuel cell will be shortened, and the maximum tracking efficiency of the photovoltaic group will also be affected. For DC-DC converters. A high transfer efficiency is a particularly important requirement. Zero-voltage switching (ZVS) of power devices can improve converter efficiency. Phase-shifted full-bridge (PSFB) DC-DC converters are a popular approach to realize zero-voltage soft switching. But there are some problems that need to be overcome. First, under light load, the lagging bridge arm cannot realize soft switching; second, the loss of duty cycle due to transformer leakage inductance is particularly serious at low voltage input. The loss of duty cycle makes it difficult for the converter to output the rated voltage at normal ratio. The loss of duty cycle can be compensated by increasing the transformation ratio of the transformer, but this will increase the voltage stress of the output rectifier and the current stress of the primary side. Because a diode with a high withstand voltage has a higher conduction voltage drop, this will increase the conduction loss of the diode, and at the same time, the conduction loss of the primary switch will also increase. In order to reduce the voltage spike of the rectifier diode on the secondary side, an additional clamping circuit is required. Therefore, the efficiency of the phase-shifted full-bridge DC-DC converter will be greatly limited when the input voltage is low. Therefore, the phase-shifting full-bridge DC-DC converter is not suitable for low-voltage input and high-voltage output applications.
一种由全桥构成的电压型混合谐振DC-DC变换器在IEEE Transaction onindustry application【电力电子期刊】于2014年发表的“A Hybrid Resonant ConverterUtilizing a Bidirectional GaN AC Switch for High-Efficiency PV Applications”【一种采用GaN AC开关管的混合谐振高效光伏双向变换器】一文中被提出。变压器漏感既作为谐振电感又作为升压电感。虽然可以实现零电压开关，但是控制器的结构必须根据输入电压的值在降压(buck)模式和升压(boost)模式之间切换，会增加设计的难度。IEEETransaction on power electronics【电力电子期刊】于2015年发表的文献“A Family ofSoft-Switching DC–DC Converters Based on a Phase-Shift-Controlled ActiveBoost Rectifier”【基于移相控制有源升压整流的软开关DC-DC变换器族】中的单相移控制使得控制可以忽视电压的变化，但由于开关管工作在硬开关条件，变换器需要额外的箝位电路。A voltage-type hybrid resonant DC-DC converter composed of a full bridge was published in "A Hybrid Resonant Converter Utilizing a Bidirectional GaN AC Switch for High-Efficiency PV Applications" in IEEE Transaction onindustry application [Journal of Power Electronics] in 2014 [一A hybrid resonant high-efficiency photovoltaic bidirectional converter using GaN AC switching tube] is proposed in this paper. The transformer leakage inductance acts as both the resonant inductor and the boost inductor. Although zero-voltage switching can be realized, the structure of the controller must switch between buck mode and boost mode according to the value of the input voltage, which will increase the difficulty of design. IEEE Transaction on power electronics [Journal of Power Electronics] published in 2015 "A Family of Soft-Switching DC–DC Converters Based on a Phase-Shift-Controlled ActiveBoost Rectifier" [based on phase-shift control active boost rectifier soft-switching DC -The single-phase-shift control in the DC converter family] enables the control to ignore voltage changes, but since the switching tube works under hard switching conditions, the converter requires an additional clamping circuit.
升压半桥不仅可以降低输入电流纹波，同时也可以减少附加的启动电路。使用传统的脉宽控制不能实现在宽输入电压范围内所有开关管的软开关。可以如IEEETransaction on power electronics【电力电子期刊】于2013年发表的文献“Anintegrated boost resonant converter for photovoltaic applications”【一种应用于光伏场合的集成升压谐振电感的变换器】中所述，将变压器漏感设计成与输入侧箝位电容和输出侧电容谐振的方式，此时输出二极管可以实现零电流开关(ZCS)，但输入侧的下管会丢失零电压开关(ZVS)。The boost half-bridge can not only reduce the input current ripple, but also reduce the additional startup circuit. The soft switching of all switching tubes within a wide input voltage range cannot be realized using traditional pulse width control. The transformer leakage The inductance is designed to resonate with the input-side clamp capacitor and the output-side capacitor. At this time, the output diode can realize zero-current switching (ZCS), but the lower tube on the input side will lose zero-voltage switching (ZVS).
发明内容Contents of the invention
为了克服上述的变换器不能实现全负载范围内软开关、控制器设计复杂、需额外箝位电路等问题，本发明公开的一种电流型单向DC-DC变换器及用于对变换器进行控制的对称双PWM加移相控制方法，要解决的技术问题是，能够减小电流应力和环流损耗以及实现电路输入电流的低纹波运行，减小输入侧开关管电流应力和变换器的环流损耗，实现输出侧半有源全桥开关管的ZVS软开关和二极管的ZCS开关，提高变换器的效率和可靠性。In order to overcome the problems that the above-mentioned converter cannot realize soft switching in the full load range, the controller design is complicated, and an additional clamping circuit is required, the present invention discloses a current-mode unidirectional DC-DC converter and a Controlled symmetrical dual PWM plus phase-shift control method, the technical problem to be solved is to reduce the current stress and circulation loss and realize the low ripple operation of the circuit input current, reduce the current stress of the switch tube on the input side and the circulation current of the converter Loss, realize the ZVS soft switching of the semi-active full-bridge switching tube on the output side and the ZCS switching of the diode, and improve the efficiency and reliability of the converter.
本发明公开的一种电流型单向DC-DC变换器及用于对变换器进行控制的对称双PWM加移相控制方法。变换器由主电路和控制电路组成；所述主电路由输入侧和输出侧组成；控制电路包括控制器和驱动电路。基于提出的拓扑，采用对称双PWM加移相控制方法。通过同时调节占空比及输入侧和输出侧电压之间的移相角，减小在功率不传递的阶段漏感电流有效值，降低电流造成的电路通态损耗和环流损耗，同时降低电流峰值，减小功率器件应力。输出侧二极管可在全范围内实现零电流开关。移相角较小时，利用变压器漏感中偏置电流实现输出侧开关管的零电压开关，从而保证开关管在全负载范围内实现软开关，进一步降低电路损耗。经验证，本发明提出的电路拓扑和控制方法能够减小电流应力和环流损耗以及实现电路输入电流的低纹波运行，能够提高变换器的效率和可靠性。The invention discloses a current-type unidirectional DC-DC converter and a symmetrical double PWM plus phase-shift control method for controlling the converter. The converter is composed of a main circuit and a control circuit; the main circuit is composed of an input side and an output side; the control circuit includes a controller and a drive circuit. Based on the proposed topology, a symmetrical dual PWM plus phase-shift control method is adopted. By adjusting the duty cycle and the phase shift angle between the input side and output side voltage at the same time, the effective value of the leakage inductance current is reduced in the stage where the power is not transmitted, the circuit on-state loss and circulation loss caused by the current are reduced, and the current peak value is reduced at the same time. , Reduce the stress of power devices. Output-side diodes enable zero-current switching over the full range. When the phase shift angle is small, the bias current in the leakage inductance of the transformer is used to realize the zero-voltage switching of the switching tube on the output side, so as to ensure the soft switching of the switching tube in the full load range and further reduce the circuit loss. It has been verified that the circuit topology and control method proposed by the invention can reduce current stress and circulation loss, realize low ripple operation of circuit input current, and improve the efficiency and reliability of the converter.
本发明的目的是通过下述技术方案实现的。The purpose of the present invention is achieved through the following technical solutions.
一种电流型单向DC-DC变换器，主要由主电路和控制电路组成。主电路主要由输入侧和输出侧构成，输入侧用于将输入的低压进行第一次升压，从而降低对变压器变比的要求，减小输入侧开关管的电流应力，并且减小输入的纹波；输出侧用于实现对变压器交流电压的整流，实现额定电压输出；所述的输入侧包括直流电感(L1、L2)和由开关管(Q1、Q2、Q1a、Q2a)和箝位电容(Cd1、Cd2)组成的箝位电路，输出侧由半有源全桥和输出电容(Co)组成，半有源全桥包括开关管(S3、S4)和二极管(Dr1、Dr2)。A current-mode unidirectional DC-DC converter is mainly composed of a main circuit and a control circuit. The main circuit is mainly composed of an input side and an output side. The input side is used to boost the input low voltage for the first time, thereby reducing the requirements for the transformation ratio of the transformer, reducing the current stress of the switching tube on the input side, and reducing the input voltage. Ripple; the output side is used to rectify the AC voltage of the transformer to achieve rated voltage output; the input side includes DC inductance (L1 , L2 ) and switching tubes (Q1 , Q2 , Q1a , Q2a ) and clamping capacitors (Cd1 , Cd2 ), the output side is composed of a semi-active full-bridge and an output capacitor (Co ), and the semi-active full-bridge includes switches (S3 , S4 ) and diodes (Dr1 , Dr2 ).
连接关系是：输入侧的直流电感L1和L2的一端同时和输入电源正极相连，直流电感L1和直流电感L2的另一端分别与升压半桥的中点a和b相连，同时a点分别与开关管Q1a的源极、Q1的漏极、电感Lr一端相连，开关管Q1a的漏极与箝位电容Cd1一端连接，开关管Q2a的漏极连接到箝位电容Cd2一端，开关管Q1的源极、开关管Q2的源极、箝位电容Cd1与箝位电容Cd2的一端连接到输入电源负极，电感Lr的另一端与变压器Tr原边一端连接，变压器Tr原边另一端连接到b点；输入侧和输出侧通过变压器Tr产生电气连接；变压器Tr副边的两端分别连接到半有源全桥的中点c和d，c点与二极管Dr1的阳极和开关管S3的漏极相连，d点与Dr2的阳极和开关管S4的漏极相连，二极管Dr1、二极管Dr2的阴极、输出电容Co的一端都连接到输出负载的一端，开关管S3、开关管S4的阴极、Co的一端同时与负载的另一端相连。The connection relationship is: one end of the DC inductance L1 and L2 on the input side is connected to the positive pole of the input power supply at the same time, and the other ends of the DC inductance L1 and DC inductance L2 are respectively connected to the midpoints a and b of the boost half bridge, and at the same time The point a is respectively connected to the source of the switching tubeQ1a , the drain of Q1, andone end of the inductorLr , the drain of the switching tubeQ1a is connected to one end of the clamp capacitorCd1 , and the drain of the switching tubeQ2a is connected to the clamp One end of bit capacitor Cd2 , the source of switching tube Q1, the source of switching tubeQ2 , clamping capacitor Cd1 and one end of clamping capacitor Cd2 are connected to the negativepole of the input power supply, and the other end of inductor Lr is connected to the transformer Tr One end of the primary side is connected, and the other end of the primary side of the transformer Tr is connected to point b; the input side and the output side are electrically connected through the transformer Tr; the two ends of the secondary side of the transformer Tr are respectively connected to midpoints c and d of the semi-active full bridge, Point c is connected to the anode of diode Dr1 and the drain of switch tube S3, pointd is connected to the anode of Dr2 and the drain of switch tubeS4 , the cathode of diode Dr1 , diode Dr2 , and the output capacitor Co One end is connected to one end of the output load, and the cathodes of the switching tube S3 , the switching tube S4 , and one end of Co are connected to the other end of the load at the same time.
DC-DC变换器控制电路主要由控制器和驱动电路构成；控制器是以DSP控制器为核心，用于对由传感器采样得到的电压电流采样信号进行转换，并依据对称双PWM加移相控制方法产生PWM驱动信号，调节占空比及输入侧和输出侧电压之间的移相角，能够减小功率不传递的阶段漏感电流有效值，降低电流造成的电路通态损耗和环流损耗，同时降低电流峰值，减小开关管和二极管应力；使输出侧二极管在全范围内实现零电流开关；移相角较小时，利用变压器漏感中偏置电流实现输出侧开关管的零电压开关，从而保证开关管在全负载范围内实现软开关，进一步降低电路损耗。驱动电路用于接收来自控制器的PWM信号，经过隔离和电压增强后为主电路的开关管(Q1、Q2、Q1a、Q2a、S3、S4)提供驱动电压。The DC-DC converter control circuit is mainly composed of a controller and a drive circuit; the controller is based on a DSP controller, which is used to convert the voltage and current sampling signals obtained from the sensor sampling, and is controlled according to the symmetrical dual PWM plus phase shifting. The method generates a PWM drive signal, adjusts the duty cycle and the phase shift angle between the input side and the output side voltage, can reduce the effective value of the leakage inductance current in the stage where the power is not transmitted, and reduce the circuit on-state loss and circulation loss caused by the current. At the same time, the current peak value is reduced, and the stress of the switch tube and diode is reduced; the output side diode can realize zero current switching in the whole range; when the phase shift angle is small, the bias current in the leakage inductance of the transformer can be used to realize the zero voltage switch of the output side switch tube, Therefore, it is ensured that the switching tube realizes soft switching in the full load range, further reducing circuit loss. The driving circuit is used to receive the PWM signal from the controller, and provide driving voltage for the switching tubes (Q1 , Q2 , Q1a , Q2a , S3 , S4 ) of the main circuit after isolation and voltage enhancement.
作为优选，所述开关管(Q1、Q2、Q1a、Q2a、S3、S4)为存在反并联的体二极管和漏源极的寄生电容的开关管。Preferably, the switch tubes (Q1 , Q2 , Q1a , Q2a , S3 , S4 ) are switch tubes with anti-parallel body diodes and drain-source parasitic capacitances.
用于对一种电流型单向DC-DC变换器控制的对称双PWM加移相控制方法，包括对称双PWM控制环和移相控制环两个控制环路；具体控制步骤如下：A symmetrical double PWM plus phase-shift control method for controlling a current-mode unidirectional DC-DC converter, including two control loops: a symmetrical double PWM control loop and a phase-shift control loop; the specific control steps are as follows:
步骤一：通过低压侧电压vab和高压侧电压vcd之间的移相控制环的调节，使输入侧电压vab波形超前于输出侧电压vcd相角Φ；在轻载时，使变压器原边漏感电流在变压器原边电压为零时电流同时保持小值，降低环流损耗，提高系统的变换效率。所述的小值指远小于漏感电流峰值ILr。Step 1: Through the adjustment of the phase shift control loop between the low-voltage side voltagevab and the high-voltage side voltagevcd , the waveform of the input side voltagevab is ahead of the phase angle Φ of the output side voltagevcd ; at light load, the transformer The leakage inductance current of the primary side keeps a small value at the same time when the primary side voltage of the transformer is zero, which reduces the circulation loss and improves the conversion efficiency of the system. The small value mentioned is far smaller than the peak value ILr of the leakage inductance current.
输入侧电压vab和输出侧电压vcd之间的移相控制环通过电压传感器采样输出侧电压Vo的值作为移相控制环的电压反馈，计算电压给定值Vref与Vo的差值作为移相控制环电压数字PI调节器1的输入，电压数字PI调节器1的输出经限幅器限幅后作为移相控制环电流数字PI调节器2的给定；通过电流传感器采样输入侧电流iin的值作为移相控制环的电流反馈，计算电流数字PI调节器1给定值与iin的差值作为电流数字PI调节器2的输入，电流数字PI调节器2的输出经限幅器限幅后作为输出侧开关管载波与基准载波的移相角Φ；基准载波Vtr1经载波移相控制器移相180°得到输入侧滞后桥臂载波Vtr2，基准载波Vtr1经载波移相控制器相移角Φ叠加后得到相位为Φ的载波Vtr3，将Vtr3移相180°后得到输出侧滞后桥臂载波Vtr4。The phase shift control loop between the input side voltage vab and the output side voltage vcd samples the value of the output side voltage Vo through the voltage sensor as the voltage feedback of the phase shift control loop, and calculates the difference between the voltage reference value Vref and Vo The value is used as the input of the phase-shift control loop voltage digital PI regulator 1, and the output of the voltage digital PI regulator 1 is limited by the limiter as the given of the phase-shift control loop current digital PI regulator 2; the input is sampled through the current sensor The value of the side current iin is used as the current feedback of the phase-shift control loop, and the difference between the given value of the current digital PI regulator 1 and iin is calculated as the input of the current digital PI regulator 2, and the output of the current digital PI regulator 2 is passed through After the limiter is limited, it is used as the phase shift angle Φ between the carrier of the switching tube on the output side and the reference carrier; the reference carrier Vtr1 is shifted by 180° through the carrier phase shift controller to obtain the lagging bridge arm carrier Vtr2 on the input side, and the reference carrier Vtr1 is passed through The phase shift angle Φ of the carrier phase shift controller is superimposed to obtain the carrier Vtr3 with a phase of Φ, and the phase shift of Vtr3 by 180° is obtained to obtain the lagging bridge arm carrier Vtr4 on the output side.
步骤二：通过输入侧PWM控制环的调节，调节开关管占空比，使输出电压在不同输入电压时保持稳定，并使输入侧电压vab和输出侧电压vcd产生相同的波形，实现输入侧电压vab和输出侧电压vcd的匹配，降低漏感电流峰值，能够减小输入侧开关管的电流应力；使开关管Q1、Q2交错导通，直流电感L1、L2上电流有180°相位差，直流电感电流叠加可以降低输入电流纹波。Step 2: Through the adjustment of the PWM control loop on the input side, adjust the duty ratio of the switching tube to keep the output voltage stable at different input voltages, and make the input side voltage vab and the output side voltage vcd generate the same waveform to realize the input The matching of the side voltage vab and the output side voltage vcd reduces the peak value of the leakage inductance current, which can reduce the current stress of the input side switch tube; make the switch tubes Q1 and Q2 conduct alternately, and the DC inductance L1 and L2 The current has a 180° phase difference, and the superposition of the DC inductor current can reduce the input current ripple.
通过电压传感器采样输入侧箝位电容Cd1和Cd2上的电压Vc(即变压器输入侧电压vab)，与经过计算出来的电压给定Vref/n，对Vref/n进行求差后作为PWM控制环的数字PI调节器3的输入，将PWM控制环数字PI调节器3的输出经限幅器限幅后得到调节器输出Vm，将Vm与载波载波Vtr1、Vtr2、Vtr3和Vtr4比较，分别得到开关管Q1、Q2、S3和S4的PWM驱动信号，将开关管Q1和Q2的驱动信号反向后分别作为开关管Q1a和Q2a的PWM驱动信号；因此开关管Q1、Q2、S3和S4有相同占空比。所述的n＝N1：N2。Sampling the voltage Vc on the clamping capacitors Cd1 and Cd2 on the input side through the voltage sensor (that is, the voltage vab on the input side of the transformer), and calculating the given voltage Vref /n to calculate the difference between Vref /n Finally, as the input of the digital PI regulator 3 of the PWM control loop, the output of the digital PI regulator 3 of the PWM control loop is limited by the limiter to obtain the regulator output Vm , and the Vm and the carrier carrier Vtr1 , Vtr2 , Vtr3 and Vtr4 are compared to obtain the PWM driving signals of the switching tubes Q1 , Q2 , S3 and S4 respectively, and the driving signals of the switching tubes Q1 and Q2 are reversed as the switching tubes Q1a and Q2a PWM drive signal; therefore, the switches Q1 , Q2 , S3 and S4 have the same duty cycle. Said n=N1 :N2 .
有益效果：Beneficial effect:
1、本发明公开的隔离式电流型单向DC-DC变换器及用于对变换器进行控制的对称双PWM加移相控制方法，通过对输入侧和输出侧电压都采用PWM控制，产生相同的等效的PWM宽度，而输出侧和输入侧之间采用移相控制，以控制功率的方向和大小。其中输入侧和输出侧占空比设计成一致，电压vab和vcd波形一致，只是vab波形超前于vcd相角并Φ；在轻载时，变压器原边漏感电流在变压器原边电压为零时电流同时保持小值，降低环流损耗，提高系统的变换效率。1. The isolated current-type unidirectional DC-DC converter disclosed by the present invention and the symmetrical double PWM plus phase-shift control method for controlling the converter can generate the same voltage by using PWM control on the input side and output side voltage The equivalent PWM width, and the phase shift control is used between the output side and the input side to control the direction and magnitude of the power. The duty cycle of the input side and the output side is designed to be consistent, and the waveforms of the voltage vab and vcd are consistent, except that the vab waveform is ahead of the phase angle of vcd and Φ; at light load, the leakage inductance current of the primary side of the transformer is on the primary side of the transformer When the voltage is zero, the current keeps a small value at the same time, which reduces the circulation loss and improves the conversion efficiency of the system.
2、本发明的隔离式电流型单向DC-DC变换器的对称双PWM加移相控制方法，通过低压侧PWM控制环的调节，调节开关管占空比，使输出电压在不同输入电压时保持稳定，并使输入侧电压vab和输出侧电压vcd产生相同的波形，实现输入侧电压vab和输出侧电压vcd的匹配，降低漏感电流峰值，能够减小输入侧开关管的电流应力。2. The symmetrical double PWM plus phase-shift control method of the isolated current-type unidirectional DC-DC converter of the present invention, through the adjustment of the low-voltage side PWM control loop, adjusts the duty ratio of the switching tube, so that the output voltage is at different input voltages Maintain stability, and make the input side voltage vab and output side voltage vcd generate the same waveform, realize the matching of input side voltage vab and output side voltage vcd , reduce the peak value of leakage inductance current, and reduce the input side switching tube current stress.
3、通过本发明的隔离式电流型单向DC-DC变换器的输入侧直流电感对输入电流的作用，使输入电流的纹波减小，提高燃料电池和光伏组的使用寿命，提高最大功率跟踪的效果。3. Through the action of the DC inductance on the input side of the isolated current-type unidirectional DC-DC converter of the present invention on the input current, the ripple of the input current is reduced, the service life of the fuel cell and the photovoltaic group is improved, and the maximum power is increased Tracking effect.
附图说明Description of drawings
图1为本发明实施例隔离式单向DC-DC变换器电路结构示意图；FIG. 1 is a schematic structural diagram of an isolated unidirectional DC-DC converter circuit according to an embodiment of the present invention;
图2为本发明实施例的对称双PWM加移相控制方法框图；FIG. 2 is a block diagram of a symmetrical dual PWM plus phase-shift control method according to an embodiment of the present invention;
图3a)为本发明实例主要波形图；Fig. 3 a) is the main waveform diagram of the example of the present invention;
图3b)为本发明实例主要波形图。Fig. 3b) is the main waveform diagram of the example of the present invention.
具体实施方式detailed description
下面将结合附图和实施例对本发明加以详细说明，同时也叙述了本发明技术方案解决的技术问题及有益效果，需要指出的是，所描述的实施例仅旨在便于对本发明的理解，而对其不起任何限定作用。The present invention will be described in detail below in conjunction with accompanying drawing and embodiment, also described the technical problem and beneficial effect that the technical solution of the present invention solves simultaneously, it should be pointed out that described embodiment is only intended to facilitate the understanding of the present invention, and It has no limiting effect on it.
一种电流型单向DC-DC变换器，主要由主电路和控制电路组成。如图1所示，主电路主要由输入侧和输出侧构成，输入侧用于将输入的低压进行第一次升压，从而降低对变压器变比的要求，减小输入侧MOS管(Q1、Q2、Q1a、Q2a)的电流应力，并且减小输入的纹波；输出侧用于实现对变压器Tr交流电压vcd的整形，实现额定电压输出。所述的输入侧包括直流电感(L1、L2)和由MOS管(Q1、Q2、Q1a、Q2a)和箝位电容(Cd1、Cd2)组成的箝位电路，输出侧由半有源全桥和输出电容Co组成，半有源全桥包括MOS管(S3、S4)和二极管(Dr1、Dr2)。其连接关系是：输入侧的直流电感L1和L2的一端同时与输入电源正极相连，直流电感L1和直流电感L2的另一端分别和升压半桥的中点a和b相连，同时a点分别与MOS管Q1a的源极、Q1的漏极、电感Lr一端相连，MOS管Q1a的漏极与箝位电容Cd1一端连接，MOS管Q2a的漏极连接到箝位电容Cd2一端，MOS管Q1的源极、MOS管Q2的源极、箝位电容Cd1与箝位电容Cd2的一端连接到输入电源负极，电感Lr的另一端与变压器Tr原边一端连接，变压器Tr原边另一端连接到b点；输入侧和输出侧通过变压器Tr产生电气连接；变压器Tr副边的两端分别连接到半有源全桥的中点c和d，c点与二极管Dr1的阳极和MOS管S3的漏极相连，d点与二极管Dr2的阳极、MOS管S4的漏极相连，二极管Dr1的阴极、二极管Dr2的阴极、输出电容Co的一端都连接到输出负载的一端，MOS管S3的阴极、MOS管S4的阴极、Co的一端同时与负载的另一端相连。A current-mode unidirectional DC-DC converter is mainly composed of a main circuit and a control circuit. As shown in Figure 1, the main circuit is mainly composed of an input side and an output side. The input side is used to boost the input low voltage for the first time, thereby reducing the requirements for the transformation ratio of the transformer and reducing the input side MOS tube (Q1 , Q2 , Q1a , Q2a ) current stress, and reduce the input ripple; the output side is used to realize the shaping of the AC voltage vcd of the transformer Tr, and realize the rated voltage output. The input side includes a DC inductance (L1 , L2 ) and a clamping circuit composed of MOS tubes (Q1 , Q2 , Q1a , Q2a ) and clamping capacitors (Cd1 , Cd2 ), and the output The side is composed of a semi-active full bridge and an output capacitor Co , and the semi-active full bridge includes MOS transistors (S3 , S4 ) and diodes (Dr1 , Dr2 ). The connection relationship is: one end of the DC inductance L1 and L2 on the input side is connected to the positive pole of the input power supply at the same time, and the other ends of the DC inductance L1 and DC inductance L2 are respectively connected to the midpoints a and b of the boost half bridge, At the same time, point a is connected to the source of MOS transistorQ1a , the drain of Q1, andone end of inductorLr , the drain of MOS transistorQ1a is connected to one end of clamp capacitorCd1 , and the drain of MOS transistorQ2a is connected to One end of the clamping capacitor Cd2 , the source of the MOS transistor Q1, the source of the MOS transistorQ2 , the clamping capacitor Cd1 and one end of the clamping capacitor Cd2 are connected to the negativepole of the input power supply, and the other end of the inductor Lr is connected to the transformer One end of the primary side of Tr is connected, and the other end of the primary side of the transformer Tr is connected to point b; the input side and the output side are electrically connected through the transformer Tr; the two ends of the secondary side of the transformer Tr are respectively connected to midpoints c and d of the semi-active full bridge , point c is connected to the anode of diode Dr1 and the drain of MOS transistor S3, pointd is connected to the anode of diode Dr2 and the drain of MOS transistorS4 , the cathode of diode Dr1 , the cathode of diode Dr2 , the output One end of the capacitor Co is connected to one end of the output load, the cathodeof the MOS transistorS3 , the cathode of the MOS transistor S4, and one end of Co are connected to the other end of the load at the same time.
如图2所示，DC-DC变换器控制电路主要由控制器和驱动电路构成；控制器是以DSP控制器为核心，用于对由传感器采样得到的电压电流采样信号进行转换，并依据对称双PWM加移相控制方法产生PWM驱动信号，调节占空比D及输入侧和输出侧电压之间的移相角Φ，能够减小功率不传递的阶段漏感电流有效值，降低电流造成的电路通态损耗和环流损耗，同时降低漏感电流iLr峰值，减小MOS管(Q1、Q2、Q1a、Q2a)的电流应力。如图3a)所示，当移相角Φ<2π(D-0.5)时，利用变压器漏感中偏置电流实现输出侧MOS管的零电压开关；如图3b)所示，当移相角Φ>2π(D-0.5)时,漏感电流足以保证输出侧MOS管(S3、S4)的零电压开关，从而保证MOS管(S3、S4)能够在全负载范围内实现软开关驱动电路用于接收来自控制器的PWM信号，经过隔离和电压增强后为主电路的开关管提供驱动电压。在图3a)和3b)中可以知道在一个周期内，漏感电流iLr有两个过零换向的时刻，在这个时刻输出侧电流is同时过零，此时二极管自然关断，从而实现二极管(Dr1、Dr2)的零电流开关，进一步降低电路损耗。As shown in Figure 2, the DC-DC converter control circuit is mainly composed of a controller and a drive circuit; the controller is based on the DSP controller, which is used to convert the voltage and current sampling signals obtained by the sensor sampling, and according to the symmetrical The dual-PWM plus phase-shift control method generates a PWM drive signal, adjusts the duty cycle D and the phase-shift angle Φ between the input side and output side voltage, which can reduce the effective value of the leakage inductance current in the stage where the power is not transmitted, and reduce the current caused by The on-state loss and circulation loss of the circuit, while reducing the peak value of the leakage inductance current iLr , reduces the current stress of the MOS transistors (Q1 , Q2 , Q1a , Q2a ). As shown in Figure 3a), when the phase shift angle Φ<2π(D-0.5), the bias current in the leakage inductance of the transformer is used to realize the zero-voltage switching of the MOS tube on the output side; as shown in Figure 3b), when the phase shift angle When Φ>2π(D-0.5), the leakage inductance current is sufficient to ensure the zero-voltage switching of the MOS transistors (S3 , S4 ) on the output side, thereby ensuring that the MOS transistors (S3 , S4 ) can realize soft switching in the full load range. The switch drive circuit is used to receive the PWM signal from the controller, and provide the drive voltage for the switch tube of the main circuit after isolation and voltage enhancement. In Fig. 3a) and 3b), it can be known that in one cycle, the leakage inductance current iLr has two moments of zero-crossing commutation, at which moment the output side current is crosses zero at the same time, and the diode is naturally turned off at this time, thus The zero-current switching of the diodes (Dr1 , Dr2 ) is realized, and the circuit loss is further reduced.
作为优选，所述MOS管Q1、Q2、Q1a、Q2a、S3、S4为存在反并联的体二极管和漏源极的寄生电容的MOS管。Preferably, the MOS transistors Q1 , Q2 , Q1a , Q2a , S3 , and S4 are MOS transistors with antiparallel body diodes and drain-source parasitic capacitances.
如图1所示，iin为输入侧电流，iL1、iL2分别为输入电感L1、L2的电流，iLr为漏感电流，变换器输出电压为Vo，变压器原边电压为a、b两点之间的电压vab，变压器副边电压为c、d两点之间的电压vcd。S3、S4、Q1、Q1a、Q2、Q2a分别代表对应MOS管的门极信号。用于对一种电流型单向DC-DC变换器控制的对称双PWM加移相控制方法，包括对称双PWM控制环和移相控制环两个控制环路，其框图如图2所示；具体控制步骤如下：As shown in Figure 1, iin is the input side current, iL1 and iL2 are the currents of the input inductors L1 and L2 respectively, iLr is the leakage inductance current, the output voltage of the converter is Vo , and the primary side voltage of the transformer is The voltage vab between two points a and b, the secondary voltage of the transformer is the voltage vcd between two points c and d. S3 , S4 , Q1 , Q1a , Q2 , and Q2a respectively represent the gate signals of the corresponding MOS transistors. A symmetrical dual PWM plus phase-shift control method for controlling a current-mode unidirectional DC-DC converter, including two control loops, a symmetrical dual PWM control loop and a phase-shift control loop, the block diagram of which is shown in Figure 2; The specific control steps are as follows:
步骤一：通过输入侧电压vab和输出侧电压vcd之间的移相控制环的调节，使输入侧电压vab波形超前于输出侧电压vcd相角Φ；在轻载时，使变压器原边漏感电流iLr在变压器原边电压vab为零时同时保持小值，能够降低环流损耗，提高系统的变换效率。所述的小值指远小于漏感电流峰值ILr。Step 1: Through the adjustment of the phase shift control loop between the input side voltage vab and the output side voltage vcd , make the waveform of the input side voltage vab lead the phase angle Φ of the output side voltage vcd ; at light load, make the transformer The leakage inductance current iLr of the primary side maintains a small value when the primary side voltage vab of the transformer is zero, which can reduce the circulation loss and improve the conversion efficiency of the system. The small value mentioned is far smaller than the peak value ILr of the leakage inductance current.
输入侧电压vab和输出侧电压vcd之间的移相控制环通过电压传感器采样输出侧电压Vo的值作为移相控制环的电压反馈，计算电压给定值Vref与Vo的差值作为移相控制环电压数字PI调节器1的输入，电压数字PI调节器1的输出经限幅器限幅后作为移相控制环电流数字PI调节器2的给定；通过电流传感器采样输入侧电流iin的值作为移相控制环的电流反馈，计算电流数字PI调节器1给定值与iin的差值作为电流数字PI调节器2的输入，电流数字PI调节器2的输出经限幅器限幅后作为输出侧MOS管载波与基准载波的移相角Φ；基准载波Vtr1经载波移相控制器移相180°得到输入侧滞后桥臂载波Vtr2，基准载波Vtr1经载波移相控制器相移角Φ叠加后得到相位为Φ的载波Vtr3，将Vtr3移相180°后得到输出侧滞后桥臂载波Vtr4。The phase shift control loop between the input side voltage vab and the output side voltage vcd samples the value of the output side voltage Vo through the voltage sensor as the voltage feedback of the phase shift control loop, and calculates the difference between the voltage reference value Vref and Vo The value is used as the input of the phase-shift control loop voltage digital PI regulator 1, and the output of the voltage digital PI regulator 1 is limited by the limiter as the given of the phase-shift control loop current digital PI regulator 2; the input is sampled through the current sensor The value of the side current iin is used as the current feedback of the phase-shift control loop, and the difference between the given value of the current digital PI regulator 1 and iin is calculated as the input of the current digital PI regulator 2, and the output of the current digital PI regulator 2 is passed through After the limiter is limited, it is used as the phase shift angle Φ between the output side MOS tube carrier and the reference carrier; the reference carrier Vtr1 is shifted by 180° through the carrier phase shift controller to obtain the lagging bridge arm carrier Vtr2 on the input side, and the reference carrier Vtr1 is passed through The phase shift angle Φ of the carrier phase shift controller is superimposed to obtain the carrier Vtr3 with a phase of Φ, and the phase shift of Vtr3 by 180° is obtained to obtain the lagging bridge arm carrier Vtr4 on the output side.
步骤二：通过输出侧PWM控制环的调节，调节开关管占空比，使输出电压在不同输入电压时保持稳定，并使输入侧电压vab和输出侧电压vcd产生相同的波形，实现输入侧电压vab和输出侧电压vcd的匹配，降低漏感电流峰值，能够减小输入侧MOS管的电流应力；使开关管Q1、Q2交错导通，直流电感L1、L2上电流有180°相位差，直流电感电流叠加可以降低输入电流纹波。Step 2: Adjust the duty cycle of the switching tube through the adjustment of the PWM control loop on the output side, so that the output voltage remains stable at different input voltages, and the input side voltage vab and the output side voltage vcd generate the same waveform to realize the input The matching of the side voltage vab and the output side voltage vcd reduces the peak value of the leakage inductance current, which can reduce the current stress of the MOS tube on the input side; make the switch tubes Q1 and Q2 conduct alternately, and the DC inductance L1 and L2 The current has a 180° phase difference, and the superposition of the DC inductor current can reduce the input current ripple.
通过电压传感器采样输入侧钳位电容Cd1和Cd2上的电压Vc(即变压器输入侧电压vab)，与经过计算出来的电压给定Vref/n，对Vref/n进行求差后作为PWM控制环的数字PI调节器3的输入，将PWM控制环数字PI调节器3的输出经限幅器限幅后得到调节器输出Vm，将Vm与载波载波Vtr1、Vtr2、Vtr3和Vtr4比较，分别得到MOS管Q1、Q2、S3和S4的PWM驱动信号，将Q1和Q2的驱动信号反向后分别作为MOS管Q1a和Q2a的PWM驱动信号；因此MOS管Q1、Q2、S3和S4有相同占空比。所述的n＝N1：N2。Sampling the voltage Vc on the clamping capacitors Cd1 and Cd2 on the input side through the voltage sensor (that is, the voltage vab on the input side of the transformer), and calculating the given voltage Vref /n to calculate the difference between Vref /n Finally, as the input of the digital PI regulator 3 of the PWM control loop, the output of the digital PI regulator 3 of the PWM control loop is limited by the limiter to obtain the regulator output Vm , and the Vm and the carrier carrier Vtr1 , Vtr2 , Vtr3 and Vtr4 are compared to obtain the PWM driving signals of MOS transistors Q1 , Q2 , S3 and S4 respectively, and the driving signals of Q1 and Q2 are reversed as the MOS transistors Q1a and Q2a respectively PWM drive signal; therefore, MOS transistors Q1 , Q2 , S3 and S4 have the same duty cycle. Said n=N1 :N2 .
本实施例及其电路拓扑工作过程如下：The present embodiment and its circuit topology work process are as follows:
变换器上电开始工作后，对于移相控制环，数字运算控制器通过传感器采样输入侧电流iin和输出电压Vo作为反馈。Vref为输出电压给定，计算电压给定值Vref与Vo的差值，该差值作为移相控制环电压数字PI调节器1的输入，电压数字PI调节器1的输出经限幅器限幅后作为移相控制环电流数字PI调节器2的给定；通过电流传感器采样低压侧电流iin的值作为移相控制环的电流反馈，计算电流数字PI调节器1给定值与iin的差值作为电流数字PI调节器2的输入，电流数字PI调节器2的输出经限幅器限幅后作为高压侧开关管载波与基准载波的移相角Φ；基准载波Vtr1经载波移相控制器移相180°得到输入侧滞后桥臂载波Vtr2，基准载波Vtr1经载波移相控制器移相与移相角Φ叠加后得到相位为Φ的载波Vtr3，将Vtr3移相180°后得到输出侧滞后桥臂载波Vtr4。After the converter is powered on and starts to work, for the phase-shift control loop, the digital operation controller samples the input side current iin and the output voltage Vo through the sensor as feedback. Vref is the given output voltage, calculate the difference between the voltage given value Vref and Vo , the difference is used as the input of phase-shift control loop voltage digital PI regulator 1, and the output of voltage digital PI regulator 1 is limited After being limited by the controller, it is used as the given value of the current digital PI regulator 2 of the phase-shift control loop; the value of the low-voltage side current iin is sampled by the current sensor as the current feedback of the phase-shift control loop, and the current digital PI regulator 1 set value and The difference of iin is used as the input of the current digital PI regulator 2, and the output of the current digital PI regulator 2 is limited by the limiter as the phase shift angle Φ between the high-voltage side switching tube carrier and the reference carrier; the reference carrier Vtr1 is passed through The carrier phase-shifting controller shifts the phase by 180° to obtain the lagging bridge arm carrier Vtr2 at the input side. The reference carrier Vtr1 is phase-shifted by the carrier phase-shifting controller and superimposed with the phase-shift angle Φ to obtain the carrier Vtr3 with a phase of Φ. Vtr3 After a phase shift of 180°, the carrier Vtr4 of the lagging bridge arm at the output side is obtained.
通过数字运算控制器对输入侧箝位电压给定Vref/n和箝位电压反馈Vc(即两个电容Cd1和Cd2的电压)进行比较后经过数字PI调节器3和限幅器后得到调制波Vm，将Vm与载波Vtr1、Vtr2、Vtr3和Vtr4比较，分别得到MOS管Q1、Q2、S3和S4的PWM驱动信号，将Q1和Q2的驱动信号反向后分别作为MOS管Q1a和Q2a的PWM驱动信号；因此MOS管Q1、Q2、S3和S4有相同占空比。其中，载波Vtr2滞后Vtr1相位相差180，从而实现Q1和Q2的交错开通，降低了输入侧输入电流的脉动，提高了蓄电池寿命。S3和S4滞后于Q1、Q2移相角Φ，从而保证功率的传递的方向是从输入侧到输出侧。Through the digital operation controller, the clamp voltage reference Vref /n on the input side is compared with the clamp voltage feedback Vc (that is, the voltages of the two capacitors Cd1 and Cd2 ), and then passed through the digital PI regulator 3 and the limiter After obtaining the modulation wave Vm , compare Vm with the carrier Vtr1 , Vtr2 , Vtr3 and Vtr4 to obtain the PWM driving signals of the MOS transistors Q1 , Q2 , S3 and S4 respectively, and compare Q1 and Q2 drive signals are reversed as PWM drive signals for MOS transistors Q1a and Q2a respectively; therefore MOS transistors Q1 , Q2 , S3 and S4 have the same duty cycle. Among them, the carrier Vtr2 lags behind Vtr1 with a phase difference of 180°, so that the interleaved opening of Q1 and Q2 is realized, the pulsation of the input current at the input side is reduced, and the battery life is improved. S3 and S4 lag behind Q1 and Q2 by phase shift angle Φ, so as to ensure that the direction of power transmission is from the input side to the output side.
PWM控制环的具体控制信号变化过程如下：当Vc>Vref/n时，即输入侧箝位电容电压Vc高于输入侧箝位电容电压给定Vref/n，此时为了使得变压器输入侧电压跟随其给定，调节器输出值变小，低压侧开关的占空比降低，使得箝位电容电压Vc减小，也就是使得变压器输入侧电压vab的幅值减小(变压器输入侧电压vab幅值为输入侧箝位电容电压Vc)。同理，当Vc<Vref/n时，此时变压器低压侧箝位电容电压Vc低于给定，这时调节器就会增大MOS管的占空比D，使得变压器输入侧电压vab增大，从而实现变压器原边电压与变压器副边电压相匹配。The specific control signal change process of the PWM control loop is as follows: when Vc >Vref /n, that is, the input side clamp capacitor voltage Vc is higher than the input side clamp capacitor voltage given Vref /n, at this time in order to make the transformer The voltage on the input side follows its setting, the output value of the regulator becomes smaller, and the duty cycle of the switch on the low-voltage side decreases, so that the clamp capacitor voltage Vc decreases, that is, the amplitude of the voltage vab on the input side of the transformer decreases (transformer The magnitude of the input side voltage vab is the input side clamp capacitor voltage Vc ). Similarly, when Vc < Vref /n, the clamp capacitor voltage Vc on the low-voltage side of the transformer is lower than a given value, and the regulator will increase the duty cycle D of the MOS tube, so that the voltage on the input side of the transformer Vab increases, so that the voltage on the primary side of the transformer matches the voltage on the secondary side of the transformer.
通过分析，变换器工作的过程中没有额外的环流阶段，能减小变压器漏感电流iLr峰值，从而较好的降低变换器输入侧的环流损耗，也减小输入侧MOS管的电流应力。Through analysis, there is no additional circulating current stage in the working process of the converter, which can reduce the peak value of the leakage inductance current iLr of the transformer, thereby better reducing the circulating current loss at the input side of the converter and reducing the current stress of the MOS tube at the input side.
如图3中所示，当移相角Φ<2π(D-0.5)时，在vab和vcd电压同时为零的时刻，漏感电流减小到零后，由于输出侧MOS管寄生结电容CS3和CS4存储有能量，此时结电容CS3或CS4会和变压器漏感谐振，直到结电容上电压降为零，MOS管的体二极管导通，从而实现输出侧MOS管的零电压开关。谐振结束后结电容能量转移到漏感中，所以漏感电流会保持一个小值。当移相角Φ>2π(D-0.5)时，在vab和vcd电压不存在同时为零的时刻，输出侧电流可以保证输出侧MOS管的零电压开关。从分析可知，输出侧MOS管可以实现全负载范围内的零电压开关。如图3中所见，漏感电流iLr在一个周期内存在两次下降到零时刻，这个时刻副边电流is也会同时为零，此时输出侧二极管(Dr1、Dr2)没有电流流过，自然关断，所以二极管可实现零电流关断。通过分析可知，输出侧的所有MOS管和二极管都可以实现软开关，从而降低了损耗，提高了变换器的效率和可靠性。As shown in Figure 3, when the phase shift angle Φ<2π(D-0.5), at the moment when the voltages of vab and vcd are zero at the same time, after the leakage inductance current decreases to zero, due to the parasitic junction of the MOS transistor on the output side Capacitors CS3 and CS4 store energy. At this time, the junction capacitance CS3 or CS4 will resonate with the leakage inductance of the transformer until the voltage drop on the junction capacitance is zero, and the body diode of the MOS transistor is turned on, thereby realizing the output side of the MOS transistor. Zero voltage switching. After the resonance is over, the junction capacitance energy is transferred to the leakage inductance, so the leakage inductance current will keep a small value. When the phase shift angle Φ>2π(D-0.5), the current at the output side can ensure the zero-voltage switching of the MOS transistor at the output side at the moment when the voltages of vab and vcd are zero at the same time. It can be seen from the analysis that the MOS tube on the output side can realize zero-voltage switching within the full load range. As shown in Figure 3, the leakage inductance current iLr drops to zero twice in one cycle, and the secondary current is will also be zero at the same time. At this time, the output side diodes (Dr1 , Dr2 ) have no The current flows and is naturally turned off, so the diode can be turned off with zero current. It can be seen through analysis that all MOS transistors and diodes on the output side can realize soft switching, thereby reducing losses and improving the efficiency and reliability of the converter.
综上，通过本发明提出的对称双PWM加移相控制方法，可以很好对提出的DC-DC变换器上实现对称双PWM加移相的控制，能够减小输入侧MOS管电流应力和变换器的环流损耗，实现输出侧半有源全桥MOS的ZVS软开关和二极管的ZCS开关，提高变换器的效率和可靠性。To sum up, through the symmetrical dual PWM plus phase shift control method proposed by the present invention, the proposed DC-DC converter can be well implemented with symmetrical dual PWM plus phase shift control, which can reduce the current stress and transformation of the MOS tube on the input side. The circulating current loss of the converter realizes the ZVS soft switching of the semi-active full-bridge MOS on the output side and the ZCS switch of the diode, which improves the efficiency and reliability of the converter.
以上所述的具体描述，对发明的目的、技术方案和有益效果进行进一步详细说明，所应理解的是，以上所述仅为本发明的具体实施例而已，并不用于限定本发明的保护范围，凡在本发明的精神和原则之内，所做的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。The specific description above is to further describe the purpose, technical solution and beneficial effect of the invention in detail. It should be understood that the above description is only a specific embodiment of the present invention and is not used to limit the protection scope of the present invention. , Any modifications, equivalent replacements, improvements, etc. made within the spirit and principles of the present invention shall be included within the protection scope of the present invention.
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|---|---|
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|---|---|
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN106787755A (en)* | 2016-12-29 | 2017-05-31 | 北京理工大学 | The optimal feed forward control method of the active doube bridge DC DC converters of current source type |
| CN106712522B (en)* | 2017-01-20 | 2019-02-15 | 北京理工大学 | PWM-Phase-shifting Composite Control Method of Semi-active Bridge DC-DC Converter |
| WO2018216294A1 (en)* | 2017-05-25 | 2018-11-29 | シャープ株式会社 | Dc/dc converter |
| DE102017212568A1 (en)* | 2017-07-21 | 2019-01-24 | Robert Bosch Gmbh | Electric machine |
| CN109004841B (en)* | 2018-07-20 | 2023-05-02 | 福建星云电子股份有限公司 | AC-DC-DC converter and double-loop feedforward secondary ripple suppression method thereof |
| CN110011528B (en)* | 2019-04-18 | 2020-07-24 | 珠海格力电器股份有限公司 | Bridge circuit soft start method, controller and equipment |
| CN110729912B (en)* | 2019-10-30 | 2021-05-14 | 渤海大学 | High frequency induction heating series resonance soft switching inverter control method |
| CN111669057B (en)* | 2020-05-26 | 2021-05-18 | 中国科学院电工研究所 | DC boost converter and its control method |
| CN113285598B (en)* | 2021-05-19 | 2022-11-25 | 哈尔滨工业大学 | Hybrid Control Structure and Method for Interleaved Parallel Boost Converter |
| TWI765740B (en)* | 2021-06-09 | 2022-05-21 | 崑山科技大學 | Symmetrical switching type high boost dc converter |
| CN113507212B (en)* | 2021-07-16 | 2022-07-22 | 中南大学 | Dual active bridge DC-DC converter and power control method based on three-winding coupled inductor |
| CN113890376B (en)* | 2021-10-14 | 2023-08-22 | 东南大学 | Isolation type DC-DC converter topology with wide input voltage and control method thereof |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN103872920A (en)* | 2014-03-13 | 2014-06-18 | 北京理工大学 | Leakage inductance current slope direct control method of isolation type two-way three-level converter |
| CN104242664A (en)* | 2014-09-25 | 2014-12-24 | 西安交通大学 | Modulation method of three-phase isolation type bidirectional direct-current converter under double PWM (pulse width modulation) |
| CN104506040A (en)* | 2014-09-22 | 2015-04-08 | 北京理工大学 | Dual-PWM and phase-shifting control method with the same duty ratio |
| CN104539164A (en)* | 2014-12-31 | 2015-04-22 | 北京理工大学 | Current type bidirectional DC-DC converter unequal-width PWM and double phase-shifting control method |
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN103872920A (en)* | 2014-03-13 | 2014-06-18 | 北京理工大学 | Leakage inductance current slope direct control method of isolation type two-way three-level converter |
| CN104506040A (en)* | 2014-09-22 | 2015-04-08 | 北京理工大学 | Dual-PWM and phase-shifting control method with the same duty ratio |
| CN104242664A (en)* | 2014-09-25 | 2014-12-24 | 西安交通大学 | Modulation method of three-phase isolation type bidirectional direct-current converter under double PWM (pulse width modulation) |
| CN104539164A (en)* | 2014-12-31 | 2015-04-22 | 北京理工大学 | Current type bidirectional DC-DC converter unequal-width PWM and double phase-shifting control method |
| Title |
|---|
| A High Step-up Current-Fed Phase-Shift Double-Input DC-DC Converter Using Asymmetrical PWM Switching Strategy;M.Mohammadi,J.等;《Power Electronics Drive Systems and Technologies Conference》;20150204;第400-405页* |
| Publication number | Publication date |
|---|---|
| CN105305829A (en) | 2016-02-03 |
| Publication | Publication Date | Title |
|---|---|---|
| CN105305829B (en) | The unidirectional DC DC converters of current mode and symmetric double PWM add phase-shifting control method | |
| CN107370386B (en) | Optimal Duty Cycle Voltage Mismatch Control Method for Current Mode Bidirectional DC-DC Converter | |
| CN108900100B (en) | A kind of single-phase high efficiency high frequency isolated form rectifier | |
| CN104539164B (en) | The not wide PWM of the two-way DC DC converters of current mode adds two-track phase control method | |
| WO2022134278A1 (en) | Current source input high-frequency isolation matrix converter and regulation and control method thereof | |
| CN104506040B (en) | Dual-PWM and phase-shifting control method with the same duty ratio | |
| CN112928919B (en) | Isolated high-frequency resonant DC-DC converter with wide output voltage range and method | |
| CN103441680B (en) | A kind of soft switching full-bridge direct-current converter reducing circulation loss | |
| CN102364860B (en) | A Secondary-side Phase Shift Controlled Full-Bridge Converter | |
| CN109104108B (en) | A soft-switching single-stage high-frequency isolated rectifier with active clamping | |
| CN109921653B (en) | A single-phase power electronic transformer topology structure and its control method | |
| CN110855163A (en) | A single-stage isolated three-phase rectifier and its control method | |
| CN102201739A (en) | Symmetrical half-bridge LLC resonant bidirectional DC-DC converter | |
| Guo et al. | Control strategy of AC-DC converter based on dual active bridge with minimum current stress and soft switching | |
| CN114583967A (en) | Isolated direct-current boost converter of two-phase parallel boost circuit and control method thereof | |
| CN111431415B (en) | High-boost isolated DC converter with parallel input and series output | |
| CN106655785A (en) | Bidirectional hybrid bridge DC-DC converter and half-cycle volt-second area balance control method | |
| CN105207486B (en) | A kind of two-way resonance DC converter and its control method | |
| CN101604916B (en) | Zero-Voltage Switching Full-Bridge DC Converter Based on π-type Auxiliary Network | |
| CN216819713U (en) | Wide-range bidirectional resonant soft-switching direct-current converter | |
| CN110445387B (en) | A kind of topology structure and control method of component power supply | |
| CN101847936A (en) | Soft switching full-bridge direct-current converter with lag leg connected with auxiliary network in parallel | |
| CN108880268A (en) | The multi-mode control method of the semi-active bridge DC-DC converter of voltage-source type | |
| CN109742957B (en) | Double-ring full-resonance type soft switching converter | |
| CN111884521B (en) | Single-stage Boost full-bridge boost zero-current switching DC converter and its control method |
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