技术领域technical field
本发明涉及开关式电源供应器(switched mode power supply)，尤其涉及可以预估输出至一负载的一输出电流的电源供应器。The present invention relates to a switched mode power supply, in particular to a power supply capable of estimating an output current output to a load.
背景技术Background technique
开关式电源供应器一般采用一功率开关来控制流经一电感元件的一电流。跟其他一般电源供应器相较之下，开关式电源供应器具有较小的产品体积以及较优越的转换效率，所以广受业界的欢迎与采用。Switching mode power supplies generally use a power switch to control a current flowing through an inductive element. Compared with other general power supplies, switching power supplies have smaller product size and superior conversion efficiency, so they are widely welcomed and adopted by the industry.
反驰式(flyback)架构的供应器，因为其提供了电隔绝的效果，所以广泛的被采用。反驰式架构以一个变压器，把连接到市电插座的输入电源线，直流电隔绝于连接到负载的输出电源线。初级侧一般是指电连接到输入电源线的那些电路所在的那一侧；相对的，次级侧一般指的是电连接到输出电源线的那些电路所在的那一侧。所谓次级侧控制是采用电阻或一些电子元件，放在次级侧，来检测次级侧的负载上的输出电压或是流过的输出电流。次级侧控制可以简单的达到不错的输出电压或是输出电流调节(regulation)，但是因为次级侧的电阻或是电子元件不断的消耗电能，因此次级侧控制可能会有比较低的转换效率。美国专利申请公开号US20100321956A1揭露了许多开关式电源供应器，其采用初级侧控制来调节次级侧的最大额定输出电流或是额定输出电压。美国专利申请公开号US20100321956A1可以使次级侧的最大额定输出电流，大约为一个不随输入电源线的输入电压而变化的定值。The flyback type (flyback) structure of the supply, because it provides the effect of electrical isolation, so it is widely used. The flyback architecture uses a transformer to isolate the DC power of the input power line connected to the mains outlet from the output power line connected to the load. The primary side generally refers to the side where those circuits electrically connected to the input power line are located; in contrast, the secondary side generally refers to the side where those circuits electrically connected to the output power line are located. The so-called secondary side control is to use resistors or some electronic components on the secondary side to detect the output voltage or the output current flowing through the load on the secondary side. The secondary side control can easily achieve a good output voltage or output current regulation (regulation), but because the secondary side resistors or electronic components continue to consume power, the secondary side control may have relatively low conversion efficiency . US Patent Application Publication No. US20100321956A1 discloses many switching power supplies, which adopt primary-side control to adjust the maximum rated output current or rated output voltage of the secondary side. US Patent Application Publication No. US20100321956A1 can make the maximum rated output current of the secondary side approximately a fixed value that does not vary with the input voltage of the input power line.
负载补偿(load compensation)是一种在对负载的输出电流增加时，就增加输出电压的一种技术，可以用来补偿在电源供应器与负载之间传输线上的电压损失。一种做到负载补偿的传统方法，是采用流经一变压器的一峰值电流，来大约当作输出至一负载的输出电流的代表，据以改变电源供应器所调节的输出电压的目标值。但是，如同业界所知道的，峰值电流跟输出电流是有相当大的差异的，所以，峰值电流根本不能用来代表输出电流。Load compensation is a technique that increases the output voltage when the output current to the load increases, and can be used to compensate for the voltage loss on the transmission line between the power supply and the load. One conventional approach to load compensation is to use a peak current flowing through a transformer as an approximate representation of the output current to a load to vary the target value of the output voltage regulated by the power supply. However, as known in the industry, there is a considerable difference between the peak current and the output current, so the peak current cannot be used to represent the output current at all.
发明内容Contents of the invention
实施例揭示有一种电源控制器，适用于一开关式电源供应器。该开关式电源供应器包含有串联在一起的一电感元件以及一功率开关。该电源控制器包含有一输出电流估计器以及一电流限制器。该输出电流估计器，架构来提供一电流检测信号以及一放电时间信号。该电流检测信号代表流经该电感元件的一电感电流。该放电时间信号指示该电感元件的一放电时间。并依据该电流检测信号以及该放电时间信号，该输出电流估计器产生一充电电流，使其大约对应该开关式电源供应器对一负载输出的一输出电流。该充电电流被限制不大于一最大值。当该充电电流等于该最大值时，该电流限制器架构来用以限制该电流检测信号。The embodiment discloses a power controller suitable for a switching power supply. The switching power supply includes an inductance element and a power switch connected in series. The power controller includes an output current estimator and a current limiter. The output current estimator is configured to provide a current detection signal and a discharge time signal. The current detection signal represents an inductor current flowing through the inductor element. The discharge time signal indicates a discharge time of the inductance element. And according to the current detection signal and the discharge time signal, the output current estimator generates a charging current to approximately correspond to an output current output by the switching power supply to a load. The charging current is limited not to be greater than a maximum value. When the charging current is equal to the maximum value, the current limiter structure is used to limit the current detection signal.
实施例揭示有一种控制方法，适用于一开关式电源供应器中，作为输出电流检测。该开关式电源供应器包含有一电感元件以及一功率开关，串接在一起。该控制方法包含有：接收一电流检测信号，其大致代表流经该电感元件的一电感电流；检测该电感元件，以产生一放电时间信号，大致指示该电感元件的一放电时间；依据该电流检测信号以及该放电时间信号，产生一充电电流，其中，该放电电流大约代表了该开关式电源供应器输出至一负载的一输出电流；限制使该充电电流不超过一最大值；以及，当该充电电流等于该最大值时，压制该电流检测信号。The embodiment discloses a control method suitable for a switch mode power supply as output current detection. The switching power supply includes an inductance element and a power switch connected in series. The control method includes: receiving a current detection signal, which roughly represents an inductive current flowing through the inductive element; detecting the inductive element to generate a discharge time signal, roughly indicating a discharge time of the inductive element; according to the current detection signal and the discharge time signal, generating a charging current, wherein the discharging current approximately represents an output current output from the switching power supply to a load; limiting the charging current so that it does not exceed a maximum value; and, when When the charging current is equal to the maximum value, the current detection signal is suppressed.
实施例另揭示有一种电源控制器，适用于一开关式电源供应器。该开关式电源供应器包含有串联在一起的一电感元件以及一功率开关。该电源控制器包含有一输出电流估计器以及一负载补偿器。该输出电流估计器架构来接收一电流检测信号以及一放电时间信号。该电流检测信号大致代表流经该电感元件的一电感电流，该放电时间信号大约指示该电感元件的一放电时间。该输出电流估计器依据该电流检测信号以及该放电时间信号，产生一充电电流。该充电电流大致对应该开关式电源供应器对一负载输出的一输出电流。该负载补偿器，据该充电电流，架构来从一电阻汲取一偏压电流，流到一接地线。该电感元件包含有一辅助绕组，且该电阻连接在该辅助绕组以及该负载补偿器之间。The embodiment further discloses a power controller suitable for a switching power supply. The switching power supply includes an inductance element and a power switch connected in series. The power controller includes an output current estimator and a load compensator. The output current estimator is configured to receive a current detection signal and a discharge time signal. The current detection signal roughly represents an inductor current flowing through the inductor element, and the discharge time signal roughly indicates a discharge time of the inductor element. The output current estimator generates a charging current according to the current detection signal and the discharge time signal. The charging current roughly corresponds to an output current output by the switching power supply to a load. The load compensator is configured to draw a bias current from a resistor to a ground line according to the charging current. The inductance element includes an auxiliary winding, and the resistor is connected between the auxiliary winding and the load compensator.
附图说明Description of drawings
图1显示依据本发明所实施的一开关式电源供应器。FIG. 1 shows a switching power supply implemented according to the present invention.
图2显示图1中的一些信号的波形。Figure 2 shows the waveforms of some of the signals in Figure 1.
图3举例图1中的电源控制器。FIG. 3 exemplifies the power controller in FIG. 1 .
图4举例图3中的输出电流估计器。FIG. 4 illustrates the output current estimator in FIG. 3 .
图5A显示在一些实施例中，充电电流ICHARGE跟电压VM的关系。FIG. 5A shows the relationship between the charging current ICHARGE and the voltage VM in some embodiments.
图5B显示在一些实施例中，偏压电流IOFFSET跟充电电流ICHARGE的关系。FIG. 5B shows the relationship between the bias current IOFFSET and the charging current ICHARGE in some embodiments.
【符号说明】【Symbol Description】
具体实施方式Detailed ways
本发明的一实施例具有设置于初级侧的一电源控制器，其可以依据流经一变压器之初级侧绕组的一电流，以及该变压器的一放电时间，来产生估计信号。这些估计信号的其中之一是一充电电流，其产生的方法将在此说明书中解释，并证明此充电电流相当程度地可以代表一电源供应器对一负载所输出的一输出电流。而且，通过限制该充电电流的最大值，该输出电流也可以被准确地调节的不超过该电源供应器的最大额定输出电流。此外，此充电电流非常精准地或是相当程度地可以代表该输出电流，所以该充电电流可以作为一个输入，来产生要作为负载补偿的一偏压电流(offset current)，而得到一个精准控制的结果。An embodiment of the present invention has a power controller disposed on the primary side, which can generate an estimation signal according to a current flowing through a primary side winding of a transformer and a discharge time of the transformer. One of these estimated signals is a charging current, the method of which is generated will be explained in this specification, and it is proved that the charging current can represent an output current output by a power supply to a load to a considerable extent. Moreover, by limiting the maximum value of the charging current, the output current can also be accurately regulated so as not to exceed the maximum rated output current of the power supply. In addition, the charging current can represent the output current very accurately or to a considerable extent, so the charging current can be used as an input to generate a bias current (offset current) to be used as load compensation to obtain a precisely controlled result.
图1显示依据本发明所实施的一开关式电源供应器10，其采用初级侧控制。桥式整流器20提供全波式整流，将市电插座供应来的交流电(AC)电源，转换成直流电(DC)输入电源VIN。输入电源VIN的电压，可能具有M型的波形，或是大致为一个不变常数。电源控制器26可以是一集成电路，其具有数个引脚，可以电连接到周边装置或元件。通过驱动端GATE，电源控制器26可以周期地开启或是关闭功率开关34。当功率开关34开启时，该变压器的初级侧绕组PRM储能；当功率开关34关闭时，该变压器通过次级侧绕组SEC与辅助绕组AUX释能，分别建立出输出电源VOUT以及操作电源VCC，分别供应给负载24与电源控制器26。连接在负载24与二次绕组之间有传输线38，其寄生电阻以标示于其中的电阻表示。FIG. 1 shows a switching mode power supply 10 implemented according to the present invention, which adopts primary side control. The bridge rectifier 20 provides full-wave rectification to convert the alternating current (AC) power supplied by the mains socket into a direct current (DC) input power VIN . The voltage of the input power supply VIN may have an M-shaped waveform, or be roughly a constant. The power controller 26 can be an integrated circuit with several pins that can be electrically connected to peripheral devices or components. Through the driving terminal GATE, the power controller 26 can periodically turn on or turn off the power switch 34 . When the power switch 34 is turned on, the primary side winding PRM of the transformer stores energy; when the power switch 34 is turned off, the transformer releases energy through the secondary side winding SEC and the auxiliary winding AUX, respectively establishing the output power supply VOUT and the operating power supply VCC is supplied to the load 24 and the power controller 26 respectively. Connected between the load 24 and the secondary winding is a transmission line 38 whose parasitic resistance is represented by the resistance marked therein.
电阻28与30可以视为一个分压器，用来检测辅助绕组AUX上的跨压VAUX，然后提供一反馈电压信号VFB，到电源控制器26的一反馈端FB。当功率开关34关闭时，跨压VAUX大约会是比例于次级侧绕组SEC上的跨压的一映射电压(reflective voltage)。依据反馈电压信号VFB，电源控制器26控制功率开关34的工作周期(duty cycle)。通过电流检测端CS，电源控制器26检测电流检测电压VCS，其代表了流经电流检测电阻36、功率开关34以及初级侧绕组PRM的电流IPRM。Resistors 28 and 30 can be regarded as a voltage divider for detecting the voltage across the auxiliary winding AUX VAUX , and then providing a feedback voltage signal VFB to a feedback terminal FB of the power controller 26 . When the power switch 34 is turned off, the voltage across VAUX is approximately a reflective voltage proportional to the voltage across the secondary winding SEC. According to the feedback voltage signal VFB , the power controller 26 controls the duty cycle of the power switch 34 . Through the current sensing terminal CS, the power controller 26 detects the current sensing voltage VCS , which represents the current IPRM flowing through the current sensing resistor 36 , the power switch 34 and the primary winding PRM.
图2显示图1中的一些信号的波形。在驱动端GATE上的驱动信号VGATE为逻辑上的1时，功率开关34开启。功率开关34维持在开启的时间，称为开启时间(ON time)TON；相对的，功率开关34维持在关闭的时间，称为关闭时间(OFF time)TOFF，如同图2所示。一个开关周期(cycle time)TCYC等于一个开启时间TON与一个关闭时间TOFF的总和，如同图2所显示的。在开启时间TON时，辅助绕组AUX上的跨压VAUX是负值，映射输入电源VIN的电压。此时，初级侧绕组PRM储能，代表电流IPRM的电流检测电压VCS将随着时间而增加。在功率开关34被关闭的瞬间，次级侧绕组SEC开始产生次级侧绕组电流ISEC，其随着时间而减少，直到次级侧绕组SEC完全的释能。如同图2所示，次级侧绕组SEC释能的时间，或是次级侧绕组电流ISEC大于0的时间，称为放电时间TDIS。在放电时间TDIS时，跨压VAUX为正值，大约映射输出电源VOUT的电压。在放电时间TDIS结束后，跨压VAUX开始振荡，往0V收敛。在图2中，放电时间TDIS仅仅是关闭时间TOFF中的一部分，因为开关式电源供应器10是假定操作在不连续导通模式(discontinuous conduction mode，DCM)。万一开关式电源供应器10是操作于连续导通模式(continuousconduction mode，CCM)，放电时间TDIS大约会等于关闭时间TOFF。Figure 2 shows the waveforms of some of the signals in Figure 1. When the driving signal VGATE on the driving terminal GATE is logic 1, the power switch 34 is turned on. The time when the power switch 34 is turned on is referred to as ON time TON ; on the contrary, the time when the power switch 34 is turned off is referred to as OFF time TOFF , as shown in FIG. 2 . A switching cycle (cycle time) TCYC is equal to the sum of an on time TON and an off time TOFF , as shown in FIG. 2 . During the turn-on time TON , the voltage VAUX across the auxiliary winding AUX is negative, reflecting the voltage of the input power supply VIN . At this time, the primary side winding PRM stores energy, and the current detection voltage VCS representing the current IPRM will increase with time. At the moment when the power switch 34 is turned off, the secondary winding SEC starts to generate a secondary winding current ISEC , which decreases with time until the secondary winding SEC is fully discharged. As shown in FIG. 2 , the time when the secondary side winding SEC releases energy, or the time when the secondary side winding current ISEC is greater than 0, is called the discharge time TDIS . During the discharge time TDIS , the voltage across VAUX is positive, approximately mirroring the voltage of the output power supply VOUT . After the discharge time TDIS ends, the cross voltage VAUX starts to oscillate and converges to 0V. In FIG. 2 , the discharge time TDIS is only a part of the off time TOFF because the switch mode power supply 10 is supposed to operate in a discontinuous conduction mode (DCM). If the switch mode power supply 10 is operated in continuous conduction mode (CCM), the discharge time TDIS is approximately equal to the off time TOFF .
图3举例图1中的电源控制器26。振荡器72周期地设置SR寄存器(SRregister)78，因此驱动信号VGATE变成逻辑上的1，所以开始了开启时间TON。采样保持电路62适时地在放电时间TDIS时，对反馈电压信号VFB进行采样，然后误差放大器68比较采样保持电路62的采样结果，跟一预设的目标电压VTAR，来产生补偿电压VCOM。当电流检测电压VCS超过了补偿电压VCOM时，比较器74重置SR寄存器78，开始了关闭时间TOFF。在关闭时间TOFF内，电流检测电压VCS突然地降到0V，如同图2所示。换句话说，补偿电压VCOM控制或是压制了电流检测电压VCS的峰值。FIG. 3 illustrates the power controller 26 in FIG. 1 . Oscillator 72 periodically sets SR register (SRregister) 78, so that drive signal VGATE becomes logic 1, so on time TON begins. The sample-and-hold circuit 62 samples the feedback voltage signal VFB in a timely manner during the discharge time TDIS , and then the error amplifier 68 compares the sampling result of the sample-and-hold circuit 62 with a preset target voltage VTAR to generate a compensation voltage VCOM . When the current sense voltage VCS exceeds the compensation voltage VCOM , the comparator 74 resets the SR register 78 and the off-time TOFF begins. During the off time TOFF , the current detection voltage VCS suddenly drops to 0V, as shown in FIG. 2 . In other words, the compensation voltage VCOM controls or suppresses the peak value of the current sensing voltage VCS .
放电时间决定器64电连接到反馈端FB，通过检测反馈电压信号VFB的波形，放电时间决定器64提供一放电时间信号SDIS，来指示放电时间TDIS的时间。放电时间信号SDIS不必要一定跟放电时间TDIS完全的同步。举例来说，在一实施例中，放电时间信号SDIS在放电时间TDIS开始一段时间之后，才变成逻辑上的1，然后在放电时间TDIS结束一段时间后，才变成逻辑上的0，因此，放电时间信号SDIS为逻辑上1的时间，大约等于放电时间TDIS。The discharge time determiner 64 is electrically connected to the feedback terminal FB. By detecting the waveform of the feedback voltage signal VFB , the discharge time determiner 64 provides a discharge time signal SDIS to indicate the discharge time TDIS . The discharge time signal SDIS does not have to be completely synchronized with the discharge time TDIS . For example, in one embodiment, the discharge time signal SDIS becomes logically 1 after the discharge time TDIS starts for a period of time, and then becomes logically 1 after the discharge time TDIS ends for a period of time. 0, therefore, the time during which the discharge time signal SDIS is logically 1 is approximately equal to the discharge time TDIS .
依据放电时间信号SDIS以及电流检测电压VCS，输出电流估计器70提供负载代表电压VLC给负载补偿电路66。这里，负载代表电压VLC对应到一充电电流ICHARGE。稍后将说明，充电电流ICHARGE大致上比例于图1中输出到负载24的输出电流IOUT。负载补偿电路66产生偏压电流IOFFSET，其在放电时间，从反馈端FB汲取电流，流到接地线。一般而言，输出电流IOUT越大，充电电流ICHARGE越大，偏压电流IOFFSET越大，跨压VAUX就需要越大来维持采样保持电路62的采样结果大约等于目标电压VTAR，所以输出电源VOUT的电压越高。因此，负载补偿电路66可以使得输出电源VOUT的电压大致等于“IOUT*K1+K2*VTAR”，其中K1与K2是两个定值。只要适当地选取电阻28与30的电阻值，IOUT*K1就可以大约等于图1中的传输线38上的跨压损耗，因此负载24可以收到调节结果相当不错的电源电压(=K2*VTAR)。负载补偿就此准确的实现。According to the discharge time signal SDIS and the current detection voltage VCS , the output current estimator 70 provides a load representative voltage VLC to the load compensation circuit 66 . Here, the load representative voltage VLC corresponds to a charging current ICHARGE . It will be explained later that the charging current ICHARGE is roughly proportional to the output current IOUT output to the load 24 in FIG. 1 . The load compensation circuit 66 generates a bias current IOFFSET , which draws current from the feedback terminal FB and flows to the ground line during the discharge time. Generally speaking, the larger the output current IOUT , the larger the charging current ICHARGE , the larger the bias current IOFFSET , the larger the cross voltage VAUX is required to maintain the sampling result of the sample and hold circuit 62 approximately equal to the target voltage VTAR , So the voltage of the output power supply VOUT is higher. Therefore, the load compensation circuit 66 can make the voltage of the output power VOUT roughly equal to “IOUT *K1 +K2 *VTAR ”, where K1 and K2 are two constant values. As long as the resistance values of the resistors 28 and 30 are properly selected, IOUT *K1 can be approximately equal to the voltage loss on the transmission line 38 in FIG. *VTAR ). Load compensation is thus accurately realized.
输出电流估计器70还提供了限制电压VLIMIT给比较器76。一但电流检测信号VCS超过了限制电压VLIMIT，比较器76重置SR寄存器78，结束开启时间TON，并开始了关闭时间TOFF。因此，限制电压VLIMIT也可以控制或压制电流检测信号VCS的峰值。Output current estimator 70 also provides a limit voltage VLIMIT to comparator 76 . Once the current detection signal VCS exceeds the limit voltage VLIMIT , the comparator 76 resets the SR register 78 to end the on-time TON and start the off-time TOFF . Therefore, the limit voltage VLIMIT can also control or suppress the peak value of the current sense signal VCS .
图4举例输出电流估计器70，其具有转导器90、电位转换器(levelshifter)92与94、一更新电路96、一收集电容98、一开关104、一电压控制电流源(voltage-controlled current source)102、以及一CS峰值电压检测器100。Fig. 4 illustrates output current estimator 70, and it has transducer 90, potential converter (levelshifter) 92 and 94, an update circuit 96, a collecting capacitor 98, a switch 104, a voltage-controlled current source (voltage-controlled current). source) 102, and a CS peak voltage detector 100.
CS峰值电压检测器100产生电压VCS-PEAK，其代表了电流检测信号VCS的一峰值。举例来说，美国专利申请公开号US20100321956A1中的图10就提供了CS峰值电压检测器100的一例子。在一些实施例中，CS峰值电压检测器100可以用美国专利申请公开号US20100321956A1的图17或18图中所举例的平均电流检测器所取代。电压控制电流源102将电压VCS-PEAK转换成放电电流IDIS，其仅有在放电时间信号SDIS为逻辑上的1时，对收集端ACC放电。换句话说，放电电流IDIS对收集端ACC的放电时间，等效上大约等于放电时间TDIS。在一些实施例中，图4中的开关104可以省略，取而代之，放电时间信号SDIS用来启动(activate)或是关闭(deactivate)电压控制电流源102。在电容99上的电压VM，被位移转换后，送给转导器90，用来跟一预设参考电压VREF比较。转导器90依据比较结果，来输出充电电流ICHARGE，对收集端ACC持续地充电。通过检测充电电流ICHARGE，可以产生负载代表电压VLC。更新电路96受更新信号SUPDATE所触发，对收集端ACC上的反馈电压VACC采样，来更新电压VM，可以每次开关周期来更新一次。更新信号SUPDATE并不必要每次开关周期就使得更新电路96执行更新一次，举例来说，也可以每两个开关周期执行更新一次。在一实施例中，更新信号SUPDATE可以等同于驱动信号VGATE，意味着更新的动作在关闭时间TOFF一开始时被执行。电压VM平时都是保持在一个定值，直到更新电路96对它更新后，才会变成另一个定值。依据电压VM，电位转换器94提供限制电压VLIMIT。从以上说明可以发现，当电压VM不变时，充电电流ICHARGE也会维持不变。The CS peak voltage detector 100 generates a voltage VCS-PEAK representing a peak value of the current sense signal VCS . For example, FIG. 10 in US Patent Application Publication No. US20100321956A1 provides an example of a CS peak voltage detector 100 . In some embodiments, the CS peak voltage detector 100 can be replaced by an average current detector as exemplified in FIG. 17 or 18 of US Patent Application Publication No. US20100321956A1. The voltage-controlled current source 102 converts the voltage VCS-PEAK into a discharge current IDIS , which discharges the collector ACC only when the discharge time signal SDIS is logic 1. In other words, the discharge time of the collecting terminal ACC by the discharge current IDIS is equivalent to approximately equal to the discharge time TDIS . In some embodiments, the switch 104 in FIG. 4 can be omitted, and instead, the discharge time signal SDIS is used to activate or deactivate the voltage-controlled current source 102 . The voltage VM on the capacitor 99 is shifted and then sent to the transducer 90 for comparison with a preset reference voltage VREF . According to the comparison result, the transconductor 90 outputs a charging current ICHARGE to continuously charge the collecting terminal ACC. By detecting the charging current ICHARGE , a load representative voltage VLC can be generated. The update circuit 96 is triggered by the update signal SUPDATE , and samples the feedback voltage VACC on the collecting terminal ACC to update the voltage VM , which can be updated once every switching cycle. The update signal SUPDATE does not necessarily cause the update circuit 96 to perform an update every switching cycle, for example, it may also perform an update every two switching cycles. In an embodiment, the update signal SUPDATE may be equal to the drive signal VGATE , which means that the update action is performed at the beginning of the off-time TOFF . The voltage VM is usually kept at a constant value, and will not change to another constant value until the updating circuit 96 updates it. According to the voltage VM , the level shifter 94 provides the limit voltage VLIMIT . It can be found from the above description that when the voltage VM remains unchanged, the charging current ICHARGE will also remain unchanged.
类似美国专利申请公开号US20100321956A1中所分析的，当充电电流ICHARGE为一个定值，且反馈电压VACC在被采样时的值，等于上一次被采样时的值，那充电电流ICHARGE就会是跟输出到负载24的输出电流IOUT成比例。为了使充电电流ICHARGE跟输出电流IOUT成比例，所以反馈电压VACC每次被采样时的值，必须要一样或是稳定。更新电路96、电位转换器92、以及转导器90一起形成了具有负回路增益(negative loop gain)的一回路，而这个回路最后可能可以使得反馈电压VACC每次被采样时的值，稳定在一个值。举例来说，如果充电电流ICHARGE大于跟输出电流IOUT成比例的一期望值，那反馈电压VACC在下次的采样时，就会变大，造成更新后的电压VM也随着变大，因此，充电电流ICHARGE就会变小。反之亦然。所以，充电电流ICHARGE可以在最后，变的大约跟输出电流IOUT成比例。Similar to the analysis in US Patent Application Publication No. US20100321956A1, when the charging current ICHARGE is a fixed value, and the value of the feedback voltage VACC when being sampled is equal to the value when it was sampled last time, then the charging current ICHARGE will be is proportional to the output current IOUT output to the load 24 . In order to make the charging current ICHARGE proportional to the output current IOUT , the value of the feedback voltage VACC every time it is sampled must be the same or stable. The update circuit 96, the potential converter 92, and the transconductor 90 together form a loop with a negative loop gain, and this loop may finally stabilize the value of the feedback voltage VACC each time it is sampled. in a value. For example, if the charging current ICHARGE is greater than an expected value proportional to the output current IOUT , then the feedback voltage VACC will increase in the next sampling, causing the updated voltage VM to increase accordingly. Therefore, the charging current ICHARGE will become smaller. vice versa. Therefore, the charging current ICHARGE can be changed approximately in proportion to the output current IOUT at the end.
图5A显示在一些实施例中，充电电流ICHARGE跟电压VM的关系。电压VM通过电位转换器92以及转导器90，控制了充电电流ICHARGE。如同图5A所示的，转导器90所输出的充电电流ICHARGE并没有负值，最低就是0A。当电压VM低于预设电压VREF-M(其对应到图4中的参考电压VREF)时，充电电流ICHARGE大约是一极大值，也就是图5A上所示的IMAX。FIG. 5A shows the relationship between the charging current ICHARGE and the voltage VM in some embodiments. The voltage VM controls the charging current ICHARGE through the potential converter 92 and the transducer 90 . As shown in FIG. 5A , the charging current ICHARGE output by the transconductor 90 has no negative value, and the lowest value is 0A. When the voltage VM is lower than the predetermined voltage VREF-M (which corresponds to the reference voltage VREF in FIG. 4 ), the charging current ICHARGE is about a maximum value, which is IMAX shown in FIG. 5A .
当负载24不大或是很轻时，输出电流IOUT还没有到最大额定输出电流，所以电压VM将稳定在大于预设电压VREF-M的某个值，使得充电电流ICHARGE跟输出电流IOUT成比例。此时，图3中的电源控制器26操作在定电压模式，调节输出电源VOUT的电压，使其稳定在预设值。但是当负载24非常重时，充电电流ICHARGE会固定在IMAX，且导致电压VM掉到预设电压VREF-M之下。此时，如果输出电流IOUT超过了其最大额定输出电流(跟IMAX比例)，那电压VM与限制电压VLIMIT两者都会随着一个个开关周期，而减少，直到限制电压VLIMIT压低了电压VCS-PEAK，使得输出电流IOUT等于最大额定输出电流。换句话说，当电压VM低于预设电压VREF-M时，电源控制器26会操作在定电流模式。When the load 24 is not heavy or very light, the output current IOUT has not yet reached the maximum rated output current, so the voltage VM will stabilize at a certain value greater than the preset voltage VREF-M , so that the charging current ICHARGE and the output current IOUT is proportional. At this time, the power controller 26 in FIG. 3 operates in a constant voltage mode, and regulates the voltage of the output power VOUT to stabilize it at a preset value. But when the load 24 is very heavy, the charging current ICHARGE will be fixed at IMAX and cause the voltage VM to drop below the preset voltage VREF-M . At this time, if the output current IOUT exceeds its maximum rated output current (proportional to IMAX ), both the voltage VM and the limit voltage VLIMIT will decrease with each switching cycle until the limit voltage VLIMIT is lowered The voltage VCS-PEAK is increased so that the output current IOUT is equal to the maximum rated output current. In other words, when the voltageVM is lower than the preset voltage VREF-M , the power controller 26 will operate in the constant current mode.
在定电压操作时，转导器90的转导增益大致决定了电压VM值的位置。转导增益越大，电压VM的可能位置范围越窄，反馈电压VACC就可以有更多的操作电压空间可以使用。但是，转导增益也不能太大，因为提高转导增益同时也提高了负回路增益，而过大的负回路增益将可能导致振荡，使得电压VM无法稳定。When operating at a constant voltage, the transconductance gain of the transducer 90 roughly determines the value of the voltage VM . The larger the transconductance gain, the narrower the possible position range of the voltage VM is, and the more operating voltage space of the feedback voltage VACC can be used. However, the transconductance gain should not be too large, because increasing the transconductance gain also increases the negative loop gain, and too large negative loop gain may cause oscillation, making the voltage VM unstable.
图5B显示在一些实施例中，偏压电流IOFFSET跟充电电流ICHARGE的关系，这关系可由负载补偿电路66所执行。在一实施例中，负载代表电压VLC与偏压电流IOFFSET可以通过映射充电电流ICHARGE来产生。一些实施例中，当负载很轻或是无载时，负载补偿并不需要产生。因此，如同图5B所示的，当充电电流ICHARGE比一预设值IREF来的低时，偏压电流IOFFSET就消失，等于0A。当充电电流ICHARGE比一预设值IREF高时，两者大约是一个线性关系，如同图5B中的斜直线103所举例的。FIG. 5B shows the relationship between the bias current IOFFSET and the charging current ICHARGE in some embodiments, which can be implemented by the load compensation circuit 66 . In an embodiment, the load representative voltage VLC and the bias current IOFFSET can be generated by mapping the charging current ICHARGE . In some embodiments, load compensation does not need to occur when the load is very light or no load. Therefore, as shown in FIG. 5B , when the charging current ICHARGE is lower than a preset value IREF , the bias current IOFFSET disappears and is equal to 0A. When the charging current ICHARGE is higher than a predetermined value IREF , there is approximately a linear relationship between them, as exemplified by the inclined line 103 in FIG. 5B .
输出电流估计器70仅仅使用了一个内部、具有负回路增益的一回路，就达到了两个重要的功能：提供跟输出电流IOUT成比例的充电电流ICHARGE，以及控制输出电流IOUT不超过最大额定输出电流。The output current estimator 70 only uses an internal primary loop with a negative loop gain to achieve two important functions: providing a charging current ICHARGE proportional to the output current IOUT , and controlling the output current IOUT not to exceed maximum rated output current.
以上所述仅为本发明的优选实施例，凡依本发明权利要求书所做的均等变化与修饰，皆应属本发明的涵盖范围。The above descriptions are only preferred embodiments of the present invention, and all equivalent changes and modifications made according to the claims of the present invention shall fall within the scope of the present invention.
| Application Number | Priority Date | Filing Date | Title |
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| CN201310698183.4ACN104734506A (en) | 2013-12-18 | 2013-12-18 | Power supply controller capable of providing load compensation and related control method |
| Application Number | Priority Date | Filing Date | Title |
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| CN201310698183.4ACN104734506A (en) | 2013-12-18 | 2013-12-18 | Power supply controller capable of providing load compensation and related control method |
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| CN201310698183.4APendingCN104734506A (en) | 2013-12-18 | 2013-12-18 | Power supply controller capable of providing load compensation and related control method |
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