技术领域technical field
本发明涉及电力控制设备，尤其是一种反激功率因数校正变换器的控制方法及其装置。The invention relates to power control equipment, in particular to a control method and device for a flyback power factor correction converter.
背景技术Background technique
近年来，电力电子技术迅速发展，作为电力电子领域重要组成部分的电源技术逐渐成为应用和研究的热点。开关电源以其效率高、功率密度高的特点而确立了其在电源领域中的主流地位，但其通过整流器接入电网时会存在一个致命的弱点：功率因数较低(一般仅为0.45～0.75)，且在电网中会产生大量的电流谐波和无功功率而污染电网。抑制开关电源产生谐波的方法主要有两种：一是被动法，即采用无源滤波或有源滤波电路来旁路或消除谐波；二是主动法，即设计新一代高性能整流器，它具有输入电流为正弦波、谐波含量低以及功率因数高等特点，即具有功率因数校正功能。开关电源功率因数校正研究的重点主要是功率因数校正电路拓扑的研究和功率因数校正控制集成电路的开发。传统的有源功率因数校正电路一般采用Boost-升压拓扑，这是因为Boost变换器具有控制容易、驱动简单以及功率因数可以接近于1等优点，但是Boost功率因数校正变换器在低输出电压时，其功率因数却很低。在小功率的应用场合中，主要采用Buck-降压拓扑和反激变换器，但是在Buck电路实现PFC时，由于当输入电压低于输出电压时，不传递能量，输入电流为0，交越失真严重；而反激变换器在整个工频周期内都可以传递能量，功率因数和总谐波畸变都优于Buck变换器，因此更加适用于PFC领域当中。反激功率因数校正变换器通常有断续模式和临界连续模式两种工作模式。传统的临界连续模式反激功率因数校正变换器的控制复杂，且输入电流畸变严重，因此断续模式的反激功率因数校正变换器应用更加广泛。传统的断续模式反激功率因数校正变换器可以自动获得单位功率因数，但是由于其较大的峰值电流，使得原边开关管、副边二极管的导通损耗很大从而影响变换器的效率。传统的断续模式反激功率因数校正变换器的控制方式如图1所示，采用单电压环的控制方法，保持原边开关管占空比恒定，使变换器获得单位功率因数；但由于变压器的漏感吸收一部分能量且不能被利用，导致变换器的工作效率非常低。In recent years, power electronics technology has developed rapidly, and power supply technology, which is an important part of the power electronics field, has gradually become a hot spot in application and research. Switching power supply has established its mainstream position in the field of power supply due to its high efficiency and high power density, but there is a fatal weakness when it is connected to the power grid through a rectifier: the power factor is low (generally only 0.45 to 0.75 ), and a large number of current harmonics and reactive power will be generated in the grid to pollute the grid. There are two main methods for suppressing harmonics generated by switching power supplies: one is the passive method, that is, using passive filtering or active filtering circuits to bypass or eliminate harmonics; the other is the active method, that is, designing a new generation of high-performance rectifiers, which It has the characteristics of sine wave input current, low harmonic content and high power factor, that is, it has the function of power factor correction. The focus of the research on power factor correction of switching power supply is mainly the research of power factor correction circuit topology and the development of power factor correction control integrated circuit. The traditional active power factor correction circuit generally adopts the Boost-boost topology, because the Boost converter has the advantages of easy control, simple driving, and power factor close to 1, but the Boost power factor correction converter has the advantages of low output voltage , but its power factor is very low. In low-power applications, Buck-step-down topology and flyback converter are mainly used, but when the Buck circuit implements PFC, because when the input voltage is lower than the output voltage, no energy is transferred, the input current is 0, and the crossover The distortion is serious; while the flyback converter can transfer energy in the whole power frequency cycle, the power factor and total harmonic distortion are better than the Buck converter, so it is more suitable for the PFC field. Flyback power factor correction converters usually have two operating modes: discontinuous mode and critical continuous mode. The traditional critical continuous mode flyback power factor correction converter has complex control and serious input current distortion, so discontinuous mode flyback power factor correction converter is more widely used. The traditional discontinuous mode flyback power factor correction converter can automatically obtain unity power factor, but due to its large peak current, the conduction loss of the primary side switch and secondary side diode is very large, which affects the efficiency of the converter. The control method of the traditional discontinuous mode flyback power factor correction converter is shown in Figure 1. The single voltage loop control method is adopted to keep the duty cycle of the primary side switch constant, so that the converter can obtain unity power factor; but because the transformer The leakage inductance absorbs part of the energy and cannot be used, resulting in very low efficiency of the converter.
发明内容Contents of the invention
本发明的目的是提供一种新颖的反激功率因数校正变换器的拓扑及控制方法，采用该拓扑结构和控制方法可使反激功率因数校正变换器获得高功率因数和高效率。The object of the present invention is to provide a novel topology and control method of a flyback power factor correction converter, which can make the flyback power factor correction converter obtain high power factor and high efficiency.
本发明实现其发明目的，所采用的技术方案是一种断续模式的高功率因数高效率反激变换器的控制装置，其具体作法是：The present invention realizes its purpose of the invention, and the technical solution adopted is a control device of a high power factor high efficiency flyback converter of discontinuous mode, and its concrete practice is:
一种断续模式的高功率因数高效率反激变换器的控制装置，反激变换器的控制电路包括输出电压采样电路；误差放大器；PWM脉冲产生电路；占空比运算电路；其中，反激变压器副边串联一个谐振电容Cr和并联一个快恢复二极管D01，谐振电容Cr与变压器副边漏感Llk发生谐振，将漏感中的能量全部吸收利用；PWM脉冲产生电路由比较器和锯齿波发生器组成；误差放大器的负输入端连接到由R7和R8分压电阻网络构成的输出电压采样电路的输出端，误差放大器的正输入端为基准电压Vref，输出电压采样电路的输出信号与基准电压相比较产生误差放大器的输出信号Vcomp；占空比运算电路的输出信号VD(t)连接到PWM脉冲产生电路中比较器的负输入端，锯齿波发生器的输出信号连接到PWM脉冲产生电路中比较器的正输入端。所述占空比运算电路包括第一运算放大器、第二运算放大器、第三运算放大器和乘法器：由电阻R1和R2分压电阻网络采样的输入电压连接到第一运算放大器的正输入端，第一运算放大器的负输入端连接到其输出端；第一运算放大器的输出端还通过电阻R3连接到峰值计算网络中整流二极管D3的阳极，整流二极管D3的阴极通过充电电容C3接地；整流二极管D3的阴极还连接到第二运算放大器的正输入端，第二运算放大器的负输入端连接到其输出端；第一运算放大器和第二运算放大器的输出端分别通过电阻R4和电阻R5连接到第三运算放大器的负输入端；变压器的辅助绕组电压通过二极管D2、电容C2整流滤波后连接到第三运算放大器的正输入端；第三运算放大器的负输入端还通过电阻R6连接到其输出端；误差放大器与第三运算放大器的输出端分别连接到乘法器的两个输入端；锯齿波发生器和乘法器的输出端分别连接到比较器的正、负输入端，比较器的输出端连接到开关管的驱动端。A control device for a discontinuous mode high power factor and high efficiency flyback converter, the control circuit of the flyback converter includes an output voltage sampling circuit; an error amplifier; a PWM pulse generating circuit; a duty ratio operation circuit; wherein, the flyback The secondary side of the transformer is connected in series with a resonant capacitor Cr and a fast recovery diode D01 in parallel. The resonant capacitor Cr resonates with the leakage inductance Llk of the secondary side of the transformer to absorb and utilize all the energy in the leakage inductance; the PWM pulse generating circuit consists of a comparator and a sawtooth wave generator; the negative input terminal of the error amplifier is connected to the output terminal of the output voltage sampling circuit composed of R7 and R8 voltage dividing resistor network, the positive input terminal of the error amplifier is the reference voltage Vref , the output voltage sampling circuit The output signal is compared with the reference voltage to generate the output signal Vcomp of the error amplifier; the output signal VD (t) of the duty cycle operation circuit is connected to the negative input terminal of the comparator in the PWM pulse generation circuit, and the output signal of the sawtooth wave generator Connect to the positive input of the comparator in the PWM pulse generating circuit. The duty cycle operation circuit includes a first operational amplifier, asecond operational amplifier, a third operational amplifier anda multiplier: the input voltage sampled by the resistor R1 and R2 voltage divider resistor network is connected to the positive input of the first operational amplifier Terminal, the negative input terminal of the first operational amplifier is connected to its output terminal; the output terminal of the first operational amplifier is also connected to the anode of the rectifier diodeD3 in the peak calculation network through the resistorR3 , and the cathode of the rectifier diodeD3 is charged through the charging capacitor C3 is grounded; the cathode of the rectifier diode D3 is also connected to the positive input of the second operational amplifier, and the negative input of the second operational amplifier is connected to its output; the output terminals of the first operational amplifier and the second operational amplifier are respectively passed Resistor R4 and resistor R5 are connected to the negative input terminal of the third operational amplifier; the auxiliary winding voltage of the transformer is rectified and filtered by diode D2 and capacitor C2 and then connected to the positive input terminal of the third operational amplifier; the third operational amplifier The negative input terminal is also connected to its output terminal through a resistor R6; the output terminals of the error amplifier and the third operational amplifier are respectively connected to the two input terminals of the multiplier; the output terminals of the sawtooth wave generator and the multiplier are respectively connected to the comparator The positive and negative input terminals of the comparator are connected to the drive terminal of the switch tube.
本发明的另一目的是提供一种断续模式的高功率因数高效率反激变换器的控制方法，以上述装置为实施硬件，其具体方案为：Another object of the present invention is to provide a control method of a discontinuous mode high power factor high efficiency flyback converter, with the above-mentioned device as the implementation hardware, the specific scheme is:
一种断续模式的高功率因数高效率反激变换器的控制方法，采用上述装置，采用占空比运算电路的运算方法实现导通占空比在半个工频周期内随着输入电压和输出电压的变化而变化，从而获得高功率因数，包括如下的处理过程：匝比n的2倍与输出电压Vo相乘后的值2n*Vo，减去输入电压的瞬时值Vin(t)以及K0倍的输入电压峰值K0*Vm，相减的结果的D0倍再乘以误差放大器的输出电压Vcomp，所得结果再除以即得到占空比运算电路运算的输出值；其中匝比n为反激变换器的变压器原边绕组匝数与副边绕组匝数的比值。A control method for a high power factor and high efficiency flyback converter in discontinuous mode, using the above-mentioned device, using the calculation method of the duty ratio operation circuit to realize that the conduction duty ratio increases with the input voltage and the input voltage within half a power frequency cycle The output voltage changes to obtain a high power factor, including the following process: the value 2n*Vo after multiplying the output voltage Vo by twice the turn ratio n, and subtracting the instantaneous value of the input voltage Vin ( t) and K0 times the input voltage peak value K0 *Vm , the result of the subtraction is multiplied by D0 times the output voltage Vcomp of the error amplifier, and the result is divided by That is, the output value of the duty cycle operation circuit is obtained; the turn ratio n is the ratio of the number of turns of the primary winding of the flyback converter to the number of turns of the secondary winding of the flyback converter.
在实际实施中，匝比n与输出电压Vo相乘后的值n*Vo，是通过反激变换器辅助绕组获得，即n*Vo＝n*Vcc/nas，其中Vcc为反激变换器的辅助绕组电压经过二极管D2、电容C2整流滤波后的电压值，nas为反激变换器辅助绕组与副边绕组匝数的比值。In actual implementation, the value n*Vo obtained by multiplying the turn ratio n by the output voltage Vo is obtained through the auxiliary winding of the flyback converter, that is, n*Vo =n*Vcc /nas , where Vcc is the voltage value of the auxiliary winding voltage of the flyback converter rectified and filtered by the diode D2 and the capacitor C2 , and nas is the ratio of the number of turns of the auxiliary winding of the flyback converter to the number of turns of the secondary winding.
与现有技术相比，本发明的有益效果是：Compared with prior art, the beneficial effect of the present invention is:
相对于传统断续模式的反激功率因数校正变换器，采用本发明的断续模式高功率因数高效率反激变换器的拓扑及其控制方式，可以获得较高的工作效率，同时还保留了传统断续模式工作方式的高功率因数和低总谐波畸变的特征。Compared with the traditional discontinuous mode flyback power factor correction converter, the topology and control method of the discontinuous mode high power factor high efficiency flyback converter of the present invention can obtain higher working efficiency while retaining High power factor and low total harmonic distortion are characteristic of traditional discontinuous mode operation.
本发明的另一目的是提供一种实现以上高功率因数和高效率方法的装置。其具体构造采用：Another object of the present invention is to provide an apparatus for realizing the above high power factor and high efficiency method. Its specific structure adopts:
断续模式的高功率因数高效率反激变换器的控制电路包括输出电压采样与误差放大器电路、PWM脉冲产生电路、占空比运算电路以及驱动电路；其中PWM脉冲产生电路由比较器和锯齿波发生器组成；误差放大器的负向输入端为由R7和R8分压电阻网络采样的输出电压，误差放大器的正向输入端为基准电压Vref，采样的输出信号与基准电压相比较产生误差放大器的输出信号Vcomp；占空比运算电路的输出VD(t)连接到PWM脉冲产生电路中比较器的负端，锯齿波发生器的输出信号连接到PWM脉冲产生电路中比较器的正端；可见，采用以上装置可以方便可靠地实现本发明以上方法。The control circuit of the flyback converter with high power factor and high efficiency in discontinuous mode includes an output voltage sampling and error amplifier circuit, a PWM pulse generating circuit, a duty ratio operation circuit and a driving circuit; the PWM pulse generating circuit is composed of a comparator and a sawtooth wave generator; the negative input of the error amplifier is the output voltage sampled by the R7 and R8 voltage divider resistor network, the positive input of the error amplifier is the reference voltage Vref , and the sampled output signal is compared with the reference voltage to generate The output signal Vcomp of the error amplifier; the output VD (t) of the duty cycle operation circuit is connected to the negative terminal of the comparator in the PWM pulse generating circuit, and the output signal of the sawtooth wave generator is connected to the comparator in the PWM pulse generating circuit Positive end; It can be seen that the above method of the present invention can be realized conveniently and reliably by using the above device.
附图说明Description of drawings
图1为传统断续模式的反激功率因数校正变换器的系统结构框图。Figure 1 is a block diagram of the system structure of a traditional discontinuous mode flyback power factor correction converter.
图2为本发明的断续模式的高功率因数高效率反激变换器的系统结构框图。Fig. 2 is a system structural block diagram of a discontinuous mode high power factor high efficiency flyback converter of the present invention.
图3为图2所示电路框图的一个开关周期的工作模态波形图。FIG. 3 is a working mode waveform diagram of one switching cycle of the circuit block diagram shown in FIG. 2 .
图4为本发明实施例子的电路结构示意图。Fig. 4 is a schematic circuit structure diagram of an embodiment of the present invention.
图5为图1传统断续模式的反激功率因数校正变换器和图2本发明的断续模式的高功率因数高效率反激变换器的输入电流对比仿真波形图。FIG. 5 is a comparison simulation waveform diagram of the input current of the traditional discontinuous mode flyback power factor correction converter of FIG. 1 and the discontinuous mode high power factor high efficiency flyback converter of the present invention shown in FIG. 2 .
图6为图1传统断续模式的反激功率因数校正变换器和图2本发明的断续模式的高功率因数高效率反激变换器变压器的原副边电流对比仿真波形图。Fig. 6 is a comparison simulation waveform diagram of the primary and secondary side currents of the traditional discontinuous mode flyback power factor correction converter in Fig. 1 and the discontinuous mode high power factor high efficiency flyback converter transformer in Fig. 2 of the present invention.
图7为图1传统断续模式的反激功率因数校正变换器和图2本发明的断续模式的高功率因数高效率反激变换器变压器的PF值对比。Fig. 7 is a comparison of the PF values of the traditional discontinuous mode flyback power factor correction converter in Fig. 1 and the discontinuous mode high power factor high efficiency flyback converter transformer of the present invention in Fig. 2 .
图8为图1传统断续模式的反激功率因数校正变换器和图2本发明的断续模式的高功率因数高效率反激变换器变压器的效率对比。FIG. 8 is a comparison of the efficiency of the traditional discontinuous mode flyback power factor correction converter in FIG. 1 and the discontinuous mode high power factor high efficiency flyback converter transformer of the present invention in FIG. 2 .
具体实施方式detailed description
下面通过具体的实例并结合附图对本发明做进一步详细的描述。The present invention will be further described in detail through specific examples and in conjunction with the accompanying drawings.
图2为本发明的系统结构框图，图4为本发明的一种具体实施方式，断续模式的高功率因数高效率反激变换器的拓扑结构和控制方法。FIG. 2 is a system structure diagram of the present invention, and FIG. 4 is a specific embodiment of the present invention, a topological structure and a control method of a flyback converter with high power factor and high efficiency in discontinuous mode.
其连接方式为：由R1和R2分压电阻网络采样的输入电压连接到运算放大器1(电压跟随器)；运算放大器1的输出连接到峰值计算网络，峰值计算网络得到的峰值电压连接到运算放大器2(电压跟随器)；运算放大器1、2的输出分别通过电阻R4和R5连接到运算放大器3的负向输入端；变压器的辅助绕组电压通过二极管整流后连接到运算放大器3的正向输入端。由R7和R8分压电阻网络采样的输出电压连接到误差放大器的负向输入端，误差放大器的正向输入端为基准电压；误差放大器与运算放大器3的输出分别连接到乘法器的两个输入端；三角波发生器和乘法器的输出分别连接到比较器的正负输入端，比较器的输出连接到开关管的驱动端。Its connection method is: the input voltage sampled by the R1 and R2 voltage divider resistor network is connected to the operational amplifier1 (voltage follower); the output of the operational amplifier 1 is connected to the peak calculation network, and the peak voltage obtained by the peak calculation network is connected to Operational amplifier 2 (voltage follower); the outputs of operational amplifiers 1 and2 are connected to the negative input of operational amplifier3 through resistors R4 and R5 respectively; the auxiliary winding voltage of the transformer is rectified by a diode and then connected to the negative input of operational amplifier 3 positive input. The output voltage sampled by the R7 and R8 divider resistor network is connected to the negative input terminal of the error amplifier, and the positive input terminal of the error amplifier is the reference voltage; the output of the error amplifier and the operational amplifier 3 are respectively connected to the two terminals of the multiplier The outputs of the triangular wave generator and the multiplier are respectively connected to the positive and negative inputs of the comparator, and the output of the comparator is connected to the drive terminal of the switch tube.
其具体作法是：反激变换器的控制电路包括输出电压采样与误差放大器电路、PWM脉冲产生电路、占空比运算电路以及驱动电路。误差放大器的负向输入端为由R7和R8分压电阻网络采样的输出电压，误差放大器的正向输入端为基准电压Vref，采样的输出信号与基准电压相比较产生误差放大器的输出信号Vcomp。开关管导通占空比运算电路有输入电压Vin(t)、输出电压Vo和误差放大器的输出信号Vcomp三路输入信号，占空比运算电路包括3个运算放大器、1个乘法器和若干电阻网络；整流后的输入电压Vin(t)经过R1、R2分压，再经过由运算放大器1组成的电压跟随器后，得到VA＝K*Vin(t)，K为R1、R2电压网络的分压系数；VA通过峰值计算网络并通过由运算放大器2组成的电压跟随器得到其峰值Vm，采集原边辅助绕组上的电压Vcc代替输出电压Vo，Vcc＝nA*Vo/nS；R4、R5、R6和运算放大器3共同构成加法器，VA、Vm和Vcc同时作为加法器的输入端，通过设定K、R4、R5、R6、nA和nP，使其满足：R6*(1/R4+1/R5+1/R6)＝2*nP/nA，K*R6/R5＝K0，K*R6/R4＝1，nP、nA为变压器原边、辅助边绕组匝数。加法器的输出Vy和Vcomp分别连接到乘法器的输入端，乘法器的输出信号为占空比运算电路的输出信号VD(t)；占空比运算电路的输出VD(t)信号连接到PWM脉冲产生电路中比较器的负端，PWM脉冲产生电路中比较器的正端接入锯齿波发生器的输出信号，当占空比运算电路的输出信号VD(t)大于锯齿波发生器的输出信号时，开关管Q导通；反之，开关管Q关断。反激变换器原边开关管的导通占空比受误差放大器输出电压Vcomp、输入电压Vin(t)和输出电压Vo的控制；设定误差放大器电路的补偿使整个环路的截止频率远小于工频(一般为10～20Hz)，误差放大器输出信号Vcomp在半个工频周期内维持不变。The specific method is as follows: the control circuit of the flyback converter includes an output voltage sampling and error amplifier circuit, a PWM pulse generation circuit, a duty ratio calculation circuit and a drive circuit. The negative input terminal of the error amplifier is the output voltage sampled by the resistor network of R7 and R8 , the positive input terminal of the error amplifier is the reference voltage Vref , the output signal of the sample is compared with the reference voltage to generate the output of the error amplifier Signal Vcomp . The switching tube conduction duty cycle operation circuit has three input signals of input voltage Vin (t), output voltage Vo and error amplifier output signal Vcomp , and the duty cycle operation circuit includes 3 operational amplifiers and 1 multiplier and a number of resistor networks; the rectified input voltage Vin (t) is divided by R1 and R2 , and then passes through the voltage follower composed of operational amplifier 1 to obtain VA =K*Vin (t), K is the voltage division coefficient of the R1 and R2 voltage network; VA obtains its peak value Vm through the peak calculation network and the voltage follower composed of the operational amplifier 2, and collects the voltage Vcc on the auxiliary winding of the primary side instead of the output voltage Vo , Vcc ＝nA *Vo /nS ; R4 , R5 , R6 and the operational amplifier 3 together constitute an adder, and VA , Vm and Vcc serve as the input terminals of the adder at the same time, by setting K, R4 , R5 , R6 , nA and nP , so that it satisfies: R6 *(1/R4 +1/R5 +1/R6 )=2*nP /nA , K *R6 /R5 ＝K0 , K*R6 /R4 ＝1, nP , nA are the turns of primary and auxiliary windings of the transformer. The output Vy and Vcomp of the adder are respectively connected to the input terminals of the multiplier, and the output signal of the multiplier is the output signal VD (t) of the duty cycle operation circuit; the output VD (t) of the duty cycle operation circuit The signal is connected to the negative terminal of the comparator in the PWM pulse generating circuit, and the positive terminal of the comparator in the PWM pulse generating circuit is connected to the output signal of the sawtooth wave generator. When the output signal VD (t) of the duty cycle operation circuit is greater than the sawtooth When the output signal of the wave generator is applied, the switching tube Q is turned on; otherwise, the switching tube Q is turned off. The on-duty ratio of the switch tube on the primary side of the flyback converter is controlled by the error amplifier output voltage Vcomp , the input voltage Vin (t) and the output voltage Vo ; setting the compensation of the error amplifier circuit makes the entire loop cut-off The frequency is much lower than the power frequency (generally 10-20 Hz), and the output signal Vcomp of the error amplifier remains unchanged within half of the power frequency cycle.
图5和图6是利用PSIM仿真软件得到的仿真波形。从图5可以看出：传统断续模式的反激功率因数校正变换器的输入电流自动跟踪输入电压波形，功率因数较高；采用变占空比控制断续模式的高功率因数高效率反激变换器的输入电流也很好地跟踪了输入电压的波形，具有较高的功率因数。从图6可以看出：传统断续模式反激功率因数校正变换器的导通时间不变，即占空比不变；而本发明的断续模式高功率因数高效率反激变换器的占空比实时变化，其值与输入电压瞬时值、输出电压有关。Figure 5 and Figure 6 are simulation waveforms obtained by using PSIM simulation software. It can be seen from Figure 5 that the input current of the traditional discontinuous mode flyback power factor correction converter automatically tracks the input voltage waveform, and the power factor is high; the high power factor and high efficiency flyback of the discontinuous mode is controlled by variable duty ratio The input current of the converter also tracks the waveform of the input voltage very well, and has a high power factor. It can be seen from Fig. 6 that the conduction time of the traditional discontinuous mode flyback power factor correction converter remains unchanged, that is, the duty cycle remains unchanged; while the duty cycle of the discontinuous mode high power factor high efficiency flyback converter of the present invention is The empty ratio changes in real time, and its value is related to the instantaneous value of the input voltage and the output voltage.
图7为传统断续模式的反激功率因数校正变换器和本发明的断续模式的高功率因数高效率反激变换器的PF值对比。从图7中可以看出：本发明的变换器基本上保留了传统变换器的高功率因数的特征。Fig. 7 is a comparison of PF values between a traditional discontinuous mode flyback power factor correction converter and the discontinuous mode high power factor high efficiency flyback converter of the present invention. It can be seen from Fig. 7 that the converter of the present invention basically retains the characteristics of high power factor of the traditional converter.
图8传统断续模式的反激功率因数校正变换器和本发明的断续模式的高功率因数高效率反激变换器的效率对比。从图8可以看出：在两种变换器参数完全一样的条件下，本发明的断续模式的高功率因数高效率反激变换器的工作效率相比于传统断续模式的反激功率因数校正变换器得到了极大的提高。Fig. 8 Efficiency comparison of a conventional discontinuous mode flyback power factor correction converter and a discontinuous mode high power factor high efficiency flyback converter of the present invention. It can be seen from Fig. 8 that under the same condition of the parameters of the two converters, the operating efficiency of the flyback converter with high power factor and high efficiency in the discontinuous mode of the present invention is higher than that of the flyback power factor in the traditional discontinuous mode Correction transformers have been greatly improved.
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| CN201410768023.7ACN104578797B (en) | 2014-12-12 | 2014-12-12 | Method and device for controlling discontinuous mode flyback converter with high power factor and high efficiency |
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| CN201410768023.7ACN104578797B (en) | 2014-12-12 | 2014-12-12 | Method and device for controlling discontinuous mode flyback converter with high power factor and high efficiency |
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