CN104242717B

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Publication number: CN104242717B
Application number: CN201410466131.9A
Authority: CN
Inventors: 刘芳; 张兴; 石荣亮; 徐海珍; 胡超; 余畅舟; 李文超; 毛福斌
Original assignee: Hefei University of Technology
Current assignee: Hefei Luyang Technology Innovation Group Co ltd
Priority date: 2014-09-12
Filing date: 2014-09-12
Publication date: 2016-09-14
Anticipated expiration: 2034-09-12
Also published as: CN104242717A

Abstract

本发明公开了一种基于虚拟同步机的自适应输出阻抗控制方法。它首先提出了一种稳态输出阻抗控制器来增大稳态输出阻抗，以保证并联稳定运行时的并联均流度，然后给出了一种电容电压复合微分控制器来增大系统的阻尼，增强系统稳定性，最后，采用一种动态输出阻抗控制器，减小动态过程的输出阻抗，减小电压跳变，从而提高了输出电压的动态响应。它可广泛地应用于不同负载条件下的微网逆变器控制，以同时既满足微网逆变器离网运行时的动稳态性能，又能多台并联运行。

The invention discloses an adaptive output impedance control method based on a virtual synchronous machine. It firstly proposes a steady-state output impedance controller to increase the steady-state output impedance to ensure parallel current sharing during parallel stable operation, and then provides a capacitor-voltage composite differential controller to increase the damping of the system , to enhance the stability of the system, and finally, a dynamic output impedance controller is used to reduce the output impedance of the dynamic process and reduce the voltage jump, thereby improving the dynamic response of the output voltage. It can be widely used in micro-grid inverter control under different load conditions, so as to satisfy the dynamic and steady-state performance of micro-grid inverters when they are running off-grid, and to allow multiple parallel operation of micro-grid inverters.

Description

Translated fromChinese

基于虚拟同步机的自适应输出阻抗控制方法Adaptive output impedance control method based on virtual synchronous machine

技术领域technical field

本发明涉及一种自适应输出阻抗控制方法，尤其是一种基于虚拟同步机的自适应输出阻抗控制方法。The invention relates to an adaptive output impedance control method, in particular to an adaptive output impedance control method based on a virtual synchronous machine.

背景技术Background technique

近年来，虚拟同步机技术作为微网逆变器的一种新型的发电模式，受到了学者们的大量关注。采用虚拟同步机技术的微网逆变器叫做虚拟同步机。虚拟同步机(VirtualSynchronous Generator，VSG)需要运行在两种模式下，并网和孤岛并联运行。VSG在离网运行时需要带不同的负载，如，阻感容负载、电机马达负载、整流桥负载，以及各种有源负载。在不同的负载情况下，其动态响应和稳态性能各不相同。为了达到良好的动稳态性能，特别是较快的动态响应，要求尽量减小VSG的输出阻抗。然而，VSG又需要并联运行，并联稳定运行的前提条件是多台机器之间能够按照功率等级进行功率分配，这就需要VSG增大输出阻抗以满足功率分配的需求。总之，输出阻抗是决定VSG动稳态输出误差、功率均分精度的重要因素。如何既能保持良好的动稳态性能，又能提高功率均分精度，合理的设计输出阻抗成为VSG亟待解决的关键问题之一。In recent years, virtual synchronous machine technology, as a new type of power generation mode for microgrid inverters, has received a lot of attention from scholars. The microgrid inverter using virtual synchronous machine technology is called virtual synchronous machine. The Virtual Synchronous Generator (VSG) needs to run in two modes, grid-connected and island-connected. VSG needs to carry different loads when running off-grid, such as resistance-inductance-capacitance loads, motor loads, rectifier bridge loads, and various active loads. Under different load conditions, its dynamic response and steady-state performance are different. In order to achieve good dynamic steady-state performance, especially faster dynamic response, it is required to minimize the output impedance of VSG. However, VSG needs to run in parallel. The prerequisite for stable parallel operation is that multiple machines can be distributed according to the power level, which requires VSG to increase the output impedance to meet the power distribution requirements. In short, the output impedance is an important factor that determines the VSG dynamic steady-state output error and power sharing accuracy. How to not only maintain good dynamic stability performance, but also improve the accuracy of power sharing, and reasonably design the output impedance has become one of the key issues to be solved urgently for VSG.

针对输出阻抗设计问题，国内外的专家学者们提出了一些方法，主要有：For the problem of output impedance design, experts and scholars at home and abroad have proposed some methods, mainly including:

题为“Output Impedance Design of Parallel-Connected UPS Inverters WithWireless Load-Sharing Control”，J.M.Guerrero，《Industrial Electronics，IEEETransactions on》，vol.52，pp.1126-1135，2005(“基于无互联线负载均流控制的并联UPS输出阻抗设计”，《IEEE学报-工业电子期刊》，2005年第52卷1126～1135页)的文章。该文提出了将输出阻抗设计成阻感性，低频呈感性、高频呈阻性；然而所提出的控制方法中，输出阻抗并没有减小控制系统高频处所固有的输出阻抗，在负载动态变化时，输出电压将剧烈变化，同时也没有考虑大功率应用场合。Entitled "Output Impedance Design of Parallel-Connected UPS Inverters With Wireless Load-Sharing Control", J.M.Guerrero, "Industrial Electronics, IEEE Transactions on", vol.52, pp.1126-1135, 2005 ("Based on Wireless Load-Sharing Control Controlled parallel UPS output impedance design", "IEEE Transactions-Journal of Industrial Electronics", 2005, Vol. 52, pp. 1126-1135). This paper proposes to design the output impedance to be resistive and inductive, which is inductive at low frequencies and resistive at high frequencies; however, in the proposed control method, the output impedance does not reduce the inherent output impedance at high frequencies of the control system. , the output voltage will change drastically, and high-power applications are not considered.

题为“Analysis，Design，and Implementation of Virtual Impedance forPower Electronics Interfaced Distributed Generation”，H.Jinwei and L.Yun Wei，《Industry Applications，IEEE Transactions on》，vol.47，pp.2525-2538，2011(“基于电力电子装置接口的分布式发电虚拟阻抗分析，设计及实现”，《IEEE学报-工业应用期刊》，2011年第47卷2525～2538页)的文章。该文提出了将输出阻抗设计成阻感性，并提出了根据无功功率来自适应变化输出阻抗大小，根据有功和无功自适应变化输出电压幅值的方法；然而所提控制方法中，有功和无功功率的计算中存在低通滤波器，且需要通过内环控制才能实现，响应速度仍然不够快速，无法达到负载突变时的输出电压动态响应要求。且此控制方法也没有考虑大功率应用场合。Titled "Analysis, Design, and Implementation of Virtual Impedance for Power Electronics Interfaced Distributed Generation", H.Jinwei and L.Yun Wei, "Industry Applications, IEEE Transactions on", vol.47, pp.2525-2538, 2011 (" Analysis, Design and Implementation of Virtual Impedance of Distributed Generation Based on Power Electronics Device Interface", "IEEE Journal-Industrial Application Journal", 2011, Vol. 47, pp. 2525-2538). This paper proposes to design the output impedance to be resistive and inductive, and proposes a method of adaptively changing the output impedance according to reactive power, and adaptively changing the output voltage amplitude according to active and reactive power; however, in the proposed control method, active power and There is a low-pass filter in the calculation of reactive power, and it needs to be realized through the inner loop control. The response speed is still not fast enough to meet the dynamic response requirements of the output voltage when the load changes suddenly. Moreover, this control method does not consider high-power applications.

题为“Control of Inverters Via a Virtual Capacitor to AchieveCapacitive Output Impedance”，Q.C.Zhong and Y.Zeng，《Power Electronics,IEEETransactions on》，vol.29，pp.5568-5578，2014(“基于虚拟电容的逆变器容性输出阻抗控制”，《IEEE学报-电力电子期刊》，2014年第29卷5568～5578页)的文章。该文提出了将输出阻抗控制为容性，可以滤除直流分量，并减小低次谐波；这些方法虽在小功率单相系统中得到了很好的验证，但将其应用到三相大功率场合时却存在困难，且也未阐述其动态响应问题。Titled "Control of Inverters Via a Virtual Capacitor to Achieve Capacitive Output Impedance", Q.C.Zhong and Y.Zeng, "Power Electronics, IEEE Transactions on", vol.29, pp.5568-5578, 2014 ("Inverter based on virtual capacitor capacitive output impedance control", "IEEE Transactions-Journal of Power Electronics", 2014, Vol. 29, pp. 5568-5578). This paper proposes to control the output impedance to be capacitive, which can filter out the DC component and reduce low-order harmonics; although these methods have been well verified in low-power single-phase systems, they are applied to three-phase However, there are difficulties in high-power occasions, and the dynamic response problem has not been described.

题为“A Simple Control Method for High-Performance UPS InvertersThrough Output-Impedance Reduction”，Heng Deng and Ramesh Oruganti，《IndustrialElectronics，IEEE Transactions on》55(2)，888-898，2008(“一种基于减小输出阻抗的高性能UPS控制方法”，《IEEE学报-工业电子期刊》，2008年第55卷第2期888～898页)的文章。该文提出了一种减小UPS输出阻抗的控制方法，该方法提出的数字反馈控制器可以极大地减小UPS的输出阻抗，具有良好的动稳态性能；然而，当UPS并联运行时过小的输出阻抗将不利于负载均流，给UPS并联运行带来困难。Entitled "A Simple Control Method for High-Performance UPS Inverters Through Output-Impedance Reduction", Heng Deng and Ramesh Oruganti, "Industrial Electronics, IEEE Transactions on" 55(2), 888-898, 2008 ("A Simple Control Method Based on Reducing Output-Impedance Reduction" Impedance high-performance UPS control method", "IEEE Transactions-Journal of Industrial Electronics", 2008, Vol. 55, No. 2, pp. 888-898). This paper proposes a control method to reduce the output impedance of UPS. The digital feedback controller proposed by this method can greatly reduce the output impedance of UPS and has good dynamic and steady-state performance; The output impedance of the load will not be conducive to load current sharing, which will bring difficulties to the parallel operation of UPS.

总之，现有技术未提及输出动态响应和并联均流对输出阻抗要求的矛盾问题。并且，在大功率三相VSG应用场合，其输出阻抗特性又有不同。首先三相系统的abc坐标系变量向dq坐标系转换时存在着两个分量，一个是静态分量，一个是动态分量，其输出阻抗与单相系统的等效电路并不相同；另外，大功率VSG的开关频率较低，控制延时较大，给系统增加了不稳定的极点，尤其是多机并联运行时，其稳定性进一步降低，这使得系统必须降低带宽以维持系统稳定性，带宽的降低使得系统的动态性能进一步下降，这极大的限制了控制器和VSG无源参数的设计准则，使得大功率VSG的整体控制性能下降。目前，对于这两个问题，现有技术也鲜有论述和解决的方案。In short, the prior art does not mention the contradiction between output dynamic response and parallel current sharing requirements on output impedance. Moreover, in high-power three-phase VSG applications, the output impedance characteristics are different. First of all, there are two components when the abc coordinate system variable of the three-phase system is converted to the dq coordinate system, one is a static component and the other is a dynamic component, and its output impedance is different from the equivalent circuit of a single-phase system; in addition, high-power The switching frequency of VSG is low, and the control delay is large, which adds an unstable pole to the system, especially when multiple machines are running in parallel, its stability is further reduced, which makes the system must reduce the bandwidth to maintain system stability. The reduction further degrades the dynamic performance of the system, which greatly limits the design criteria of the controller and VSG passive parameters, and degrades the overall control performance of the high-power VSG. At present, for these two problems, there are few discussions and solutions in the prior art.

发明内容Contents of the invention

本发明要解决的技术问题为克服上述各种技术方案的局限性，针对基于虚拟同步机的微网逆变器、UPS等装置离网并联运行时，同一输出阻抗难以同时满足输出电压波形动静态性能和功率均分问题，提供一种基于虚拟同步机的自适应输出阻抗控制方法。The technical problem to be solved by the present invention is to overcome the limitations of the above-mentioned various technical solutions. When the virtual synchronous machine-based micro-grid inverter, UPS and other devices are running in parallel off-grid, it is difficult for the same output impedance to satisfy the dynamic and static state of the output voltage waveform at the same time. For performance and power sharing problems, an adaptive output impedance control method based on a virtual synchronous machine is provided.

为解决本发明的技术问题，所采用的技术方案为：基于虚拟同步机的自适应输出阻抗控制方法包括微网逆变器输出电容电压的采集，特别是，In order to solve the technical problem of the present invention, the technical solution adopted is: the adaptive output impedance control method based on the virtual synchronous machine includes the collection of the output capacitor voltage of the microgrid inverter, especially,

步骤1，先采集微网逆变器的输出电容电压U_ca,U_cb,U_cc、桥臂侧电感电流I_la,I_lb,I_lc和输出电流I_oa,I_ob,I_oc，再经过单同步旋转坐标变换得到输出电容电压dq的分量U_cd,U_cq、桥臂侧电感电流dq的分量I_ld,I_lq和输出电流dq的分量I_od,I_oq；Step 1, first collect the output capacitor voltage U_ca , U_cb , U_cc of the microgrid inverter, the bridge arm side inductor current I_la , I_lb , I_lc and the output current I_oa , I_ob , I_oc , and then pass The single synchronous rotation coordinate transformation obtains the components U_cd , U_cq of the output capacitor voltage dq, the components I_ld , I_lq of the inductor current dq on the bridge arm side, and the components I_od , I_oq of the output current dq;

步骤2，根据步骤1中得到的输出电流dq的分量I_od,I_oq，经过稳态输出阻抗控制方程得到电容电压dq的分量指令信号增量ΔU_dref,ΔU_qref；Step 2, according to the components I_od , I_oq of the output current dq obtained in the step 1, through the steady-state output impedance control equation, the component command signal increments ΔU_dref , ΔU_qref of the capacitor voltage dq are obtained;

步骤3，根据步骤1中得到的输出电容电压dq的分量U_cd,U_cq和输出电流dq的分量I_od,I_oq，经过有功功率计算方程和无功功率计算方程得到平均有功功率和平均无功功率Step 3, according to the components U_cd , U_cq of the output capacitor voltage dq obtained in step 1 and the components I_od , I_oq of the output current dq, the average active power is obtained through the active power calculation equation and the reactive power calculation equation and average reactive power

步骤4，根据步骤3中得到的平均无功功率和微网逆变器给定的无功功率指令Q_ref、电压指令U_ref，经过无功控制方程得到微网逆变器电容电压dq的分量基准信号U_dref,U_qref；Step 4, according to the average reactive power obtained in step 3 and the reactive power command Q_ref and voltage command U_ref given by the microgrid inverter, through the reactive power control equation, the component reference signals U_dref and U_qref of the capacitor voltage dq of the microgrid inverter are obtained;

步骤5，根据步骤2得到的电容电压dq的分量指令信号增量ΔU_dref,ΔU_qref和步骤4中得到的电容电压dq的分量基准信号U_dref,U_qref，将两者分别相加，得到电容电压dq的分量指令信号Step 5, according to the component command signal increment ΔU_dref and ΔU_qref of the capacitor voltage dq obtained in step 2 and the component reference signal U_dref and U_qref of the capacitor voltage dq obtained in step 4, add the two respectively to obtain the capacitance Component command signal of voltage dq

步骤6，先根据步骤5得到的电容电压dq的分量指令信号以及步骤1中的输出电容电压dq的分量U_cd,U_cq，通过电压控制方程得到电容电流dq的分量指令信号再根据电容电流dq的分量指令信号和步骤1中的桥臂侧电感电流dq的分量I_ld,I_lq和输出电流dq的分量I_od,I_oq，通过电流控制方程得到控制信号U_d1,U_q1；Step 6, first according to the component command signal of the capacitor voltage dq obtained in step 5 And the components U_cd , U_cq of the output capacitor voltage dq in step 1, the component command signal of the capacitor current dq is obtained through the voltage control equation Then according to the component command signal of the capacitor current dq And the components I_ld , I_lq of the bridge arm side inductor current dq in step 1 and the components I_od , I_oq of the output current dq, the control signals U_d1 , U_q1 are obtained through the current control equation;

步骤7，根据步骤1得到的输出电容电压dq的分量U_cd,U_cq，经过电容电压复合微分控制方程得到控制信号U_d2,U_q2；Step 7, according to the components U_cd , U_cq of the output capacitor voltage dq obtained in step 1, the control signals U_d2 , U_q2 are obtained through the compound differential control equation of the capacitor voltage;

步骤8，根据步骤1中得到的输出电流dq的分量I_od,I_oq，经过动态输出阻抗控制方程得到控制信号U_d3,U_q3；Step 8, according to the components I_od , I_oq of the output current dq obtained in Step 1, the control signals U_d3 , U_q3 are obtained through the dynamic output impedance control equation;

步骤9，根据步骤6中的控制信号U_d1,U_q1、步骤7中的控制信号U_d2,U_q2和步骤8中的控制信号U_d3,U_q3，将三者分别相加，得到控制信号U_d,U_q；Step 9, according to the control signals U_d1 , U_q1 in step 6, the control signals U_d2 , U_q2 in step 7, and the control signals U_d3 , U_q3 in step 8, add the three respectively to obtain the control signal U_d , U_q ;

步骤10，根据步骤3中得到的平均有功功率和微网逆变器给定的有功功率指令P_ref、微网逆变器给定的角频率指令ω_ref，经过功角控制方程得到虚拟同步机的角频率ω，对角频率ω积分得到虚拟同步机的矢量角θ；Step 10, according to the average active power obtained in step 3 and the active power command P_ref given by the microgrid inverter, and the angular frequency command ω_ref given by the microgrid inverter, the angular frequency ω of the virtual synchronous machine is obtained through the power angle control equation, and the virtual angular frequency ω is integrated to obtain the virtual The vector angle θ of the synchronous machine;

步骤11，先根据步骤9中的控制信号U_d,U_q和步骤10中得到的矢量角θ，经过单同步旋转坐标反变换得到三相桥臂电压控制信号U_a,U_b,U_c，再根据三相桥臂电压控制信号U_a,U_b,U_c生成微网逆变器逆变桥开关管的PWM控制信号。Step 11: First, according to the control signals U_d , U_q in step 9 and the vector angle θ obtained in step 10, the three-phase bridge arm voltage control signals U_a , U_b , U_c are obtained through inverse transformation of single synchronous rotating coordinates, Then according to the three-phase bridge arm voltage control signals U_a , U_b , U_c generate the PWM control signal of the switch tube of the inverter bridge of the microgrid inverter.

作为基于虚拟同步机的自适应输出阻抗控制方法的进一步改进：As a further improvement of the adaptive output impedance control method based on the virtual synchronous machine:

优选地，步骤2中的稳态输出阻抗控制方程为Preferably, the steady-state output impedance control equation in step 2 is

$\begin{matrix} {ΔU Δ U}_{d d r r e e f f} = = {K K}_{11} {ω ω}_{00} {LI LI}_{o o d d} \\ {ΔU Δ U}_{q q r r e e f f} = = - - {K K}_{11} {ω ω}_{00} {LI LI}_{o o q q} \end{matrix},,$

其中，K₁为补偿系数、ω₀为基波角频率、L为微网逆变器桥臂侧电感值。Among them, K₁ is the compensation coefficient, ω₀ is the fundamental angular frequency, and L is the inductance value of the bridge arm side of the microgrid inverter.

优选地，步骤3中的有功功率计算方程为Preferably, the active power calculation equation in step 3 is

$\overset{&OverBar; &OverBar;}{P P} = = ((\underset{h h}{Π Π} \frac{{s the s}^{22} + + {ω ω}_{h h}^{22}}{{s the s}^{22} + + 22 {Qω Qω}_{h h} s the s + + {ω ω}_{h h}^{22}})) \cdot &Center Dot; \frac{1.5 1.5}{τ τ s the s + + 11} \cdot \cdot (({U u}_{c c q q} {I I}_{o o q q} + + {U u}_{c c d d} {I I}_{o o d d})),,$

其中，ω_h为陷波器需要滤除的谐波角频率、τ为一阶低通滤波器的时间常数、s为拉普拉斯算子、Q为谐振控制器品质因数。Among them, ω_h is the harmonic angular frequency that needs to be filtered out by the notch filter, τ is the time constant of the first-order low-pass filter, s is the Laplacian operator, and Q is the quality factor of the resonance controller.

优选地，步骤3中的无功功率计算方程为Preferably, the reactive power calculation equation in step 3 is

$\overset{&OverBar; &OverBar;}{Q Q} = = ((\underset{h h}{Π Π} \frac{{s the s}^{22} + + {ω ω}_{h h}^{22}}{{s the s}^{22} + + 22 {Qω Qω}_{h h} s the s + + {ω ω}_{h h}^{22}})) \cdot &Center Dot; \frac{1.5 1.5}{τ τ s the s + + 11} \cdot &Center Dot; (({U u}_{c c d d} {I I}_{o o q q} - - {U u}_{c c q q} {I I}_{o o d d})),,$

优选地，步骤4中的无功控制方程为Preferably, the reactive control equation in step 4 is

$\begin{matrix} {U u}_{d d r r e e f f} = = {U u}_{r r e e f f} + + n no (({Q Q}_{r r e e f f} - - \overset{&OverBar; &OverBar;}{Q Q})) \\ {U u}_{q q r r e e f f} = = 00 \end{matrix},,$

其中，U_ref为微网逆变器给定无功功率指令Q_ref时的额定输出电容电压、n为无功控制下垂系数。Among them, U_ref is the rated output capacitor voltage when the microgrid inverter is given the reactive power command Q_ref , and n is the reactive power control droop coefficient.

优选地，步骤6中的电压控制方程为Preferably, the voltage control equation in step 6 is

$\begin{matrix} {I I}_{c c d d}^{* *} = = (({K K}_{p p} + + {K K}_{i i} / / s the s)) (({U u}_{c c d d}^{* *} - - {U u}_{c c d d})) \\ {I I}_{c c q q}^{* *} = = (({K K}_{p p} + + {K K}_{i i} / / s the s)) ((00 - - {U u}_{c c q q})) \end{matrix},,$

其中，K_p为比例控制系数、K_i为积分控制系数、s为拉普拉斯算子。Among them, K_p is the proportional control coefficient, K_i is the integral control coefficient, and s is the Laplacian operator.

优选地，步骤6中的电流控制方程为Preferably, the current control equation in step 6 is

$\begin{matrix} {U u}_{d d 11} = = K K (({I I}_{c c d d}^{* *} - - {I I}_{l l d d} + + {I I}_{o o d d})) \\ {U u}_{q q 11} = = K K (({I I}_{c c q q}^{* *} - - {I I}_{l l q q} + + {I I}_{o o q q})) \end{matrix},,$

其中，K为比例控制系数。Among them, K is the proportional control coefficient.

优选地，步骤7中的电容电压复合微分控制方程为Preferably, the capacitor voltage composite differential control equation in step 7 is

$\begin{matrix} {U u}_{d d 22} = = {K K}_{22} s the s C C (({λT λT}_{s the s} s the s + + 11)) {U u}_{c c d d} \\ {U u}_{q q 22} = = {K K}_{22} s the s C C (({λT λ T}_{s the s} s the s + + 11)) {U u}_{c c q q} \end{matrix},,$

其中，K₂为补偿系数、C为微网逆变器滤波电容值、λ为采样延时时间常数、T_s为微网逆变器采样频率、s为拉普拉斯算子。Among them, K₂ is the compensation coefficient, C is the filter capacitance value of the microgrid inverter, λ is the sampling delay time constant, T_s is the sampling frequency of the microgrid inverter, and s is the Laplacian operator.

优选地，步骤8中的动态输出阻抗控制方程为Preferably, the dynamic output impedance control equation in step 8 is

$\begin{matrix} {U u}_{d d 33} = = {K K}_{33} s the s L L (({λT λT}_{s the s} s the s + + 11)) {I I}_{o o d d} \\ {U u}_{q q 33} = = {K K}_{33} s the s L L (({λT λT}_{s the s} s the s + + 11)) {I I}_{o o q q} \end{matrix},,$

其中，K₃为补偿系数、L为微网逆变器桥臂侧电感值、λ为采样延时时间常数、T_s为微网逆变器采样频率、s为拉普拉斯算子。Among them, K₃ is the compensation coefficient, L is the inductance value of the bridge arm side of the microgrid inverter, λ is the sampling delay time constant, T_s is the sampling frequency of the microgrid inverter, and s is the Laplacian operator.

优选地，步骤10中的功角控制方程为Preferably, the power angle control equation in step 10 is

$ω ω = = {ω ω}_{r r e e f f} + + \frac{m m}{{Jω Jω}_{00} m m s the s + + 11} (({P P}_{r r e e f f} - - \overset{&OverBar; &OverBar;}{P P})),,$

其中，ω_ref为微网逆变器给定有功功率指令P_ref时的额定角频率、J为虚拟同步机的虚拟转动惯量时间常数、ω₀为电网固定角频率、m为功角控制下垂系数。Among them, ω_ref is the rated angular frequency of the microgrid inverter when the active power command P_ref is given, J is the virtual moment of inertia time constant of the virtual synchronous machine, ω₀ is the fixed angular frequency of the power grid, and m is the power angle control droop coefficient .

相对于现有技术的有益效果是：The beneficial effects relative to the prior art are:

采用本发明后，基于虚拟同步机的微网逆变器、UPS等装置离网并联运行时，在其同一输出阻抗既能满足输出电压波形动静态性能，又能满足功率均分的基础上，具备了如下优点：After adopting the present invention, when devices such as micro-grid inverters and UPSs based on virtual synchronous machines are operated in parallel off-grid, on the basis that the same output impedance can satisfy both the dynamic and static performance of the output voltage waveform and the power equalization, It has the following advantages:

1.增强了基于VSG的微网逆变器、UPS等装置运行时的稳定性，尤为开关频率较低、控制延时较大的大功率应用场合。1. Enhanced the stability of VSG-based micro-grid inverters, UPS and other devices during operation, especially in high-power applications with low switching frequency and large control delay.

2.动稳态输出阻抗实现了解耦控制与设计，稳态输出阻抗保证良好的功率均分，动态输出阻抗保证负载突变等的输出电压动态波形质量，且不相互影响。2. The dynamic steady-state output impedance realizes decoupling control and design, the steady-state output impedance ensures good power sharing, and the dynamic output impedance ensures the quality of the dynamic waveform of the output voltage such as sudden load changes without mutual influence.

3.易于应用于采用VSG的三相微网逆变器、三相UPS装置中，同时也易于应用于基于dq坐标系控制系统的输出阻抗设计方法。3. It is easy to apply to three-phase micro-grid inverters and three-phase UPS devices using VSG, and is also easy to apply to the output impedance design method based on the dq coordinate system control system.

4.有功和无功功率计算方法的动态响应较快，谐波含量较少。4. The dynamic response of the active and reactive power calculation method is faster, and the harmonic content is less.

附图说明Description of drawings

图1是本发明的总体控制框图。Fig. 1 is the overall control block diagram of the present invention.

图2是本发明所采用的虚拟同步机的拓扑结构图。Fig. 2 is a topological structure diagram of the virtual synchronization machine adopted in the present invention.

图3是虚拟同步机采用本发明前、后的仿真波形对比图。其中，图3a为采用本发明前的仿真波形图，图3b为采用本发明后的仿真波形图。Fig. 3 is a comparison diagram of simulation waveforms before and after the virtual synchronous machine adopts the present invention. Among them, Fig. 3a is a simulation waveform diagram before adopting the present invention, and Fig. 3b is a simulation waveform diagram after adopting the present invention.

图4是虚拟同步机采用本发明前、后的实验波形对比图。其中，图4a为采用本发明前的实验波形图，图4b为采用本发明后的实验波形图。Fig. 4 is a comparison diagram of experimental waveforms before and after the virtual synchronous machine adopts the present invention. Among them, Fig. 4a is the experimental waveform diagram before adopting the present invention, and Fig. 4b is the experimental waveform diagram after adopting the present invention.

由图3和图4可看出，本发明在抑制了系统振荡的基础上，有效地提高了其动态响应。It can be seen from Fig. 3 and Fig. 4 that the present invention effectively improves the dynamic response on the basis of suppressing system oscillation.

具体实施方式detailed description

下面结合附图对本发明的优选方式作进一步详细的描述。The preferred modes of the present invention will be further described in detail below in conjunction with the accompanying drawings.

本发明实施时的有关电气参数设置如下：The relevant electrical parameters when the present invention is implemented are set as follows:

直流母线电压Udc为550V，输出交流线电压有效值为380V/50Hz，额定容量为100KW，微网逆变器桥臂侧电感为L＝0.5mH，微网逆变器滤波电容为C＝200μF。变压器为100KVA270/400V Dyn11型变压器。The DC bus voltage Udc is 550V, the effective value of the output AC line voltage is 380V/50Hz, the rated capacity is 100KW, the inductance of the bridge arm side of the microgrid inverter is L=0.5mH, and the filter capacitance of the microgrid inverter is C=200μF. The transformer is a 100KVA270/400V Dyn11 type transformer.

参见图1、图2、图3和图4，本发明的实施过程如下：Referring to Fig. 1, Fig. 2, Fig. 3 and Fig. 4, the implementation process of the present invention is as follows:

步骤1，先采集微网逆变器的输出电容电压U_ca,U_cb,U_cc、桥臂侧电感电流I_la,I_lb,I_lc和输出电流I_oa,I_ob,I_oc，再经过单同步旋转坐标变换得到输出电容电压dq的分量U_cd,U_cq、桥臂侧电感电流dq的分量I_ld,I_lq和输出电流dq的分量I_od,I_oq。Step 1, first collect the output capacitor voltage U_ca , U_cb , U_cc of the microgrid inverter, the bridge arm side inductor current I_la , I_lb , I_lc and the output current I_oa , I_ob , I_oc , and then pass The single synchronous rotation coordinate transformation obtains the components U_cd , U_cq of the output capacitor voltage dq, the components I_ld , I_lq of the bridge arm side inductor current dq and the components I_od , I_oq of the output current dq.

步骤2，根据步骤1中得到的输出电流dq的分量I_od,I_oq，经过稳态输出阻抗控制方程得到电容电压dq的分量指令信号增量ΔU_dref,ΔU_qref；其中，Step 2, according to the components I_od , I_oq of the output current dq obtained in step 1, through the steady-state output impedance control equation to obtain the component command signal increment ΔU_dref , ΔU_qref of the capacitor voltage dq; where,

稳态输出阻抗控制方程为The steady-state output impedance governing equation is

$\begin{matrix} {ΔU Δ U}_{q q r r e e f f} = = {K K}_{11} {ω ω}_{00} {LI LI}_{o o d d} \\ {ΔU Δ U}_{q q r r e e f f} = = - - {K K}_{11} {ω ω}_{00} {LI LI}_{o o q q} \end{matrix},,$

其中的K₁为补偿系数、ω₀为基波角频率、L为微网逆变器桥臂侧电感值。Among them, K₁ is the compensation coefficient, ω₀ is the fundamental angular frequency, and L is the inductance value of the bridge arm side of the microgrid inverter.

补偿系数K₁主要考虑最终微网逆变器的电容电压降落，一般在≤10％的范围内，因此K₁一般取值0.1≤K₁≤1；在本实施例中，取K₁＝0.5。The compensation coefficient K₁ mainly considers the capacitor voltage drop of the final microgrid inverter, which is generally within the range of ≤10%, so K₁ generally takes a value of 0.1≤K₁ ≤1; in this embodiment, K₁ =0.5 .

步骤3，根据步骤1中得到的输出电容电压dq的分量U_cd,U_cq和输出电流dq的分量I_od,I_oq，经过有功功率计算方程和无功功率计算方程得到平均有功功率和平均无功功率其中，Step 3, according to the components U_cd , U_cq of the output capacitor voltage dq obtained in step 1 and the components I_od , I_oq of the output current dq, the average active power is obtained through the active power calculation equation and the reactive power calculation equation and average reactive power in,

有功功率计算方程为The active power calculation equation is

其中的ω_h为陷波器需要滤除的谐波角频率、τ为一阶低通滤波器的时间常数、s为拉普拉斯算子、Q为谐振控制器品质因数，Among them, ω_h is the harmonic angular frequency that needs to be filtered out by the notch filter, τ is the time constant of the first-order low-pass filter, s is the Laplacian operator, and Q is the quality factor of the resonance controller.

无功功率计算方程为The reactive power calculation equation is

$\overset{&OverBar; &OverBar;}{Q Q} = = ((\underset{h h}{Π Π} \frac{{s the s}^{22} + + {ω ω}_{h h}^{22}}{{s the s}^{22} + + 22 {Qω Qω}_{h h} s the s + + {ω ω}_{h h}^{22}})) \cdot \cdot \frac{1.5 1.5}{τ τ s the s + + 11} \cdot \cdot (({U u}_{c c d d} {I I}_{o o q q} - - {U u}_{c c q q} {I I}_{o o d d})),,$

其中的ω_h为陷波器需要滤除的谐波角频率、τ为一阶低通滤波器的时间常数、s为拉普拉斯算子、Q为谐振控制器品质因数。Among them, ω_h is the harmonic angular frequency that needs to be filtered out by the notch filter, τ is the time constant of the first-order low-pass filter, s is the Laplacian operator, and Q is the quality factor of the resonance controller.

在本实施例中，考虑主要滤除的谐波次数为2次和3次谐波，因此选取h＝2,3，此时ω_h＝628.3186rad/s,942.4779rad/s。一阶低通滤波器主要考虑滤除高次谐波，且不影响动态响应，一般取τ≤2e^-3s，本例取值τ＝1.5e^-4s；品质因数Q主要考虑陷波器的滤波效果，在本例中，选取Q＝0.5。In this embodiment, it is considered that the mainly filtered out harmonics are the 2nd and 3rd harmonics, so h=2,3 is selected, and ω_h =628.3186rad/s, 942.4779rad/s at this time. The first-order low-pass filter mainly considers filtering out high-order harmonics without affecting the dynamic response. Generally, τ≤2e^-3 s is taken, and the value in this example is τ=1.5e^-4 s; the quality factor Q mainly considers the notch filter The filtering effect of , in this example, select Q=0.5.

步骤4，根据步骤3中得到的平均无功功率和微网逆变器给定的无功功率指令Q_ref、电压指令U_ref，经过无功控制方程得到微网逆变器电容电压dq的分量基准信号U_dref,U_qref；其中，Step 4, according to the average reactive power obtained in step 3 and the reactive power command Q_ref and voltage command U_ref given by the microgrid inverter, through the reactive power control equation, the component reference signals U_dref , U_qref of the capacitor voltage dq of the microgrid inverter are obtained; where,

无功控制方程为The reactive control equation is

其中的U_ref为微网逆变器给定无功功率指令Q_ref时的额定输出电容电压、n为无功控制下垂系数。Among them, U_ref is the rated output capacitor voltage when the microgrid inverter is given the reactive power command Q_ref , and n is the reactive power control droop coefficient.

无功控制下垂系数n取值原则为100％的无功功率变化时，电压幅值变化在2％之内；给定无功功率指令Q_ref和相对应的额定输出电容电压U_ref表示下垂曲线的位置关系，主要考虑微网逆变器输出无功功率为Q_ref时，其输出电压大小。The value principle of the reactive power control droop coefficient n is that when the reactive power changes by 100%, the voltage amplitude changes within 2%; the given reactive power command Q_ref and the corresponding rated output capacitor voltage U_ref represent the droop curve The positional relationship of the microgrid inverter mainly considers the output voltage of the microgrid inverter when the output reactive power is_Qref .

在本实施例中，无功控制下垂系数取值为给定无功功率指令Q_ref考虑系统输出无功功率为Q_ref＝0，此时对应的额定输出电容电压U_ref＝380V。In this embodiment, the value of reactive power control droop coefficient is The given reactive power command Q_ref considers that the system output reactive power is Q_ref =0, and the corresponding rated output capacitor voltage U_ref =380V at this time.

步骤6，先根据步骤5得到的电容电压dq的分量指令信号以及步骤1中的输出电容电压dq的分量U_cd,U_cq，通过电压控制方程得到电容电流dq的分量指令信号其中，Step 6, first according to the component command signal of the capacitor voltage dq obtained in step 5 And the components U_cd , U_cq of the output capacitor voltage dq in step 1, the component command signal of the capacitor current dq is obtained through the voltage control equation in,

电压控制方程为The voltage governing equation is

其中的K_p为比例控制系数、K_i为积分控制系数、s为拉普拉斯算子。Among them, K_p is the proportional control coefficient, K_i is the integral control coefficient, and s is the Laplacian operator.

再根据电容电流dq的分量指令信号和步骤1中的桥臂侧电感电流dq的分量I_ld,I_lq和输出电流dq的分量I_od,I_oq，通过电流控制方程得到控制信号U_d1,U_q1；其中，Then according to the component command signal of the capacitor current dq And the components I_ld , I_lq of the bridge arm side inductor current dq in step 1 and the components I_od , I_oq of the output current dq, the control signals U_d1 , U_q1 are obtained through the current control equation; where,

电流控制方程为The current governing equation is

其中的K为比例控制系数。Among them, K is the proportional control coefficient.

电压和电流控制方程中的参数主要考虑控制系统的稳定性和动稳态性能；在本实施例中，取K_p＝0.03,K_i＝0.8,K＝0.05。The parameters in the voltage and current control equations mainly consider the stability and dynamic steady-state performance of the control system; in this embodiment, K_p =0.03, K_i =0.8, K=0.05.

步骤7，根据步骤1得到的输出电容电压dq的分量U_cd,U_cq，经过电容电压复合微分控制方程得到控制信号U_d2,U_q2；其中，Step 7, according to the components U_cd , U_cq of the output capacitor voltage dq obtained in step 1, the control signals U_d2 , U_q2 are obtained through the compound differential control equation of the capacitor voltage; where,

电容电压复合微分控制方程为The compound differential control equation of capacitor voltage is

其中的K₂为补偿系数、C为微网逆变器滤波电容值、λ为采样延时时间常数、T_s为微网逆变器采样频率、s为拉普拉斯算子。Among them, K₂ is the compensation coefficient, C is the filter capacitance value of the microgrid inverter, λ is the sampling delay time constant, T_s is the sampling frequency of the microgrid inverter, and s is the Laplacian operator.

补偿系数K₂相当于电阻阻抗的数值，在本实施例中可以取K₂＝1；采样延时时间λ根据数字处理和控制器的处理方法不同而不同。一般数字处理器采用三角载波的方式进行PWM调制。当采用此种方式的时候，若在三角波的过零处和峰值处进行步骤1中的电压和电流采样，此时系统延时最大λ＝1.5；一般情况下，0.5＜λ≤1.5；在本实施例中，由于电压和电流采样均在三角波的过零点和峰值处进行，因而取λ＝1.5。The compensation coefficient K₂ is equivalent to the value of the resistance impedance, and in this embodiment, K₂ =1; the sampling delay time λ varies according to the digital processing and the processing method of the controller. General digital processors use triangular carrier waves for PWM modulation. When this method is used, if the voltage and current sampling in step 1 is carried out at the zero-crossing point and peak point of the triangular wave, the maximum delay of the system at this time is λ=1.5; in general, 0.5<λ≤1.5; In the embodiment, since both voltage and current sampling are performed at the zero-crossing point and peak value of the triangular wave, λ=1.5 is taken.

步骤8，根据步骤1中得到的输出电流dq的分量I_od,I_oq，经过动态输出阻抗控制方程得到控制信号U_d3,U_q3；其中，Step 8, according to the components I_od , I_oq of the output current dq obtained in step 1, the control signals U_d3 , U_q3 are obtained through the dynamic output impedance control equation; where,

动态输出阻抗控制方程为The dynamic output impedance governing equation is

$\begin{matrix} {U u}_{d d 33} = = {K K}_{33} s the s L L (({λT λ T}_{s the s} s the s + + 11)) {I I}_{o o d d} \\ {U u}_{q q 33} = = {K K}_{33} s the s L L (({λT λ T}_{s the s} s the s + + 11)) {I I}_{o o q q} \end{matrix},,$

其中的K₃为补偿系数、L为微网逆变器桥臂侧电感值、λ为采样延时时间常数、T_s为微网逆变器采样频率、s为拉普拉斯算子。Among them, K₃ is the compensation coefficient, L is the inductance value of the bridge arm side of the microgrid inverter, λ is the sampling delay time constant, T_s is the sampling frequency of the microgrid inverter, and s is the Laplacian operator.

补偿系数K₃主要考虑动态输出阻抗补偿的有效性。一般取值0.5≤K₃≤1；在本实施例中，取K₃＝1,λ＝1.5。_The compensation coefficient K3 mainly considers the effectiveness of dynamic output impedance compensation. Generally, the value is 0.5≦K₃ ≦1; in this embodiment, K₃ =1, λ=1.5.

步骤9，根据步骤6中的控制信号U_d1,U_q1、步骤7中的控制信号U_d2,U_q2和步骤8中的控制信号U_d3,U_q3，将三者分别相加，得到控制信号U_d,U_q。Step 9, according to the control signals U_d1 , U_q1 in step 6, the control signals U_d2 , U_q2 in step 7, and the control signals U_d3 , U_q3 in step 8, add the three respectively to obtain the control signal U_d , U_q .

步骤10，根据步骤3中得到的平均有功功率和微网逆变器给定的有功功率指令P_ref、微网逆变器给定的角频率指令ω_ref，经过功角控制方程得到虚拟同步机的角频率ω，对角频率ω积分得到虚拟同步机的矢量角θ；其中，Step 10, according to the average active power obtained in step 3 and the active power command P_ref given by the microgrid inverter, and the angular frequency command ω_ref given by the microgrid inverter, the angular frequency ω of the virtual synchronous machine is obtained through the power angle control equation, and the virtual angular frequency ω is integrated to obtain the virtual The vector angle θ of the synchronous machine; where,

功角控制方程为The power angle governing equation is

其中的ω_ref为微网逆变器给定有功功率指令P_ref时的额定角频率、J为虚拟同步机的虚拟转动惯量时间常数、ω₀为电网固定角频率、m为功角控制下垂系数。Among them,_{ωref is the rated angular frequency when the microgrid inverter is given the active power command P ref}_, J is the virtual moment of inertia time constant of the virtual synchronous machine,_ω0 is the fixed angular frequency of the power grid, and m is the power angle control droop coefficient .

功角控制方程表明了微网逆变器有功功率下垂曲线关系和虚拟惯量大小。其中，虚拟惯量标明了系统频率的变化率，为了保证系统频率变化平稳，需要有较大的虚拟惯量；然而虚拟惯量相当于在系统中加入了一阶惯性环节，太大的虚拟惯量有可能导致系统的不稳定。因而参数选择需要折中处理。为保证系统稳定性，在本实施例中，惯性时间常数范围在τ_virtual＝Jω₀m≤2e^-3s；功角控制方程中的有功功率下垂曲线关系包括三个系数，功角控制下垂系数m表示下垂曲线的斜率，取值原则为100％的有功功率变化时，频率变化0.5Hz以内；给定有功功率指令P_ref和相对应的额定角频率ω_ref表示下垂曲线的位置关系，主要考虑微网逆变器输出有功功率为P_ref时，其输出频率大小。The control equation of the power angle shows the drooping curve relationship of the active power of the microgrid inverter and the magnitude of the virtual inertia. Among them, the virtual inertia indicates the change rate of the system frequency. In order to ensure the smooth change of the system frequency, a large virtual inertia is required; however, the virtual inertia is equivalent to adding a first-order inertia link to the system, and too large virtual inertia may cause System instability. Therefore, parameter selection requires a compromise. In order to ensure system stability, in this embodiment, the inertia time constant range is τ_virtual = Jω₀ m≤2e^-3 s; the active power droop curve relationship in the power angle control equation includes three coefficients, and the power angle control droop coefficient m represents the slope of the droop curve, and the value principle is that when the active power changes by 100%, the frequency changes within 0.5Hz; the given active power command P_ref and the corresponding rated angular frequency ω_ref represent the position relationship of the droop curve, mainly considered When the output active power of the microgrid inverter is_Pref , its output frequency.

在本实施例中，电网角频率采用额定频率为50Hz时对应的角频率，即ω₀＝314.1593rad/s，功角控制下垂系数取值为根据惯性时间常数取值原则取τ_virtual＝Jω₀m＝1.5e^-3s，可得J＝0.2Kg·m²,为保证控制运行时能量不流向直流侧，给定有功功率指令取值为P_ref＝1KW，此时对应的额定角频率取值为ω_ref＝314.1593rad/s。In this embodiment, the grid angular frequency adopts the corresponding angular frequency when the rated frequency is 50Hz, that is, ω₀ =314.1593rad/s, and the power angle control droop coefficient is According to the value principle of the inertia time constant, take τ_virtual = Jω₀ m = 1.5e^-3 s to get J = 0.2Kg·m² , in order to ensure that the energy does not flow to the DC side during the control operation, the given active power command value is P_ref =1KW, and the corresponding rated angular frequency at this time is ω_ref =314.1593rad/s.

显然，本领域的技术人员可以对本发明的基于虚拟同步机的自适应输出阻抗控制方法进行各种改动和变型而不脱离本发明的精神和范围。这样，倘若对本发明的这些修改和变型属于本发明权利要求及其等同技术的范围之内，则本发明也意图包含这些改动和变型在内。Apparently, those skilled in the art can make various changes and modifications to the method for adaptive output impedance control based on a virtual synchronous machine of the present invention without departing from the spirit and scope of the present invention. Thus, if these modifications and variations of the present invention fall within the scope of the claims of the present invention and equivalent technologies, the present invention also intends to include these modifications and variations.