CN104215979A

Movatterモバイル変換

Info

Publication number: CN104215979A
Application number: CN201310206933.1A
Authority: CN
Inventors: 张延东; 朱柏承; 孟凡琛; 甘哲
Original assignee: Peking University
Current assignee: Peking University
Priority date: 2013-05-29
Filing date: 2013-05-29
Publication date: 2014-12-17

Abstract

Translated fromChinese

本发明公开了一种基于分段相关结合FFT运算的大频偏GNSS信号捕获方法。该方法通过分段相关结合FFT运算，可在码相位串行搜索的同时完成载波频率的并行搜索，极大地缩短了大频偏GNSS信号的捕获时间，且信噪比损失较小。具体来说，该方法首先将接收信号分成若干子段，分别与本地伪码做相关运算得到若干相关值，然后将相关值序列补零后进行FFT运算，最后对运算结果取模后的最大值进行判决，若该值高于判决门限，则此时的码相位为捕获所得码相位，该值对应的频点为捕获所得载波频率，若该值低于判决门限则调整本地码相位继续该过程，直到完成大频偏GNSS信号的时频二维搜索。

The invention discloses a large frequency offset GNSS signal acquisition method based on segment correlation combined with FFT operation. This method combines segmental correlation with FFT operation to complete the parallel search of the carrier frequency while the code phase is serially searched, which greatly shortens the acquisition time of GNSS signals with large frequency offsets, and the signal-to-noise ratio loss is small. Specifically, this method first divides the received signal into several sub-segments, performs correlation operations with the local pseudo code to obtain several correlation values, then fills the correlation value sequence with zeros and performs FFT operations, and finally moduloes the operation results to obtain the maximum value Make a decision. If the value is higher than the decision threshold, the code phase at this time is the captured code phase. The frequency point corresponding to this value is the captured carrier frequency. If the value is lower than the decision threshold, adjust the local code phase and continue the process. , until the time-frequency two-dimensional search of GNSS signals with large frequency offset is completed.

Description

Translated fromChinese

技术领域technical field

本发明属于全球卫星导航系统接收机扩频信号处理领域，具体的说是一种基于分段相关结合FFT运算的载波频率捕获方法。The invention belongs to the field of spread spectrum signal processing of a global satellite navigation system receiver, in particular to a carrier frequency acquisition method based on segment correlation combined with FFT operation.

背景技术Background technique

全球导航卫星系统(Global Navigation Satellite System，GNSS)可为用户提供连续性的位置、速度和时间信息，可满足导航、定位、测速、授时和救援等多种服务要求。目前全球主要有四大卫星导航系统：美国的GPS系统、俄罗斯的GLONASS系统、欧洲的Galileo系统和中国的北斗系统。这些系统均以码分多址或频分多址的方式向全球用户全天候的广播卫星信号。Global Navigation Satellite System (GNSS) can provide users with continuous position, speed and time information, which can meet various service requirements such as navigation, positioning, speed measurement, timing and rescue. At present, there are four major satellite navigation systems in the world: the GPS system of the United States, the GLONASS system of Russia, the Galileo system of Europe and the Beidou system of China. These systems all broadcast satellite signals to users all over the world in the way of code division multiple access or frequency division multiple access.

GNSS扩频信号码的捕获方案基本结构如图1所示，它是各种扩频信号码捕获方案的基础。信号捕获的基本流程为：基带扩频信号C(t)和S(t)经过与本地码相关、积分、平方求和得到检测量，对检测量进行判决，如果检测量超过检测门限则认为捕获，否则调整本地扩频码相位，进行下一个相位的检测。The basic structure of the acquisition scheme of GNSS spread spectrum signal code is shown in Figure 1, which is the basis of various spread spectrum signal code acquisition schemes. The basic process of signal acquisition is: the baseband spread spectrum signals C(t) and S(t) are correlated with the local code, integrated, and squared to obtain the detection quantity, and the detection quantity is judged. If the detection quantity exceeds the detection threshold, it is considered to be captured , otherwise adjust the phase of the local spreading code to detect the next phase.

当接收机以较高速度运行时，上述基带信号会存在较大的多普勒载波频率偏移，此时信号捕获过程中需要搜索的载波频率范围较大。当采用传统的载波频率串行搜索策略时，信号捕获时间较长。因此，有必要寻找一种快速的多普勒频偏估计方法，以实现对大频偏信号载波频率的准确估计。When the receiver runs at a higher speed, the above-mentioned baseband signal will have a larger Doppler carrier frequency offset. At this time, the carrier frequency range to be searched in the signal acquisition process is larger. When using the traditional carrier frequency serial search strategy, the signal acquisition time is longer. Therefore, it is necessary to find a fast Doppler frequency offset estimation method to achieve accurate estimation of the carrier frequency of the large frequency offset signal.

发明内容Contents of the invention

本发明的目的是克服上述背景中的不足之处，提供一种基于分段相关结合FFT运算的大频偏GNSS信号捕获方法。The purpose of the present invention is to overcome the disadvantages in the above-mentioned background, and provide a large frequency offset GNSS signal acquisition method based on segment correlation combined with FFT operation.

该方法通过分段相关结合FFT运算，可在码相位串行搜索的同时完成载波频率的并行搜索，极大地缩短了大频偏GNSS信号的捕获时间，且信噪比损失较小，其原理框图如图2所示。本发明主要包括以下几个步骤：This method combines segmental correlation with FFT operation to complete the parallel search of the carrier frequency while the code phase is serially searched, which greatly shortens the acquisition time of GNSS signals with large frequency offsets, and the signal-to-noise ratio loss is small. Its principle block diagram as shown in picture 2. The present invention mainly comprises the following steps:

第一步：数字信号采样。The first step: digital signal sampling.

接收信号经过数字下变频后，得到基带信号(含多普勒频偏)为：After the received signal is digitally down-converted, the baseband signal (including Doppler frequency offset) is obtained as:

$r r ((k k)) = = D D. ((k k)) PN PN ((k k)) {e e}^{j j ((22 π π {f f}_{d d} k k {T T}_{s the s} - - θ θ))}$

其中D(k)为二进制调制信息，PN(k)为扩频码序列，f_d为多普勒频偏，T_s为扩频码片周期，θ为载波相位。Among them, D(k) is the binary modulation information, PN(k) is the spread spectrum code sequence, f_d is the Doppler frequency offset, T_s is the spread spectrum chip period, and θ is the carrier phase.

第二步：将长度为M个码片的接收信号与本地伪码分别划分为R个子段，每段长度P＝M/R，然后将对应子段分别进行相关累加运算，在暂时不考虑二进制调制信息影响的情况下，得到相关值：Step 2: Divide the received signal and the local pseudo-code with a length of M chips into R subsections, each section length P=M/R, and then perform correlation and accumulation operations on the corresponding subsections, temporarily disregarding the binary In the case of the influence of modulation information, the correlation value is obtained:

$C C ((i i)) = = {Σ Σ}_{k k = = {((i i - - 11))}^{* *} P P + + 11}^{{i i}^{* *} P P} PN PN ((k k)) PN PN ((k k + + Δ Δ)) {e e}^{j j ((22 π π {f f}_{d d} k k {T T}_{s the s} - - θ θ))}$

$= = {Σ Σ}_{k k = = {((i i - - 11))}^{* *} P P + + 11}^{{i i}^{* *} P P} R R ((Δ Δ)) {e e}^{j j ((22 π π {f f}_{d d} k k {T T}_{s the s} - - θ θ))}$

$= = R R ((Δ Δ)) \cdot &Center Dot; {e e}^{j j [[((22 π π {f f}_{d d} {((i i - - 11))}^{* *} P P + + 11)) {T T}_{s the s} - - θ θ]]} \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}}$

其中i＝1，2，3...R，R(Δ)为扩频码的自相关函数，Δ为接收信号与本地扩频码的相位差。Where i=1, 2, 3...R, R(Δ) is the autocorrelation function of the spread spectrum code, and Δ is the phase difference between the received signal and the local spread spectrum code.

第三步：将相关值序列补零后进行FFT运算。Step 3: Carry out FFT operation after padding the correlation value sequence with zero.

将R个相关值补S-R个0后进行S点的FFT运算，得到S个FFT输出值为：Add S-R 0s to the R correlation values and then perform the FFT operation of S points, and obtain S FFT output values:

${Z Z}_{c c} ((m m)) + + {j j}^{* *} {Z Z}_{s the s} ((m m)) = = {Σ Σ}_{i i = = 11}^{S S} C C ((i i)) {e e}^{- - j j \frac{22 π π}{S S} im im} = = {Σ Σ}_{i i = = 11}^{R R} C C ((i i)) {e e}^{- - j j \frac{22 π π}{S S} im im}$

$= = {Σ Σ}_{i i = = 11}^{R R} R R ((Δ Δ)) \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot &Center Dot; {e e}^{{j j}^{{{22 π π {f f}_{d d} [[((i i - - 11)) P P + + 11]] {T T}_{s the s} - - \frac{22 π π}{S S} im im - - θ θ}}}}$

$= = R R ((Δ Δ)) \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot \cdot {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot \cdot \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}$

$= = R R ((Δ Δ)) \cdot \cdot {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot \cdot \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot &Center Dot; \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}$

第四步：对FFT运算输出结果取模。Step 4: Take the modulus of the output result of the FFT operation.

对FFT输出值取模，得到捕获检测量：Take the modulo of the FFT output value to obtain the capture detection quantity:

$Z Z ((m m)) = = | | {Z Z}^{22}_{c c} ((m m)) + + {Z Z}^{22}_{s the s} ((m m)) | | = = {| | R R ((Δ Δ)) \cdot &Center Dot; {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot &Center Dot; \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}} | |}^{22}$

$= = {| | R R ((Δ Δ)) \cdot \cdot \frac{sin sin ((π π {f f}_{d d} P P {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} \cdot &Center Dot; \frac{sin sin [[πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

$= = R R {((Δ Δ))}^{22} \cdot \cdot {| | \frac{sin sin ((\frac{M m}{R R} π π {f f}_{d d} {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} | |}^{22} \cdot \cdot {| | \frac{sin sin [[πR πR (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

第五步：对最大模值进行判决。Step 5: Make a judgment on the maximum modulus value.

当接收信号扩频码相位与本地扩频码相位未同步时，R(Δ)≈0，最大模值不超过门限值，这时需调整本地码相位，继续执行步骤二到步骤五。When the phase of the spread spectrum code of the received signal is not synchronized with the phase of the local spread spectrum code, R(Δ)≈0, and the maximum modulus value does not exceed the threshold value. At this time, the phase of the local code needs to be adjusted, and continue to perform steps 2 to 5.

当接收信号扩频码相位与本地扩频码相位基本同步时，R(Δ)≈1，捕获检测量如下：When the phase of the spread code of the received signal is basically synchronized with the phase of the local spread code, R(Δ)≈1, and the capture detection quantity is as follows:

$Z Z ((m m)) = = {| | \frac{sin sin ((\frac{M m}{R R} π π {f f}_{d d} {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} | |}^{22} \cdot &Center Dot; {| | \frac{sin sin [[πR πR (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

其中，第一项为积分段内频偏造成的信噪比损失，只与输入多普勒频差有关，与m值无关；第二项大小随m值变化，当时，对应的使该值达到最大(其中[·]表示取整操作)，此时捕获检测量Z(m)≈M²，捕获过程得到的多普勒频偏估计值Among them, the first item is the SNR loss caused by the frequency offset in the integration segment, which is only related to the input Doppler frequency difference and has nothing to do with the value of m; the size of the second item varies with the value of m, when when, the corresponding Maximize this value (where [ ] means rounding operation), at this time, capture the detected quantity Z(m)≈M² , and the estimated value of Doppler frequency offset obtained during the capture process

当为非整数时，会因相位补偿不完全造成一定的信噪比损失。使用本方法进行信号捕获过程中，信噪比总损失随输入多普勒频偏值的变化如图3所示。when When is a non-integer, it will cause a certain loss of signal-to-noise ratio due to incomplete phase compensation. In the process of signal acquisition using this method, the change of the total loss of signal-to-noise ratio with the input Doppler frequency offset value is shown in Fig. 3 .

附图说明Description of drawings

图1是扩频信号捕获基本结构框图；Fig. 1 is a block diagram of the basic structure of spread spectrum signal acquisition;

图2是本发明分段相关结合FFT运算信号捕获原理框图；Fig. 2 is a block diagram of the present invention's segmentation correlation combined with FFT operation signal acquisition;

图3是本发明分段相关结合FFT运算信号捕获过程中信噪比损失随载波多普勒频偏大小的变化曲线；Fig. 3 is the variation curve of signal-to-noise ratio loss with carrier Doppler frequency deviation size in the present invention's segmentation correlation combined with FFT operation signal capture process;

图4是本发明分段相关结合FFT运算信号捕获相关值分布图；Fig. 4 is the present invention's segmentation correlation combined with FFT operation signal capture correlation value distribution diagram;

具体实施方式Detailed ways

本发明提出的基于分段相关结合FFT运算的大频偏GNSS信号捕获方法，以GPS信号为实例说明如下。The large-frequency-offset GNSS signal acquisition method based on segment correlation combined with FFT operation proposed by the present invention is described as follows by taking GPS signal as an example.

本方法频率估计范围为 $f_{d \max} = &PlusMinus; \frac{1 / T_{s}}{2 (M / R)},$ 频率估计精度为 $f_{d 0} = \frac{2 f_{d \max}}{S},$ 根据需求的载波频偏估计范围和估计精度确定段长度M/R的大小和FFT点数S。The frequency estimation range of this method is $f_{d \max} = &PlusMinus; \frac{1 / T_{the s}}{2 (m / R)},$ The frequency estimation accuracy is $f_{d 0} = \frac{2 f_{d \max}}{S},$ Determine the size of the segment length M/R and the number of FFT points S according to the required carrier frequency offset estimation range and estimation accuracy.

第一步：数字信号采样。The first step: digital signal sampling.

接收GPS信号经过数字下变频后，得到基带信号(含多普勒频偏)为：After the received GPS signal is digitally down-converted, the baseband signal (including Doppler frequency offset) is obtained as:

其中D(k)为GPS电文调制信息；PN(k)为GPS卫星扩频码序列；f_d为多普勒频偏，大小为40KHz；T_s为扩频码片周期，大小为θ为载波相位。Among them, D(k) is the GPS message modulation information; PN(k) is the GPS satellite spread spectrum code sequence; f_d is the Doppler frequency offset, the size is 40KHz; T_s is the spread spectrum chip period, the size is θ is the carrier phase.

第二步：将长度M＝1023个码片的接收信号与本地伪码分别划分为R＝93个子段，每段长度P＝M/R＝11，然后将对应子段分别进行相关累加运算，在暂时不考虑二进制调制信息影响的情况下，得到相关值：Second step: the received signal of the length M=1023 code chips and the local pseudo-code are divided into R=93 subsections respectively, each section length P=M/R=11, and then the corresponding subsections are respectively correlated and accumulated, In the case of temporarily ignoring the influence of binary modulation information, the correlation value is obtained:

$= = R R ((Δ Δ)) \cdot \cdot {e e}^{j j [[((22 π π {f f}_{d d} {((i i - - 11))}^{* *} P P + + 11)) {T T}_{s the s} - - θ θ]]} \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}}$

其中i＝1，2，3...，93，R(Δ)为扩频码的自相关函数，Δ为接收信号与本地扩频码的相位差。Where i=1, 2, 3..., 93, R(Δ) is the autocorrelation function of the spreading code, and Δ is the phase difference between the received signal and the local spreading code.

将R＝93个相关值补S-R＝512-93＝419个0后进行S＝512点的FFT运算，得到S＝512个FFT输出值为：Complementing R=93 correlation values with S-R=512-93=419 0s and then performing the FFT operation of S=512 points to obtain S=512 FFT output values:

$= = {Σ Σ}_{i i = = 11}^{R R} R R ((Δ Δ)) \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot \cdot {e e}^{{j j}^{{{22 π π {f f}_{d d} [[((i i - - 11)) P P + + 11]] {T T}_{s the s} - - \frac{22 π π}{S S} im im - - θ θ}}}}$

$= = R R ((Δ Δ)) \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot &Center Dot; {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot &Center Dot; \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}$

$= = R R ((Δ Δ)) \cdot &Center Dot; {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot &Center Dot; \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}$

$Z Z ((m m)) = = | | {Z Z}^{22}_{c c} ((m m)) + + {Z Z}^{22}_{s the s} ((m m)) | | = = {| | R R ((Δ Δ)) \cdot &Center Dot; {e e}^{j j [[22 π π {f f}_{d d} ((11 - - p p)) {T T}_{s the s} - - θ θ]]} \cdot &Center Dot; \frac{11 - - {e e}^{j j ((22 π π {f f}_{d d} P P {T T}_{s the s}))}}{11 - - {e e}^{j j ((22 π π {f f}_{d d} {T T}_{s the s}))}} \cdot \cdot \frac{11 - - {e e}^{j j 22 πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}}{11 - - {e e}^{j j 22 π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))}} | |}^{22}$

$= = {| | R R ((Δ Δ)) \cdot &Center Dot; \frac{sin sin ((π π {f f}_{d d} P P {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} \cdot &Center Dot; \frac{sin sin [[πR πR (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} P P {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

$= = R R {((Δ Δ))}^{22} \cdot \cdot {| | \frac{sin sin ((\frac{M m}{R R} π π {f f}_{d d} {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} | |}^{22} \cdot &Center Dot; {| | \frac{sin sin [[πR πR (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

$Z Z ((m m)) = = {| | \frac{sin sin ((\frac{M m}{R R} π π {f f}_{d d} {T T}_{s the s}))}{sin sin ((π π {f f}_{d d} {T T}_{s the s}))} | |}^{22} \cdot \cdot {| | \frac{sin sin [[πR πR (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]}{sin sin [[π π (({f f}_{d d} \frac{M m}{R R} {T T}_{s the s} - - \frac{m m}{S S}))]]} | |}^{22}$

$= = {| | \frac{sin sin ((\frac{10231023}{9393} \times \times π π \times \times ((4040 \times \times 1010^{33})) \times \times ((\frac{11}{1.023 1.023} \times \times 1010^{- - 66}))))}{sin sin ((π π \times \times ((4040 \times \times 1010^{33})) \times \times ((\frac{11}{1.023 1.023} \times \times 1010^{- - 66}))))} | |}^{22} \times \times$

${| | \frac{sin sin [[π π \times \times 9393 \times \times ((((4040 \times \times 1010^{33})) \times \times \frac{10231023}{9393} \times \times ((\frac{11}{1.023 1.023} \times \times 1010^{- - 66})) - - \frac{m m}{512512}))]]}{sin sin [[π π \times \times ((((4040 \times \times 1010^{33})) \times \times \frac{10231023}{9393} \times \times ((\frac{11}{1.023 1.023} \times \times 1010^{- - 66})) - - \frac{m m}{512512}))]]} | |}^{22}$

当 $π \times 93 \times ((40 \times 10^{3}) \times \frac{1023}{93} \times (\frac{1}{1.023} \times 10^{- 6}) - \frac{m}{512}) = 0$ 时，该项取最大值，此时对应的m值大小为多普勒频偏估计值为 $f_{d} = \frac{mR}{SM T_{s}} = 39.961 KHz,$ 频率估计误差为40Hz。when $π \times 93 \times ((40 \times 10^{3}) \times \frac{1023}{93} \times (\frac{1}{1.023} \times 10^{- 6}) - \frac{m}{512}) = 0$ When , this item takes the maximum value, and the corresponding m value at this time is The estimated Doppler frequency offset is $f_{d} = \frac{mR}{SM T_{the s}} = 39.961 KHz,$ The frequency estimation error is 40Hz.

Claims

Translated fromChinese

步骤A：数字信号采样； Step A: digital signal sampling;

步骤B：整个码片划分为R个子段，并将对应子段分别进行相关累加运算； Step B: Divide the entire chip into R sub-sections, and perform correlation and accumulation operations on the corresponding sub-sections;

步骤D：对FFT运算输出结果取模； Step D: take the modulus of the output result of the FFT operation;

步骤E：对最大模值进行判决。 Step E: Judging the maximum modulus value. the

2.根据权利要求1所述的大频偏GNSS信号捕获方法，其特征在于：步骤A中提取的基带信号为其中D(k)为二进制调制信息，PN(k)为扩频码序列，f_d为多普勒频偏，T_s为扩频码片周期，θ为载波相位。 2. the large frequency offset GNSS signal acquisition method according to claim 1, is characterized in that: the baseband signal that extracts in the step A is Among them, D(k) is the binary modulation information, PN(k) is the spread spectrum code sequence, f_d is the Doppler frequency offset, T_s is the spread spectrum chip period, and θ is the carrier phase.

3.根据权利要求1所述的大频偏GNSS信号捕获方法，其特征在于：步骤B中将码片长度为M的基带信号平均分成R段，每段长度为P，且有P＝M/R。 3. the large frequency offset GNSS signal acquisition method according to claim 1 is characterized in that: in step B, the baseband signal that the chip length is M is divided into R sections on average, and every section length is P, and there is P=M/ R. the

4.根据权利要求1所述的大频偏GNSS信号捕获方法，其特征在于：步骤C中将R个相关值补S-R个0后进行S点的FFT运算。 4. The large frequency offset GNSS signal acquisition method according to claim 1, characterized in that: in step C, R correlation values are supplemented with S-R 0s and then the FFT calculation of S point is carried out. the

5.根据权利要求1所述的大频偏GNSS信号捕获方法，其特征在于：步骤D中输出结果取模检测量中，第一项与m值无关，第二项的大小随m变化而变化。 5. the large frequency offset GNSS signal acquisition method according to claim 1, is characterized in that: in the step D, the output result modulo In the detection quantity, the first item has nothing to do with the value of m, and the size of the second item changes with the change of m.

6.根据权利要求1所述的大频偏GNSS信号捕获方法，其特征在于：步骤E中比较FFT的输出结果与判决门限值，若小于门限值，则重新执行步骤B。 6. The large frequency offset GNSS signal acquisition method according to claim 1, characterized in that: in the step E, compare the output result of the FFT with the decision threshold value, if it is less than the threshold value, then re-execute the step B. the